IRU3065CLTR

Data Sheet No. PD94703 revA IRU3065(PbF) POSITIVE TO NEGATIVE DC TO DC CONTROLLER PRODUCT DATASHEET FEATURES Generate Negative Output from +5V Input 1A Maximum Output Current 1.5MHz maximum Switching Frequency Few External Components Available in 6-Pin SOT-23 DESCRIPTION The IRU3065 controller is designed to provide solutions for the applications requiring low power on board switching regulators. The IRU3065 is specifically designed for positive to negative conversion and uses few components for a simple solution. The IRU3065 operates at high switching frequency (up to 1.5MHz), resulting in smaller magnetics. The output voltage can be set by using an external resistor divider. The stability over all conditions is inherent with this architecture without any compensation. The device is available in the standard 6-Pin SOT-23. APPLICATIONS Hard Disk Drives Blue Laser for DVD R-W MR Head Bias LCD Bias GaAs FET Bias Positive-to-Negative Conversion TYPICAL APPLICATION 5V D1 BAT54 VDD Vcc C3 100pF C1 1uF C4 10uF U1 VGATE IRU3065 Gnd Q1 IRLML5203 D2 C6 10uF VOUT (-5V) L1 1.2uH R1 0.1 10BQ015 VSEN R2 10K ISEN VREF = 5V R3 10K VOUT = -VREF × R3 R2 Figure 1 - Typical application of IRU3065 for single input voltage. PACKAGE ORDER INFORMATION Basic Part (Non Lead-Free) TA (°C) 0 To 70 TA (°C) 0 To 70 DEVICE IRU3065CLTR DEVICE IRU3065CLTRPbF PACKAGE 6-Pin SOT-23 (L6) Lead-Free Part PACKAGE 6-Pin SOT-23 (L6) www.irf.com OUTPUT VOLTAGE Adjustable OUTPUT VOLTAGE Adjustable 1 IRU3065(PbF) ABSOLUTE MAXIMUM RATINGS Vcc ......................................................................... 7V VDD ......................................................................... 12V Operating Junction Temperature Range ..................... 0°C To 125°C Operating Ambient Temperature Range ..................... 0°C To 70°C Storage Temperature Range ...................................... -65°C To +150°C ESD Capability (Human Body Model) ........................ 2000V PACKAGE INFORMATION 6-PIN PLASTIC SOT-23 (L6) TOP VIEW VGATE 1 Gnd 2 VSEN 3 6 Vcc 5 VDD 4 ISEN θJA=230 C/W ELECTRICAL SPECIFICATIONS Unless otherwise specified, these specifications apply over Vcc=5V, VDD=7V, CGATE=470pF, RSEN=0.125Ω, RFDBK1=RFDBK2=10KΩ (to Vcc), fs=1.2MHz, IFL=0.25A and TJ=0°C to 125°C. Typical values refer to TJ=25°C. PARAMETER SYM TEST CONDITION Recommended Vcc Supply Vcc Note.1 Recommended VDD Supply VDD Operating Current Icc Initial Output Voltage Accuracy Measured in application TJ=25 C, Vout=-5V Output Accuracy Measured in application over temp. Vout=-5V. Voltage Feedback Sense VVSEN Voltage Feedback Input Offset VVoff Voltage Feedback Bias Current IV BIAS Peak Current Sense Voltage VIs Min Current Sense Voltage VIs Current Sense Bias Current IIBIAS Output Drivers Section Switching Frequency Note. 1 fs Max Output Duty Cycle Dmax Min Output Duty Cycle Dmin 10% to 90% Vgate high Rise Time Tr Fall Time Tf 90% to 10% Vgate going low Propagation Delay from TD Vsens=1V. Isens from 0 to 250mV. Delay time between Current Sense to Output 90% of Isens to 10% of Vgate MIN 4 4 -1% -2% 0 -10 145 50 2 1.5 100 0 40 40 100 10 2 TYP 5 3 1% +2% V mV µA mV mV µA MHz % % ns ns ns MAX UNITS V V mA Note. 1. guarantted by design 2 www.irf.com IRU3065(PbF) PIN DESCRIPTIONS PIN# 1 2 3 4 5 6 PIN SYMBOL VGATE Gnd VSEN ISEN VDD Vcc PIN DESCRIPTION Output driver for external P Channel MOSFET. This pin serves as ground pin and must be connected to the ground plane. A resistor divider from this pin to VOUT and Vcc or an external VREF, sets the output voltage. This pin sets the maximum load current by sensing the inductor current. This pin provides biasing for the output driver. This pin provides biasing for the internal blocks of the IC. BLOCK DIAGRAM Vcc 6 VDD 5 S Q R 1 VGATE 4 ISEN 3 VSEN 2 Gnd Figure 2 - Simplified block diagram of the IRU3065. www.irf.com 3 IRU3065(PbF) APPLICATION INFORMATION Introduction The IRU3065 is a controller intended for an inverting regulator solution. For example, to generate –5V from a 5V supply. The controller is simple and only has a voltage comparator, current hysteretic comparator, flipflop and MOSFET driver. It controls a typical buck boost converter configured by a P-channel MOSFET, an inductor, a diode and an output capacitor. The sensed inductor current by a sensing resistor compares with current comparator. The current comparator uses hysteresis to control the turn-on and turn-off of the transistor based upon the inductor current and gated by the output voltage level. When the inductor current rises past the hysteresis set point, the output of the current comparator goes high. The flip-flop is reset and the Pchannel MOSFET is turned off. In the mean time, the current sense reference is reduced to near zero, giving a zero reference threshold voltage level. As the inductor current passes below this threshold, which indicates that the inductor’s stored energy has been transferred to the output capacitor, the current comparator output goes high and turns on the output transistor (if the output voltage is low). By means of hysteresis, the inductor charges and discharges and functions as self oscillating. The voltage feedback comparator acts as a demand governor to maintain the output voltage at the desired level. By hysteresis control, the maximum switch current (also equals inductor current) is limited by the internal current sensing reference. The power limit is automatically achieved. The switching frequency is determined by a combination of factors including the inductance, output load current level and peak inductor current. The theoretical output voltage and switching frequency versus output current is shown in Figure 3. Output voltage Regulation mode Power limit mode When the output current is below a critical current IOCP, the output voltage is regulated at the desired value and the switching frequency increases as output current increases. At current IOCP, the switching frequency reaches its maximum fS(MAX). In this region, the operation is in regulation mode. When the current goes above IOCP, the operation goes into power limit mode. The output voltage starts to decrease and the output power is limited. The switching frequency is also reduced. Analysis shows that the current IOCP is determined by: VISEN(TH) VIN IOCP = 1 × × Rs VIN-VOUT(NOM)+VD 2 --(1) Where: Rs = Current Sensing Resistance VISEN(TH) = Upper Threshold Voltage at the current comparator (when Vcc=5V, VISEN(TH)=0.145V) VIN = Input Voltage VD = Diode Forward Voltage VOUT(NOM) = Nominal Output Voltage The maximum switching frequency is determined by: fS(MAX) = fS(MAX) = VIN×(VD-VOUT(NOM)) (VIN+VD-VOUT(NOM)×L×IPEAK VIN×(VD-VOUT(NOM))×RS VISEN(TH)×(VIN+VD-VOUT(NOM))×L ---(2) Where: IPEAK = Peak Inductor Current IPEAK is determined by: IPEAK = VISEN(TH) RS ---(3) The detailed operation can be seen in the theoretical operation section Vout f s max Switching frequency I out fs I out I ocp Figure 3 - Theoretical output voltage and switching frequency vs. output current. 4 www.irf.com IRU3065(PbF) APPLICATION EXAMPLE Design Example The following design example is for the evaluation board application for IRU3065. The schematic is shown in figure 1: Where: VIN = 5V VOUT(NOM) = -5V IOUT = 200mA fS(MAX) = Maximum Frequency fS(MAX) = 1.2MHz VD = Diode Forward Voltage VD = 0.5V Vcc = 5V VISEN(TH)=145mV ≅ 150mV Voltage Sensing Resistor The output voltage is determined by the two voltage sensing resistors R2 and R3: VOUT(NOM) = - R3 × VREF R2 If R3 is chosen as 10K, Then R2 is given by: VREF 5V R2 = × R3 = × 10K = 10KΩ VOUT(NOM) -5V Current Sensing Resistor RS In order to select RS, the desired critical current IOCP has to be determined. Considering the switching losses, for conservative, the critical current should select to be slightly greater than the nominal output current. Select: IOCP = 200mA×1.5 = 300mA Where 1.5 is the coefficient to take the efficiency into account. According to equation (1), the current IOCP is given by: IOCP = VIN 1 0.15 × × = 300mA RS VIN - VOUT(NOM) + VD 2 VIN 1 0.15 × × 2 IOCP VIN - VOUT(NOM) + VD The modified current IOCP is: 0.15 VIN 1 × × RS VIN + VD - VOUT(NOM) 2 5 1 IOCP = × × 1.5A = 357mA 5 + 0.5 - (-5) 2 IOCP = Output Inductor L The inductance is chosen by equation (2): L L VIN×(VD - VOUT(NOM)) (VIN+VDVOUT(NOM))×fS(MAX)×IPEAK -(-5 - 0.5) 5 × (5 - (-5) + 0.5)×1.2MHz 1.5A = 1.45µH Select L = 1.2µH The maximum inductor current is: IPEAK = 1.5A The maximum average inductor current equals IAVG=(VISENTH_MAX+VISENTH_MIN)/Rs/2 IAVG=(145mV+50mV)/0.1ohm/2=1A MOSFET Selection A P-channel MOSFET is required. The peak current in this case is equal to I PEAK =1.5A. The MOSFET IRLML5203, from international Rectifier with ID=3A and BVDSS=30V, is a good choice. Input Capacitor An input capacitor will help to minimize the induced ripple on the +5V supply. A 1µF to 10µF X7R ceramic capacitor is recommended. Output Capacitor An output capacitor is required to store energy from transfer to the output inductor. Its capacitance and ESR have a great impact on output voltage ripple. A 10µF to 22µF X7R Tantolum or ceramic capacitor is recommended. Output Diode The average diode current equals output current. In this case, select the diode average current larger than 300mA. The lowest block voltage is VIN+(-VOUT). In this case, It is 10V. In order to reduce the switching losses, the Schottky diode is recommended. The diode 10BQ015 from International Rectifier with ID=1A and VBR=15V is a good choice. Other Components In order to speed up the turn off of P-channel MOSFET, a fast diode 1N4148 or a 100ohm resistor and 100pF capacitor is connected to the pin VDD and VGATE as shown The current sensing resistance is calculated as: RS = RS = 5 1 0.15 × × y 0.12Ω 0.3 5 - (-5) + 0.5 2 Select RS = 0.1Ω From equation (3), the modified inductor peak current is: VISEN(TH) IPEAK = = 1.5A RS www.irf.com 5 IRU3065(PbF) in figure 1. The schottky diode can be replaced with a 100Ω resistor (Figure 28.) with a small sacrifice of efficiency but lower cost. Thermal Consideration The thermal design is to ensure maximum junction temperature of IRU3065 will not exceed the maximum operation junction temperature, which is 125 C. The junction temperature can be estimated by the following: TJ = PD×ΘJA+TA≤TJ(MAX) = 125 C Where ΘJA is the thermal resistance from junction to case which is usually provided in the specification. PD is the power dissipation. TA is the ambient temperature. The package thermal resistance of IRU3065 is estimated as 230C/W due to compact package. Assuming the maximum allowed ambient temperature is 70C, the maximum power dissipation of IRU3065 will be PDIOCP), the converter goes into power limit mode. In this mode, the maximum inductor current is limited by the internal current reference VISEN=145mV. Therefore, the turn on time of the PMOS keeps same as equation (7). For turn off time, the inductor current theorectically should decrease from IPEAK to zero if the threshold voltage is close to zero , therefore: t1 = L × IPEAK VISEN × L = -(VOUT - VD) -(VOUT - VD) × RS ---(12) And the peak current is given by: IPEAK = VIN × tON L ---(5) Where tON is the turn on time of the PMOS. Because the switch is turned off when sensed inductor current reaches threshold VISEN, the following equation holds: VIN ×tON = VISEN=150mV ---(6) RS×IPEAK = RS× L VISEN(TH) IPEAK = RS The turn on time of the PMOS can be calculated as: tON L×IPEAK VISEN×L = = VIN RS×VIN ---(7) Where VD is the forward voltage drop of output diode D2. The switching period is given by: TS = tON + t1 = L × IPEAK L × IPEAK + VIN -(VOUT - VD) ---(13) For inductor, by applying voltage and second balance approach, we have: VIN×tON+(VOUT - VD)×t1 = 0 It can be derived as: VIN×tON VISEN×L t1 = -(VOUT - VD) = -(VOUT - VD)×RS ---(8) VIN - VOUT + VD TS = L × IPEAK × -VIN ×(VOUT - VD) Where VD is the forward voltage drop of output diode D2. From Figure 18, the average current of output diode should equals the output current, resulting in: ID(AVG) = 1 t1 × IPEAK × = IOUT 2 TS ---(9) The combination of equations (12) and (13) result in the following: VIN t1 = TS VIN - VOUT + VD ---(14) The output current equals the average diode current, which is: IOUT = IOUT = 1 t1 ×IPEAK× 2 TS 1 VISEN VIN × × 2 RS VIN - VOUT + VD ---(15) 1 Where TS is the switching period and fS = TS Combination of equation (6)(8)(9) results in the relationship between output current and switching frequency: fS = -RS2×(VOUT - VD) ×IOUT×2 VISEN×VISEN×L ---(10) Where the peak current is given by equation (6). Equation (15) can be rewritten as: VOUT = VIN + VD VISEN × VIN 2RS × IOUT ---(16) Because at regulation mode, the output voltage is regulated, i.e. VOUT=VOUT(NOM). Then the equation (10) can be rewritten as: fS = -RS2×(VOUT(NOM) - VD) ×IOUT×2 VISEN×VISEN×L ---(11) The above equation shows that the output voltage at the power limit mode is not regulated. It decreases as the output current increases. 12 www.irf.com IRU3065(PbF) When IOUT=IOCP, the output voltage equals nominal voltage VOUT=VOUT(NOM). From equation (15),we have IOCP = 1 VISEN VIN × × 2 VIN - VOUT +VD RS ---(17) 6 Vout versus output current Output voltage The above equation is used to select the current sensing resistor RS. Substitution of equation (16) into equation (13) results in the relationship between frequency and output current, that is VIN 2×IOUT fS = × 1L×IPEAK I PEAK --(18) The above equation indicates that the switching frequency decreases when output current increases during power limit mode. When IOUT=IOCP, the switching frequency reaches its maximum. Substitution of VOUT=VOUT(NOM) and equation (6) into equation (13) results in the maximum switching frequency: VIN×(VD - VOUT(NOM)) fS(MAX) = (VIN + VD VOUT(NOM))×L×IPEAK VIN×(VD - VOUT(NOM))×RS fS(MAX) = ---(19) VISEN×(VIN + VD VOUT(NOM))×L Therefore, the inductance can be selected according to the maximum desired frequency as shown in the following: VIN×(VD - VOUT(NOM)) (VIN + VD VOUT(NOM))×fS(MAX)×IPEAK 5 4 3 2 1 0 0 0.2 0.4 0.6 0.8 ( ) Output current (A) Predicted (-Vout) Measured -Vout Figure 24- The comparison between predicted and measured output voltage versus output current Switching frequency versus output current Frequency(KHz) 1200 1000 800 600 400 200 0 0 0.2 0.4 0.6 0.8 Output current (amp) Predicted fs(KHz) Experiment fs (kHz) L ---(20) Fig. 24 and Fig.25 shows the theorectical predication and calculation results for the output voltage and frequency versus output current. Figure 25 - The comparison between predicted and measured switching frequency versus output current www.irf.com 13 IRU3065(PbF) Other Applications 5V 100ohm VDD Vcc C1 100pF U1 VGATE IRU3065 Gnd Q1 IRLML5203 L1 1.2uH R1 0.1 D2 C2 10uF 10BQ015 VOUT (-5V) VSEN ISEN VREF= 5V R2 10K R3 10K Fig. 26 . IRU3065 application with 100ohm resistor and 100pf cap 14 www.irf.com IRU3065(PbF) (L6) SOT-23 Package B e L E E1 e1 D α C A2 A C L A1 SYMBOL A A1 A2 B C D E E1 e e1 L MAX MIN 1.45 0.90 0.15 0.00 1.30 0.90 0.50 0.35 0.20 0.09 3.00 2.80 3.00 2.60 1.75 1.50 0.95 REF 1.90 REF 0.60 0.10 10 0 α NOTE: ALL MEASUREMENTS ARE IN MILLIMETERS. IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information Data and specifications subject to change without notice. 9/6/2005 www.irf.com 15