LT1110CS8-5#PBF

LT1110 Micropower DC-DC Converter Adjustable and Fixed 5V, 12V U DESCRIPTIO FEATURES ■ ■ ■ ■ ■ ■ ■ ■ Operates at Supply Voltages From 1.0V to 30V Works in Step-Up or Step-Down Mode Only Three External Off-the-Shelf Components Required Low-Battery Detector Comparator On-Chip User-Adjustable Current Limit Internal 1A Power Switch Fixed or Adjustable Output Voltage Versions Space-Saving 8-Pin MiniDIP or S8 Package UO APPLICATI ■ ■ ■ ■ ■ ■ ■ ■ Pagers Cameras Single-Cell to 5V Converters Battery Backup Supplies Laptop and Palmtop Computers Cellular Telephones Portable Instruments Laser Diode Drivers Hand-Held Inventory Computers The 70kHz oscillator allows the use of surface mount inductors and capacitors in many applications. Quiescent current is just 300µA, making the device ideal in remote or battery powered applications where current consumption must be kept to a minimum. The device can easily be configured as a step-up or step-down converter, although for most step-down applications or input sources greater than 3V, the LT1111 is recommended. Switch current limiting is user-adjustable by adding a single external resistor. Unique reverse battery protection circuitry limits reverse current to safe, nondestructive levels at reverse supply voltages up to 1.6V. UO ■ S The LT1110 is a versatile micropower DC-DC converter. The device requires only three external components to deliver a fixed output of 5V or 12V. The very low minimum supply voltage of 1.0V allows the use of the LT1110 in applications where the primary power source is a single cell. An on-chip auxiliary gain block can function as a low battery detector or linear post regulator. TYPICAL APPLICATI All Surface Mount Single Cell to 5V Converter SUMIDA CD54-470K 47µH Efficiency 90 85 MBRS120T3 5V I LIM 1.5V AA CELL* 2 V IN SW1 3 LT1110-5 GND 5 SENSE 8 SW2 4 + 75 VIN = 1.25V 70 VIN = 1.00V 65 60 15µF TANTALUM 55 50 OPERATES WITH CELL VOLTAGE ≥ 1.0V *ADD 10µF DECOUPLING CAPACITOR IF BATTERY IS MORE THAN 2" AWAY FROM LT1110. VIN = 1.50V 80 EFFICIENCY (%) 1 0 LT1110 • TA01 5 10 15 20 25 30 35 40 LOAD CURRENT (mA) LT1110 • TA02 1 LT1110 U U RATI GS W W W W AXI U U ABSOLUTE PACKAGE/ORDER I FOR ATIO Supply Voltage, Step-Up Mode ................................ 15V Supply Voltage, Step-Down Mode ........................... 36V SW1 Pin Voltage ...................................................... 50V SW2 Pin Voltage ......................................... – 0.5V to VIN Feedback Pin Voltage (LT1110) .............................. 5.5V Switch Current ........................................................ 1.5A Maximum Power Dissipation ............................. 500mW Operating Temperature Range ..................... 0°C to 70°C Storage Temperature Range .................. –65°C to 150°C Lead Temperature (Soldering, 10 sec.)................. 300°C TOP VIEW ILIM 1 8 FB (SENSE)* VIN 2 7 SET SW1 3 6 A0 SW2 4 5 GND N8 PACKAGE 8-LEAD PLASTIC DIP ORDER PART NUMBER LT1110CN8 LT1110CN8-5 LT1110CN8-12 *FIXED VERSIONS TJMAX = 90°C, θJA = 130°C/W TOP VIEW ILIM 1 8 FB (SENSE)* VIN 2 7 SET SW1 3 6 A0 SW2 4 5 GND LT1110CS8 LT1110CS8-5 LT1110CS8-12 S8 PART MARKING 1110 11105 11012 S8 PACKAGE 8-LEAD PLASTIC SOIC *FIXED VERSIONS TJMAX = 90°C, θJA = 150°C/W Consult factory for Industrial and Military grade parts. ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 1.5V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS IQ Quiescent Current Switch Off ● VIN Input Voltage Step-Up Mode ● Step-Down Mode ● Comparator Trip Point Voltage LT1110 (Note 1) ● 210 220 230 mV Output Sense Voltage LT1110-5 (Note 2) ● 4.75 5.00 5.25 V LT1110-12 (Note 2) ● 11.4 12.00 12.6 V Comparator Hysteresis LT1110 ● 4 8 mV Output Hysteresis LT1110-5 ● 90 180 mV LT1110-12 ● VOUT MIN 1.15 1.0 UNITS µA 12.6 12.6 V V 30 V 200 400 mV 52 70 90 kHz ● 62 69 78 % ● 7.5 10 12.5 µs 70 150 nA 100 300 nA 0.15 0.4 V ● 0.35 1.0 %/V ● 0.05 0.1 %/V Oscillator Frequency DC Duty Cycle tON Switch ON Time IFB Feedback Pin Bias Current LT1110, VFB = 0V ● ISET Set Pin Bias Current VSET = VREF ● VAO AO Output Low IAO = –300µA, VSET = 150mV ● Reference Line Regulation 1.0V ≤ VIN ≤ 1.5V 1.5V ≤ VIN ≤ 12V 2 MAX ● fOSC Full Load (VFB < VREF) TYP 300 LT1110 ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 1.5V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN VCESAT Switch Saturation Voltage Step-Up Mode VIN = 1.5V, ISW = 400mA TYP MAX 300 400 600 ● VIN = 1.5V, ISW = 500mA 400 ● VIN = 5V, ISW = 1A 700 AV A2 Error Amp Gain RL = 100kΩ (Note 3) IREV Reverse Battery Current (Note 4) ILIM Current Limit 220Ω Between ILIM and VIN ● 1000 Current Limit Temperature Coefficient ILEAK Switch OFF Leakage Current Measured at SW1 Pin VSW2 Maximum Excursion Below GND ISW1 ≤ 10µA, Switch Off The ● denotes the specifications which apply over the full operating temperature range. Note 1: This specification guarantees that both the high and low trip point of the comparator fall within the 210mV to 230mV range. UNITS mV mV 550 mV 750 mV 1000 mV 5000 V/V 750 mA 400 mA – 0.3 %/°C 1 10 µA – 400 – 350 mV Note 3: 100kΩ resistor connected between a 5V source and the AO pin. Note 4: The LT1110 is guaranteed to withstand continuous application of +1.6V applied to the GND and SW2 pins while VIN, ILIM, and SW1 pins are grounded. Note 2: This specification guarantees that the output voltage of the fixed versions will always fall within the specified range. The waveform at the sense pin will exhibit a sawtooth shape due to the comparator hysteresis. U W TYPICAL PERFOR A CE CHARACTERISTICS Oscillator Frequency Oscillator Frequency 100 Switch On Time 14 80 13 76 80 70 60 12 74 ON TIME (µs) FREQUENCY (KHz) OSCILLATOR FREQUENCY (KHz) 78 90 72 70 68 66 10 9 64 50 8 62 40 –50 11 60 –25 0 25 50 75 100 TEMPERATURE (°C) 0 3 6 9 12 15 18 21 24 27 30 INPUT VOLTAGE (V) LT1110 • TPC01 7 –50 –25 0 25 50 75 100 TEMPERATURE (°C) LT1110 • TPC02 LT1110 • TPC03 3 LT1110 U W TYPICAL PERFOR A CE CHARACTERISTICS 78 500 76 450 74 400 72 350 68 66 200 62 100 60 50 50 25 75 0 50 25 0 100 VIN = 12V OSCILLATOR FREQUENCY (KHz) 1.0 0.8 0.6 0.4 0.2 100 400 95 90 380 85 80 0°C ≤ TA ≤ 70°C 75 70 65 60 55 50 1.0 8 9 10 11 450 1.3 350 300 250 200 0 25 320 300 280 260 240 0 50 75 100 TEMPERATURE (°C) LT1110 • TPC10 3 6 9 12 15 18 21 24 27 30 INPUT VOLTAGE (V) LT1110 • TPC09 Maximum Switch Current vs RLIM Step-Up 1.5 Maximum Switch Current vs RLIM Step-Down 1.3 1.1 STEP-UP MODE VIN ≤ 5V 0.9 0.7 0.5 1.1 0.9 STEP-DOWN MODE VIN = 12V 0.7 0.5 0.3 0.1 –25 1.6 360 13 0.3 150 4 12 SWITCH CURRENT (A) 1.5 SWITCH CURRENT (A) QUIESCENT CURRENT (µA) 500 1.4 340 LT1110 • TPC08 Quiescent Current 1.2 Quiescent Current SWITCH ON TIME (µs) LT1110 • TPC07 400 0.8 1.0 200 7 ISWITCH (A) 100 –50 0.6 220 40 0.8 0.6 0.4 LT1110 • TPC06 45 0 0.4 0.2 ISWITCH (A) Minimum/Maximum Frequency vs On Time 1.4 ON VOLTAGE (V) 75 LT1110 • TPC05 Switch On Voltage Step-Down Mode 0.2 VIN = 1.0V TEMPERATURE (°C) LT1110 • TPC04 1.2 VIN = 5.0V 0 – 25 TEMPERATURE (°C) 0 VIN = 1.2V 0.6 0.2 0 –50 100 0.8 0.4 QUIESCENT CURRENT (µA) 0 VIN = 1.5V 1.0 250 150 VIN = 3.0V VIN= 2.0V 1.2 300 64 –25 1.4 VIN = 1.5V ISW = 500mA VCESAT (V) 70 58 –50 Saturation Voltage Step-Up Mode Switch Saturation Voltage VCESAT (mV) DUTY CYCLE (%) Duty Cycle 0.1 10 100 RLIM (Ω) 1000 LT1110 • TPC11 10 100 RLIM (Ω) 1000 LT1110 • TPC12 LT1110 U W TYPICAL PERFOR A CE CHARACTERISTICS Set Pin Bias Current 120 224 100 120 90 BIAS CURRENT (nA) BIAS CURRENT (nA) Reference Voltage 226 110 140 100 80 60 40 222 80 70 60 50 40 20 –25 0 25 50 75 218 214 10 0 –50 100 220 216 30 20 0 –50 FB Pin Bias Current VREF (mV) 160 –25 0 TEMPERATURE (°C) 25 50 75 212 –50 100 TEMPERATURE (°C) LT1110 • TPC13 –25 0 25 50 75 100 TEMPERATURE (°C) LT1110 • TPC15 LT1110 • TPC14 UO U U PI FU CTI S ILIM (Pin 1): Connect this pin to VIN for normal use. Where lower current limit is desired, connect a resistor between ILIM and VIN. A 220Ω resistor will limit the switch current to approximately 400mA. VIN (Pin 2): Input supply voltage. SW1 (Pin 3): Collector of power transistor. For step-up mode connect to inductor/diode. For step-down mode connect to VIN. SW2 (Pin 4): Emitter of power transistor. For step-up mode connect to ground. For step-down mode connect to inductor/diode. This pin must never be allowed to go more than a Schottky diode drop below ground. GND (Pin 5): Ground. AO (Pin 6): Auxiliary Gain Block (GB) output. Open collector, can sink 300µA. SET (Pin 7): GB input. GB is an op amp with positive input connected to SET pin and negative input connected to 220mV reference. FB/SENSE (Pin 8): On the LT1110 (adjustable) this pin goes to the comparator input. On the LT1110-5 and LT1110-12, this pin goes to the internal application resistor that sets output voltage. W LT1110 BLOCK DIAGRA + SET A2 – V IN AO GAIN BLOCK/ERROR AMP I LIM SW1 220mV REFERENCE A1 COMPARATOR GND FB OSCILLATOR Q1 DRIVER SW2 LT1110 • BD01 5 LT1110 UO LT1110 OPERATI The LT1110 is a gated oscillator switcher. This type architecture has very low supply current because the switch is cycled only when the feedback pin voltage drops below the reference voltage. Circuit operation can best be understood by referring to the LT1110 block diagram above. Comparator A1 compares the FB pin voltage with the 220mV reference signal. When FB drops below 220mV, A1 switches on the 70kHz oscillator. The driver amplifier boosts the signal level to drive the output NPN power switch Q1. An adaptive base drive circuit senses switch current and provides just enough base drive to ensure switch saturation without overdriving the switch, resulting in higher efficiency. The switch cycling action raises the output voltage and FB pin voltage. When the FB voltage is sufficient to trip A1, the oscillator is gated off. A small amount of hysteresis built into A1 ensures loop stability without external frequency compensation. When the comparator is low the oscillator and all high current circuitry is turned off, lowering device quiescent current to just 300µA for the reference, A1 and A2. The oscillator is set internally for 10µs ON time and 5µs OFF time, optimizing the device for step-up circuits where VOUT ≈ 3VIN, e.g., 1.5V to 5V. Other step-up ratios as well as step-down (buck) converters are possible at slight losses in maximum achievable power output. A2 is a versatile gain block that can serve as a low battery detector, a linear post regulator, or drive an under voltage lockout circuit. The negative input of A2 is internally connected to the 220mV reference. An external resistor divider from VIN to GND provides the trip point for A2. The AO output can sink 300µA (use a 47k resistor pull up to +5V). This line can signal a microcontroller that the battery voltage has dropped below the preset level. To prevent the gain block from operating in its linear region, a 2MΩ resistor can be connected from AO to SET. This provides positive feedback. A resistor connected between the ILIM pin and VIN adjusts maximum switch current. When the switch current exceeds the set value, the switch is turned off. This feature is especially useful when small inductance values are used with high input voltages. If the internal current limit of 1.5A is desired, ILIM should be tied directly to VIN. Propagation delay through the current limit circuitry is about 700ns. In step-up mode, SW2 is connected to ground and SW1 drives the inductor. In step-down mode, SW1 is connected to VIN and SW2 drives the inductor. Output voltage is set by the following equation in either step-up or stepdown modes where R1 is connected from FB to GND and R2 is connected from VOUT to FB.  R2  VOUT = 220mV  + 1 .  R1  ( UO W + A2 AO – V IN GAIN BLOCK/ERROR AMP I LIM SW1 220mV REF A1 COMPARATOR R1 6 Q1 DRIVER SW2 R2 300kΩ SENSE GND OSCILLATOR LT1110-5: R1 = 13.8kΩ LT1110-12: R2 = 5.6kΩ LT1110 • BD02 (01) LT1110 --5, -12 OPERATI LT1110-5, -12 BLOCK DIAGRA SET ) The LT1110-5 and LT1110-12 fixed output voltage versions have the gain setting resistors on-chip. Only three external components are required to construct a 5V or 12V output converter. 16µA flows through R1 and R2 in the LT1110-5, and 39µA flows in the LT1110-12. This current represents a load and the converter must cycle from time to time to maintain the proper output voltage. Output ripple, inherently present in gated oscillator designs, will typically run around 90mV for the LT1110-5 and 200mV for the LT1110-12 with the proper inductor/capacitor selection. This output ripple can be reduced considerably by using the gain block amp as a pre-amplifier in front of the FB pin. See the Applications section for details. LT1110 U W U UO APPLICATI S I FOR ATIO Inductor Selection — General A DC-DC converter operates by storing energy as magnetic flux in an inductor core, and then switching this energy into the load. Since it is flux, not charge, that is stored, the output voltage can be higher, lower, or opposite in polarity to the input voltage by choosing an appropriate switching topology. To operate as an efficient energy transfer element, the inductor must fulfill three requirements. First, the inductance must be low enough for the inductor to store adequate energy under the worst case condition of minimum input voltage and switch ON time. The inductance must also be high enough so maximum current ratings of the LT1110 and inductor are not exceeded at the other worst case condition of maximum input voltage and ON time. Additionally, the inductor core must be able to store the required flux; i.e., it must not saturate. At power levels generally encountered with LT1110 based designs, small surface mount ferrite core units with saturation current ratings in the 300mA to 1A range and DCR less than 0.4Ω (depending on application) are adequate. Lastly, the inductor must have sufficiently low DC resistance so excessive power is not lost as heat in the windings. An additional consideration is ElectroMagnetic Interference (EMI). Toroid and pot core type inductors are recommended in applications where EMI must be kept to a minimum; for example, where there are sensitive analog circuitry or transducers nearby. Rod core types are a less expensive choice where EMI is not a problem. Minimum and maximum input voltage, output voltage and output current must be established before an inductor can be selected. Inductor Selection — Step-Up Converter In a step-up, or boost converter (Figure 4), power generated by the inductor makes up the difference between input and output. Power required from the inductor is determined by ( )( PL = VOUT + V D – VIN MIN IOUT ) (01) where VD is the diode drop (0.5V for a 1N5818 Schottky). Energy required by the inductor per cycle must be equal or greater than PL (02) fOSC in order for the converter to regulate the output. When the switch is closed, current in the inductor builds according to –R't  V  IL ( t) = IN  1– e L  R'   (03) where R' is the sum of the switch equivalent resistance (0.8Ω typical at 25°C) and the inductor DC resistance. When the drop across the switch is small compared to VIN, the simple lossless equation () V I L t = IN t (04) L can be used. These equations assume that at t = 0, inductor current is zero. This situation is called “discontinuous mode operation” in switching regulator parlance. Setting “t” to the switch ON time from the LT1110 specification table (typically 10µs) will yield IPEAK for a specific “L” and VIN. Once IPEAK is known, energy in the inductor at the end of the switch ON time can be calculated as 1 2 LI (05) 2 PEAK EL must be greater than PL/fOSC for the converter to deliver the required power. For best efficiency IPEAK should be kept to 1A or less. Higher switch currents will cause excessive drop across the switch resulting in reduced efficiency. In general, switch current should be held to as low a value as possible in order to keep switch, diode and inductor losses at a minimum. EL = As an example, suppose 12V at 120mA is to be generated from a 4.5V to 8V input. Recalling equation (01), ( )( ) P L = 12 V + 0.5 V – 4.5 V 120mA = 960mW. (06) Energy required from the inductor is 960mW PL = = 13.7µJ. f OSC 70kHz (07) 7 LT1110 U W U UO APPLICATI S I FOR ATIO VOUT = output voltage VIN = minimum input voltage Picking an inductor value of 47µH with 0.2Ω DCR results in a peak switch current of I PEAK = −1.0 W•10ms  4.5 V  1 − e 47mH  = 862mA. 1.0 W   (08) Substituting IPEAK into Equation 05 results in EL = ( )( ) 1 47µH 0.862 A 2 = 17.5µJ. 2 Once IPEAK is known, inductor value can be derived from (09) Since 17.5µJ > 13.7µJ, the 47µH inductor will work. This trial-and-error approach can be used to select the optimum inductor. Keep in mind the switch current maximum rating of 1.5A. If the calculated peak current exceeds this, an external power transistor can be used. A resistor can be added in series with the ILIM pin to invoke switch current limit. The resistor should be picked such that the calculated IPEAK at minimum VIN is equal to the Maximum Switch Current (from Typical Performance Characteristic curves). Then, as VIN increases, switch current is held constant, resulting in increasing efficiency. Inductor Selection — Step-Down Converter After establishing output voltage, output current and input voltage range, peak switch current can be calculated by the formula 2 IOUT  V OUT + V D  DC  V IN – V SW + V D  where DC = duty cycle (0.69) VSW = switch drop in step-down mode VD = diode drop (0.5V for a 1N5818) IOUT = output current 8 L= V IN MIN – V SW – V OUT • tON IPEAK (11) where tON = switch ON time (10µs). Next, the current limit resistor RLIM is selected to give IPEAK from the RLIM Step-Down Mode curve. The addition of this resistor keeps maximum switch current constant as the input voltage is increased. As an example, suppose 5V at 250mA is to be generated from a 9V to 18V input. Recalling Equation (10), IPEAK = ( ) 2 250mA  5 + 0.5  = 498mA . (12) 0.69  9 – 1.5 + 0.5  Next, inductor value is calculated using Equation (11) The step-down case (Figure 5) differs from the step-up in that the inductor current flows through the load during both the charge and discharge periods of the inductor. Current through the switch should be limited to ~800mA in this mode. Higher current can be obtained by using an external switch (see Figure 6). The ILIM pin is the key to successful operation over varying inputs. IPEAK = VSW is actually a function of switch current which is in turn a function of VIN, L, time and VOUT. To simplify, 1.5V can be used for VSW as a very conservative value. (10) L= 9 – 1.5 – 5 • 10µs = 50µH. 498mA (13) Use the next lowest standard value (47µH). Then pick RLIM from the curve. For IPEAK = 500mA, RLIM = 82Ω. Inductor Selection — Positive-to-Negative Converter Figure 7 shows hookup for positive-to-negative conversion. All of the output power must come from the inductor. In this case, ( )( ) P L = | VOUT | + V D IOUT . (14) In this mode the switch is arranged in common collector or step-down mode. The switch drop can be modeled as a 0.75V source in series with a 0.65Ω resistor. When the LT1110 U W U UO APPLICATI S I FOR ATIO () IL + = VL R' –R't   – 1 e L     (15) where R' = 0.65Ω + DCRL VL = VIN – 0.75V As an example, suppose –5V at 75mA is to be generated from a 4.5V to 5.5V input. Recalling Equation (14), ( )( ) P L = | −5 V | + 0.5 V 75mA = 413mW. Energy required from the inductor is 413mW PL = = 5.9µJ. 70kHz fOSC (16) (17) Picking an inductor value of 56µH with 0.2Ω DCR results in a peak switch current of IPEAK = (4.5V – 0.75V) 1 – e –0.85Ω • 10µs  = 621mA . 56µH   (0.65Ω + 0.2Ω)  (18) capacitors provide still better performance at more expense. We recommend OS-CON capacitors from Sanyo Corporation (San Diego, CA). These units are physically quite small and have extremely low ESR. To illustrate, Figures 1, 2 and 3 show the output voltage of an LT1110 based converter with three 100µF capacitors. The peak switch current is 500mA in all cases. Figure 1 shows a Sprague 501D, 25V aluminum capacitor. VOUT jumps by over 120mV when the switch turns off, followed by a drop in voltage as the inductor dumps into the capacitor. This works out to be an ESR of over 240mΩ. Figure 2 shows the same circuit, but with a Sprague 150D, 20V tantalum capacitor replacing the aluminum unit. Output jump is now about 35mV, corresponding to an ESR of 70mΩ. Figure 3 shows the circuit with a 16V OS-CON unit. ESR is now only 20mΩ. 50mV/DIV switch closes, current in the inductor builds according to Substituting IPEAK into Equation (04) results in EL = ( )( ) 1 56µH 0.621A 2 = 10.8µJ. 2 5µs/DIV (19) LT1110 • TA19 Figure 1. Aluminum With this relatively small input range, RLIM is not usually necessary and the ILIM pin can be tied directly to VIN. As in the step-down case, peak switch current should be limited to ~800mA. 50mV/DIV Since 10.8µJ > 5.9µJ, the 56µH inductor will work. Capacitor Selection LT1110 • TA20 Figure 2. Tantalum 50mV/DIV Selecting the right output capacitor is almost as important as selecting the right inductor. A poor choice for a filter capacitor can result in poor efficiency and/or high output ripple. Ordinary aluminum electrolytics, while inexpensive and readily available, may have unacceptably poor Equivalent Series Resistance (ESR) and ESL (inductance). There are low ESR aluminum capacitors on the market specifically designed for switch mode DC-DC converters which work much better than general-purpose units. Tantalum 5µs/DIV 5µs/DIV LT1110 • TA21 Figure 3. OS-CON 9 LT1110 W U U UO APPLICATI S I FOR ATIO Diode Selection Speed, forward drop, and leakage current are the three main considerations in selecting a catch diode for LT1110 converters. General purpose rectifiers such as the 1N4001 are unsuitable for use in any switching regulator application. Although they are rated at 1A, the switching time of a 1N4001 is in the 10µs-50µs range. At best, efficiency will be severely compromised when these diodes are used; at worst, the circuit may not work at all. Most LT1110 circuits will be well served by a 1N5818 Schottky diode, or its surface mount equivalent, the MBRS130T3. The combination of 500mV forward drop at 1A current, fast turn ON and turn OFF time, and 4µA to 10µA leakage current fit nicely with LT1110 requirements. At peak switch currents of 100mA or less, a 1N4148 signal diode may be used. This diode has leakage current in the 1nA-5nA range at 25°C and lower cost than a 1N5818. (You can also use them to get your circuit up and running, but beware of destroying the diode at 1A switch currents.) Immediately after switch turn off, the SW1 voltage pin starts to rise because current cannot instantaneously stop flowing in L1. When the voltage reaches VOUT + VD, the inductor current flows through D1 into C1, increasing VOUT. This action is repeated as needed by the LT1110 to keep VFB at the internal reference voltage of 220mV. R1 and R2 set the output voltage according to the formula  R2  VOUT =  1 +  220mV . R1  ( ) (21) Step-Down (Buck Mode) Operation A step-down DC-DC converter converts a higher voltage to a lower voltage. The usual hookup for an LT1110 based step-down converter is shown in Figure 5. VIN R3 220 Ω + C2 I LIM V IN SW1 FB Step-Up (Boost Mode) Operation LT1110 L1 A step-up DC-DC converter delivers an output voltage higher than the input voltage. Step-up converters are not short circuit protected since there is a DC path from input to output. The usual step-up configuration for the LT1110 is shown in Figure 4. The LT1110 first pulls SW1 low causing VIN – VCESAT to appear across L1. A current then builds up in L1. At the end of the switch ON time the current in L1 is1: VIN IPEAK = t ON (20) L L1 V OUT R3* I LIM V IN SW1 LT1110 GND R2 + C1 FB SW2 R1 * = OPTIONAL LT1110 • TA14 Figure 4. Step-Up Mode Hookup. 10 GND R2 D1 1N5818 + C1 R1 LT1110 • TA15 Figure 5. Step-Down Mode Hookup When the switch turns on, SW2 pulls up to VIN – VSW. This puts a voltage across L1 equal to VIN – VSW – VOUT, causing a current to build up in L1. At the end of the switch ON time, the current in L1 is equal to D1 V IN VOUT SW2 IPEAK = VIN − VSW − VOUT L t ON . (22) When the switch turns off, the SW2 pin falls rapidly and actually goes below ground. D1 turns on when SW2 reaches 0.4V below ground. D1 MUST BE A SCHOTTKY DIODE. The voltage at SW2 must never be allowed to go below –0.5V. A silicon diode such as the 1N4933 will allow SW2 to go to –0.8V, causing potentially destructive power Note 1: This simple expression neglects the effects of switch and coil resistance. This is taken into account in the “Inductor Selection” section. LT1110 W U U UO APPLICATI S I FOR ATIO dissipation inside the LT1110. Output voltage is determined by  R2  VOUT =  1 +  220mV . R1  ( ) (23) R3 programs switch current limit. This is especially important in applications where the input varies over a wide range. Without R3, the switch stays on for a fixed time each cycle. Under certain conditions the current in L1 can build up to excessive levels, exceeding the switch rating and/or saturating the inductor. The 220Ω resistor programs the switch to turn off when the current reaches approximately 800mA. When using the LT1110 in stepdown mode, output voltage should be limited to 6.2V or less. Higher output voltages can be accommodated by inserting a 1N5818 diode in series with the SW2 pin (anode connected to SW2). Converter” section with the following conservative expression for VSW: V SW = V R1 + V SAT ≈ 0.9 V . (24) R2 provides a current path to turn off Q1. R3 provides base drive to Q1. R4 and R5 set output voltage. Inverting Configurations The LT1110 can be configured as a positive-to-negative converter (Figure 7), or a negative-to-positive converter (Figure 8). In Figure 7, the arrangement is very similar to a step-down, except that the high side of the feedback is referred to ground. This level shifts the output negative. As in the step-down mode, D1 must be a Schottky diode, and VOUTshould be less than 6.2V. More negative output voltages can be accommodated as in the prior section. +VIN R3 Higher Current Step-Down Operation Output current can be increased by using a discrete PNP pass transistor as shown in Figure 6. R1 serves as a current limit sense. When the voltage drop across R1 equals a VBE, the switch turns off. For temperature compensation a Schottky diode can be inserted in series with the ILIM pin. This also lowers the maximum drop across R1 to VBE – VD, increasing efficiency. As shown, switch current is limited to 2A. Inductor value can be calculated based on formulas in the “Inductor Selection Step-Down Q1 MJE210 OR ZETEX ZTX789A R1 0.3Ω VIN 25V MAX L1 VOUT R2 220 + VIN FB + C2 LT1110 L1 SW2 GND R1 D1 1N5818 + C1 R2 –VOUT LT1110 • TA03 Figure 7. Positive-to-Negative Converter In Figure 8, the input is negative while the output is positive. In this configuration, the magnitude of the input voltage can be higher or lower than the output voltage. A level shift, provided by the PNP transistor, supplies proper polarity feedback information to the regulator. D1 + + C1 C1 I LIM LT1110 R4 + FB GND SW1 +VOUT D1 1N5821 SW1 C2 V IN L1 R3 330 IL I LIM C2 SW2 R5 VOUT = 220mV ( R4 1 + R5 ) Figure 6. Q1 Permits Higher-Current Switching. LT1110 Functions as Controller. 2N3906 LT1110 AO GND LT1110 • TA16 V IN SW1 R1 FB SW2 R2 VOUT = 220mV + 0.6V ( R1 R2 ) –VIN LT1110 • TA04 Figure 8. Negative-to-Positive Converter 11 LT1110 U W U UO APPLICATI S I FOR ATIO Using the ILIM Pin The LT1110 switch can be programmed to turn off at a set switch current, a feature not found on competing devices. This enables the input to vary over a wide range without exceeding the maximum switch rating or saturating the inductor. Consider the case where analysis shows the LT1110 must operate at an 800mA peak switch current with a 2.0V input. If VIN rises to 4V, peak current will rise to 1.6A, exceeding the maximum switch current rating. With the proper resistor selected (see the “Maximum Switch Current vs RLIM” characteristic), the switch current will be limited to 800mA, even if the input voltage increases. switch ON times less than 3µs. Resistor values programming switch ON time for 800ns or less will cause spurious response in the switch circuitry although the device will still maintain output regulation. IL SWITCH ON OFF LT1110 • TA05 Another situation where the ILIM feature is useful occurs when the device goes into continuous mode operation. This occurs in step-up mode when VOUT + VDIODE VI N − VSW < 1 . 1 − DC IL (25) When the input and output voltages satisfy this relationship, inductor current does not go to zero during the switch OFF time. When the switch turns on again, the current ramp starts from the non-zero current level in the inductor just prior to switch turn on. As shown in Figure 9, the inductor current increases to a high level before the comparator turns off the oscillator. This high current can cause excessive output ripple and requires oversizing the output capacitor and inductor. With the ILIM feature, however, the switch current turns off at a programmed level as shown in Figure 10, keeping output ripple to a minimum. Figure 11 details current limit circuitry. Sense transistor Q1, whose base and emitter are paralleled with power switch Q2, is ratioed such that approximately 0.5% of Q2’s collector current flows in Q1’s collector. This current is passed through internal 80Ω resistor R1 and out through the ILIM pin. The value of the external resistor connected between ILIM and VIN set the current limit. When sufficient switch current flows to develop a VBE across R1 + RLIM, Q3 turns on and injects current into the oscillator, turning off the switch. Delay through this circuitry is approximately 800ns. The current trip point becomes less accurate for 12 Figure 9. No Current Limit Causes Large Inductor Current Build-Up SWITCH PROGRAMMED CURRENT LIMIT ON OFF LT1110 • TA06 Figure 10. Current Limit Keeps Inductor Current Under Control RLIM (EXTERNAL) VIN ILIM R1 80Ω (INTERNAL) Q3 SW1 DRIVER OSCILLATOR Q1 Q2 SW2 LT1110 • TA17 Figure 11. LT1110 Current Limit Circuitry Using the Gain Block The gain block (GB) on the LT1110 can be used as an error amplifier, low battery detector or linear post regulator. The gain block itself is a very simple PNP input op amp with an open collector NPN output. The negative input of the gain block is tied internally to the 220mV reference. The positive input comes out on the SET pin. LT1110 W U U UO APPLICATI S I FOR ATIO Arrangement of the gain block as a low battery detector is straightforward. Figure 12 shows hookup. R1 and R2 need only be low enough in value so that the bias current of the SET input does not cause large errors. 33kΩ for R2 is adequate. R3 can be added to introduce a small amount of hysteresis. This will cause the gain block to “snap” when the trip point is reached. Values in the 1M-10M range are optimal. The addition of R3 will change the trip point, however. +5V VIN R1 47k L1 D1 LT1110 220mV REF VBAT Output ripple of the LT1110, normally 90mV at 5VOUT can be reduced significantly by placing the gain block in front of the FB input as shown in Figure 13. This effectively reduces the comparator hysteresis by the gain of the gain block. Output ripple can be reduced to just a few millivolts using this technique. Ripple reduction works with stepdown or inverting modes as well. For this technique to be effective, output capacitor C1 must be large, so that each switching cycle increases VOUT by only a few millivolts. 1000µF is a good starting value. SET V OUT R3 270k – AO TO PROCESSOR + V IN SW1 I LIM AO VBAT + C1 LT1110 GND FB GND R2 SET SW2 R1 R3 R1 = R2 – 220mV ( VLB4.33µA ) VLB = BATTERY TRIP POINT R2 = 33kΩ R3 = 2MΩ ( )( ) VOUT = R2 + 1 220mV R1 LT1110 • TA08 LT1110 • TA07 Figure 12. Setting Low Battery Detector Trip Point Table 1. Inductor Manufacturers Figure 13. Output Ripple Reduction Using Gain Block Table 2. Capacitor Manufacturers MANUFACTURER PART NUMBERS MANUFACTURER PART NUMBERS Coiltronics International 984 S.W. 13th Court Pompano Beach, FL 33069 305-781-8900 CTX100-4 Series Surface Mount Sanyo Video Components 2001 Sanyo Avenue San Diego, CA 92173 619-661-6835 OS-CON Series Sumida Electric Co. USA 708-956-0666 CD54 CDR74 CDR105 Surface Mount Nichicon America Corporation 927 East State Parkway Schaumberg, IL 60173 708-843-7500 PL Series Sprague Electric Company Lower Main Street Sanford, ME 04073 207-324-4140 150D Solid Tantalums 550D Tantalex Matsuo 714-969-2491 267 Series Surface Mount Table 3. Transistor Manufacturers MANUFACTURER PART NUMBERS Zetex Commack, NY 516-543-7100 ZTX Series FZT Series Surface Mount 13 LT1110 UO TYPICAL APPLICATI S All Surface Mount Flash Memory VPP Generator L1* 47µH +5V ±10% MBRS12OT3 MMBT4403 10k V IN SW1 I LIM + 22µF LT1110CS8-12 1k SENSE SW2 GND 1 = PROGRAM 0 = SHUTDOWN + MMBF170 VPP 12V 120MA 47µF 20V LT1110 • TA18 *L1= SUMIDA CD105-470M 1.5V Powered Laser Diode Driver TOSHIBA TOLD-9211 22nF 220Ω 4.7k 2N3906 1N4148 1 I LIM 6 AO 1.5V 2 V IN 3 SW1 + 10Ω C1 100 µ F OS-CON 0.22 µ F CERAMIC 2Ω LT1110 8 * ADJUST R1 MJE210 FB GND 5 SET SW2 4 1N5818 7 1k* R1 L1✝ 2.2 µ H FOR CHANGE IN LASER OUTPUT POWER ✝ TOKO 262LYF-0076M • LASER DIODE CASE COMMON TO +BATTERY TERMINAL • 170mA CURRENT DRAIN FROM 1.5V CELL (50mA DIODE) • NO OVERSHOOT 1.5V Powered Laser Diode Driver 14 LT1110 • TA13 LT1110 UO TYPICAL APPLICATI S All Surface Mount 3V to 5V Step-Up Converter All Surface Mount 9V to 5V Step-Down Converter L1* 47µH 220 220 V IN SW1 I LIM I LIM 3V 2x AA CELL V IN SW1 MBRL120 LT1110-5 9V LT1110-5 GND SENSE SW2 5V 40mA + SENSE SW2 GND L1* 47µH 10µF MBRL120 5V 40mA + 10µF *L1 = COILCRAFT 1812LS-473 LT1110 • TA09 *L1 = COILCRAFT 1812LS-473 LT1110 • TA10 All Surface Mount 1.5V to +10V, +5V Dual Output Step-Up Converter L1* 82µH I LIM 1.5V AA OR AAA CELL L1* 82µH 4.7µF + +10V 3mA V IN SW1 I LIM 490k LT1110 GND All Surface Mount 1.5V to ±5V Dual Output Step-Up Converter +5V 3mA FB SW2 1.5V AA OR AAA CELL V IN SW1 4.7µF 11k 4.7µF LT1110 GND +5V 4mA SENSE SW2 + + + –5V 4mA + 4.7µF 4.7µF + = MBRL120 = MBRL120 4.7µF LT1110 • TA12 *L1 = COILCRAFT 1812LS-823 *L1 = COILCRAFT 1812LS-823 LT1110 • TA11 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT1110 U PACKAGE DESCRIPTIO Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead Plastic DIP 0.300 – 0.320 (7.620 – 8.128) 0.009 – 0.015 (0.229 – 0.381) ( +0.025 0.325 –0.015 +0.635 8.255 –0.381 ) 0.130 ± 0.005 (3.302 ± 0.127) 0.045 – 0.065 (1.143 – 1.651) 0.400 (10.160) MAX 8 7 6 5 0.065 (1.651) TYP 0.250 ± 0.010 (6.350 ± 0.254) 0.125 (3.175) MIN 0.045 ± 0.015 (1.143 ± 0.381) 0.020 (0.508) MIN 1 2 4 3 0.018 ± 0.003 (0.457 ± 0.076) 0.100 ± 0.010 (2.540 ± 0.254) S8 Package 8-Lead Plastic SOIC 0.189 – 0.197* (4.801 – 5.004) 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.053 – 0.069 (1.346 – 1.752) 0°– 8° TYP 0.016 – 0.050 0.406 – 1.270 0.014 – 0.019 (0.355 – 0.483) *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006 INCH (0.15mm). 16 Linear Technology Corporation 8 7 6 5 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) BSC 0.150 – 0.157* (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) 1 2 3 4 LT/GP 0594 2K REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7487 (408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977  LINEAR TECHNOLOGY CORPORATION 1994