LT1371
500kHz High Efficiency
3A Switching Regulator
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DESCRIPTION
FEATURES
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The LT ®1371 is a monolithic high frequency current mode
switching regulator. It can be operated in all standard
switching configurations including boost, buck, flyback,
forward, inverting and “Cuk.” A 3A high efficiency switch
is included on the die, along with all oscillator, control and
protection circuitry.
Faster Switching with Increased Efficiency
Uses Small Inductors: 4.7µH
All Surface Mount Components
Low Minimum Supply Voltage: 2.7V
Quiescent Current: 4mA Typ
Current Limited Power Switch: 3A
Regulates Positive or Negative Outputs
Shutdown Supply Current: 12µA Typ
Easy External Synchronization
The LT1371 typically consumes only 4mA quiescent
current and has higher efficiency than previous parts.
High frequency switching allows for very small inductors
to be used.
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APPLICATIONS
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New design techniques increase flexibility and maintain
ease of use. Switching is easily synchronized to an external logic level source. A logic low on the Shutdown pin
reduces supply current to 12µA. Unique error amplifier
circuitry can regulate positive or negative output voltage
while maintaining simple frequency compensation techniques. Nonlinear error amplifier transconductance reduces output overshoot on start-up or overload recovery.
Oscillator frequency shifting protects external components during overload conditions.
Boost Regulators
Laptop Computer Supplies
Multiple Output Flyback Supplies
Inverting Supplies
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATION
5V to 12V Boost Converter
L1*
4.7µH
D1
MBRS330T3
VIN
OFF
ON
LT1371
+
C1**
22µF
25V
+
FB
GND
C2
0.047µF
R3
2k
VC
R2
6.19k
1%
C4**
22µF
25V
×2
*COILCRAFT DO3316P-472 (4.7µH),
DO3316P-103 (10µH) OR
SUMIDA CD104-100MC (10µH)
**AVX TPSD226M025R0200
†
MAX IOUT
L1
IOUT
4.7µH 0.7A
10µH 0.8A
C3
0.0047µF
LT1371 • TA01
VIN = 5V
90
R1
53.6k
1%
VSW
S/S
100
VOUT†
12V
EFFICIENCY (%)
5V
12V Output Efficiency
80
70
60
50
0.01
0.1
OUTPUT CURRENT (A)
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LT1371 • TA02
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LT1371
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ABSOLUTE
RATI GS
Supply Voltage ....................................................... 30V
Switch Voltage
LT1371 ............................................................... 35V
LT1371HV .......................................................... 42V
S/S, SHDN, SYNC Pin Voltage ................................ 30V
Feedback Pin Voltage (Transient, 10ms) .............. ±10V
Feedback Pin Current ........................................... 10mA
Negative Feedback Pin Voltage
(Transient, 10ms) ............................................. ±10V
Operating Ambient Temperature Range ...... 0°C to 70°C
Operating Junction Temperature Range
Commercial .......................................... 0°C to 125°C
Industrial ......................................... – 40°C to 125°C
Short Circuit ......................................... 0°C to 150°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
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PACKAGE/ORDER I FOR ATIO
ORDER PART
NUMBER
FRONT VIEW
7
6
5
4
3
2
1
TAB
IS
GND
VIN
S/S
VSW
GND
NFB
FB
VC
TOP VIEW
LT1371CR
LT1371HVCR
LT1371IR
LT1371HVIR
R PACKAGE
7-LEAD PLASTIC DD
TJMAX = 125°C, θJA = 30°C/W
WITH PACKAGE SOLDERED TO 0.5 INCH2 COPPER
AREA OVER BACKSIDE GROUND PLANE OR INTERNAL
POWER PLANE. θJA CAN VARY FROM 20°C/W TO
> 40°C/W DEPENDING ON MOUNTING TECHNIQUE
FRONT VIEW
7
6
5
4
3
2
1
TAB
IS
GND
T7 PACKAGE
7-LEAD TO-220
ORDER PART
NUMBER
VIN
S/S
VSW
GND
NFB
FB
VC
ORDER PART
NUMBER
LT1371CT7
LT1371HVCT7
LT1371IT7
LT1371HVIT7
VC 1
20 VSW
FB 2
19 NC
NFB 3
18 VSW
GND 4
17 GND
GND 5
16 GND
GND 6
15 GND
GND 7
14 GND
SHDN 8
13 NC
SYNC 9
12 NC
VIN 10
LT1371CSW
LT1371HVCSW
LT1371ISW
LT1371HVISW
11 GND
SW PACKAGE
20-LEAD PLASTIC SO WIDE
TJMAX = 125°C, θJA = 50°C/W
θJA WILL VARY FROM APPROXIMATELY 40°C/W WITH
0.75 INCH2 OF 1 OZ COPPER TO 50°C/W WITH 0.33 INCH2
OF 1 OZ COPPER ON A DOUBLE-SIDED BOARD
TJMAX = 125°C, θJA = 50°C/W, θJC = 4°C/W
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S, SHDN, SYNC and NFB pins open, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VREF
Reference Voltage
Measured at Feedback Pin
VC = 0.8V
IFB
Feedback Input Current
●
VFB = VREF
MIN
TYP
MAX
UNITS
1.230
1.225
1.245
1.245
1.260
1.265
V
V
250
550
900
nA
nA
0.01
0.03
%/V
●
Reference Voltage Line Regulation
2
2.7V ≤ VIN ≤ 25V, VC = 0.8V
●
LT1371
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S, SHDN, SYNC and NFB pins open, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VNFB
Negative Feedback Reference Voltage
Measured at Negative Feedback Pin
Feedback Pin Open, VC = 0.8V
– 2.540
● – 2.570
INFB
Negative Feedback Input Current
VNFB = VNFR
●
Negative Feedback Reference Voltage
Line Regulation
2.7V ≤ VIN ≤ 25V, VC = 0.8V
●
Error Amplifier Transconductance
∆IC = ±25µA
gm
AV
f
MIN
– 45
UNITS
V
V
– 30
– 15
µA
0.01
0.05
%/V
1100
700
1500
●
1900
2300
µmho
µmho
120
200
350
µA
1400
2400
µA
1.95
0.40
2.30
0.52
V
V
VFB = VREF – 150mV, VC = 1.5V
●
Error Amplifier Sink Current
VFB = VREF + 150mV, VC = 1.5V
●
Error Amplifier Clamp Voltage
High Clamp, VFB = 1V
Low Clamp, VFB = 1.5V
1.70
0.25
VC Pin Threshold
Duty Cycle = 0%
0.8
1
1.25
V
Switching Frequency
2.7V ≤ VIN ≤ 25V
0°C ≤ TJ ≤ 125°C
– 40°C ≤ TJ ≤ 0°C (I Grade)
●
450
430
400
500
500
550
580
580
kHz
kHz
kHz
●
85
95
Error Amplifier Voltage Gain
500
Switch Current Limit Blanking Time
Output Switch Breakdown Voltage
130
V/ V
%
260
ns
LT1371
LT1371HV
0° C ≤ TJ ≤ 125°C
– 40°C ≤ TJ ≤ 0°C (I Grade)
●
35
47
V
●
42
40
47
V
V
0.25
0.45
Ω
3.8
3.4
5.4
5.0
A
A
Supply Current Increase During Switch ON Time
15
25
mA/A
Control Voltage to Switch Current
Transconductance
4
VSAT
Output Switch ON Resistance
ISW = 2A
●
ILIM
Switch Current Limit
Duty Cycle = 50%
Duty Cycle = 80% (Note 1)
●
●
∆IIN
∆ISW
Minimum Input Voltage
IQ
MAX
– 2.440
– 2.410
Error Amplifier Source Current
Maximum Switch Duty Cycle
BV
TYP
– 2.490
– 2.490
3.0
2.6
A/V
●
2.4
2.7
V
Supply Current
2.7V ≤ VIN ≤ 25V
●
4
5.5
mA
Shutdown Supply Current
2.7V ≤ VIN ≤ 25V, VS/S ≤ 0.6V
0° C ≤ TJ ≤ 125°C
– 40°C ≤ TJ ≤ 0°C (I Grade)
●
12
30
50
µA
µA
2.7V ≤ VIN ≤ 25V
●
0.6
1.3
2
V
●
5
12
25
µs
●
– 10
15
µA
●
600
800
kHz
Shutdown Threshold
Shutdown Delay
S/S or SHDN Pin Input Current
Synchronization Frequency Range
The ● denotes specifications which apply over the full operating
temperature range.
0V ≤ VS/S or VSHDN ≤ 5V
Note 1: For duty cycles (DC) between 50% and 90%, minimum
guaranteed switch current is given by ILIM = 1.33 (2.75 – DC).
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LT1371
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TYPICAL PERFORMANCE CHARACTERISTICS
Switch Saturation Voltage
vs Switch Current
Switch Current Limit
vs Duty Cycle
25°C
0.8
0.7
0.6
0.5
–55°C
0.4
0.3
0.2
6
3.0
5
2.8
25°C AND
125°C
4
–55°C
3
2
1
2.4
2.2
2.0
0
2.0
18
1.8
14
1.4
12
1.2
10
1.0
SHUTDOWN DELAY
0.8
6
0.6
4
0.4
2
0.2
0
–50 –25
0
SHUTDOWN THRESHOLD (V)
1.6
MINIMUM SYNCHRONIZATION VOLTAGE (VP-P)
20
8
0
25 50 75 100 125 150
TEMPERATURE (°C)
3.0
400
2.5
2.0
1.5
1.0
0.5
0
–50 –25
2
1
0
–1
–2
–3
–4
–1
0
1
2
3 4 5 6
VOLTAGE (V)
7
8
9
LT1371 • G07
4
125°C
100
0
–100
–200
–300
0
25 50 75 100 125 150
TEMPERATURE (°C)
–0.3
VREF
–0.2
–0.1
FEEDBACK PIN VOLTAGE (V)
Error Amplifier Transconductance
vs Temperature
110
2000
100
1800
90
80
70
60
50
40
30
gm =
∆I (VC)
∆V (FB)
1600
1400
1200
1000
800
600
400
200
20
10
0.1
LT1371 • G06
TRANSCONDUCTANCE (µmho)
SWITCHING FREQUENCY (% OF TYPICAL)
VIN = 5V
25°C
–55°C
200
Switching Frequency
vs Feedback Pin Voltage
3
–5
300
LT1371 • G05
S/S or SHDN Pin Input Current
vs Voltage
4
Error Amplifier Output Current
vs Feedback Pin Voltage
fSYNC = 700kHz
LT1371 • G04
5
25 50 75 100 125 150
TEMPERATURE (°C)
LT1371 • G03
Minimum Synchronization
Voltage vs Temperature
SHUTDOWN THRESHOLD
0
LT1371 • G02
Shutdown Delay and Threshold
vs Temperature
16
1.8
–50 –25
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
0
0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0
SWITCH CURRENT (A)
ERROR AMPLIFIER OUTPUT CURRENT (µA)
0
LT1371 • G01
SHUTDOWN DELAY (µs)
2.6
0.1
0
INPUT CURRENT (µA)
INPUT VOLTAGE (V)
150°C
100°C
SWITCH CURRENT LIMIT (A)
SWITCH SATURATION VOLTAGE (V)
1.0
0.9
Minimum Input Voltage
vs Temperature
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
FEEDBACK PIN VOLTAGE (V)
LT1371 • G08
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1371 • G09
LT1371
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TYPICAL PERFORMANCE CHARACTERISTICS
VC Pin Threshold and High
Clamp Voltage vs Temperature
Feedback Input Current
vs Temperature
2.4
1.8
1.6
1.4
1.2
VC THRESHOLD
1.0
0.8
700
NEGATIVE FEEDBACK INPUT CURRENT (µA)
FEEDBACK INPUT CURRENT (nA)
VC HIGH CLAMP
2.0
VC PIN VOLTAGE (V)
0
800
2.2
VFB =VREF
600
500
400
300
200
100
0.6
0.4
–50 –25
Negative Feedback Input Current
vs Temperature
0
25 50 75 100 125 150
TEMPERATURE (°C)
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1371 • G10
LT1371 • G11
VNFB =VNFR
–10
–20
–30
–40
–50
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1371 • G12
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PIN FUNCTIONS
VC: The Compensation pin is used for frequency compensation, current limiting and soft start. It is the output of the
error amplifier and the input of the current comparator.
Loop frequency compensation can be performed with an
RC network connected from the VC pin to ground.
FB: The Feedback pin is used for positive output voltage
sensing and oscillator frequency shifting. It is the inverting input to the error amplifier. The noninverting input of
this amplifier is internally tied to a 1.245V reference. Load
on the FB pin should not exceed 250µA when NFB pin is
used. See Applications Information.
NFB: The Negative Feedback pin is used for negative
output voltage sensing. It is connected to the inverting
input of the negative feedback amplifier through a 100k
source resistor.
S/S (R and T7 Packages Only): Shutdown and Synchronization Pin. The S/S pin is logic level compatible. Shutdown
is active low and the shutdown threshold is typically 1.3V.
For normal operation, pull the S/S pin high, tie it to VIN or
leave it floating. To synchronize switching, drive the S/S
pin between 600kHz and 800kHz.
SHDN: (SW Package Only): The Shutdown pin is active
low and the shutdown threshold is typically 1.3V. For
normal operation, pull the SHDN pin high, tie it to VIN or
leave it floating.
SYNC (SW Package Only): To synchronize switching,
drive the SYNC pin between 600kHz and 800kHz. If not
used, the SYNC pin can be tied high, low or left floating.
VIN: Bypass Input Supply pin with a low ESR capacitor,
10µF or more. The regulator goes into undervoltage lockout when VIN drops below 2.5V. Undervoltage lockout
stops switching and pulls the VC pin low.
VSW: The Switch pin is the collector of the power switch
and has large currents flowing through it. Keep the traces
to the switching components as short as possible to
minimize radiation and voltage spikes.
GND: Tie all Ground pins to a good quality ground plane.
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LT1371
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BLOCK DIAGRAM
VIN
SHUTDOWN
DELAY AND RESET
SHDN
SW
LOW DROPOUT
2.3V REG
ANTI-SAT
S/S*
SYNC
SYNC
LOGIC
OSC
DRIVER
SWITCH
5:1 FREQUENCY
SHIFT
+
100k
NFB
NFBA
–
COMP
50k
–
FB
+
1.245V
REF
GND SENSE
+
EA
IA
VC
AV ≈ 6
*R AND T7 PACKAGES ONLY
0.04Ω
–
GND
LT1371 • BD
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OPERATION
The LT1371 is a current mode switcher. This means that
switch duty cycle is directly controlled by switch current
rather than by output voltage. Referring to the block
diagram, the switch is turned ON at the start of each
oscillator cycle. It is turned OFF when switch current
reaches a predetermined level. Control of output voltage is
obtained by using the output of a voltage sensing error
amplifier to set current trip level. This technique has
several advantages. First, it has immediate response to
input voltage variations, unlike voltage mode switchers
which have notoriously poor line transient response.
Second, it reduces the 90° phase shift at mid-frequencies
in the energy storage inductor. This greatly simplifies
closed-loop frequency compensation under widely varying input voltage or output load conditions. Finally, it
allows simple pulse-by-pulse current limiting to provide
maximum switch protection under output overload or
short conditions. A low dropout internal regulator provides a 2.3V supply for all internal circuitry. This low
dropout design allows input voltage to vary from 2.7V to
25V with virtually no change in device performance. A
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500kHz oscillator is the basic clock for all internal timing.
It turns ON the output switch via the logic and driver
circuitry. Special adaptive anti-sat circuitry detects onset
of saturation in the power switch and adjusts driver
current instantaneously to limit switch saturation. This
minimizes driver dissipation and provides very rapid turnoff of the switch.
A 1.245V bandgap reference biases the positive input of
the error amplifier. The negative input of the amplifier is
brought out for positive output voltage sensing. The error
amplifier has nonlinear transconductance to reduce output overshoot on start-up or overload recovery. When
the feedback voltage exceeds the reference by 40mV,
error amplifier transconductance increases 10 times,
which reduces output overshoot. The feedback input also
invokes oscillator frequency shifting, which helps protect components during overload conditions. When the
feedback voltage drops below 0.6V, the oscillator frequency is reduced 5:1. Lower switching frequency allows
full control of switch current limit by reducing minimum
switch duty cycle.
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APPLICATIO S I FOR ATIO
Unique error amplifier circuitry allows the LT1371 to
directly regulate negative output voltages. The negative
feedback amplifier’s 100k source resistor is brought out
for negative output voltage sensing. The NFB pin regulates
at – 2.49V while the amplifier output internally drives the
FB pin to 1.245V. This architecture, which uses the same
main error amplifier, prevents duplicating functions and
maintains ease of use. Consult LTC Marketing for units
that can regulate down to – 1.25V.
The error signal developed at the amplifier output is
brought out externally. This pin (VC) has three different
functions. It is used for frequency compensation, current
limit adjustment and soft starting. During normal regulator operation this pin sits at a voltage between 1V (low
output current) and 1.9V (high output current). The error
amplifier is a current output (gm) type, so this voltage can
be externally clamped for lowering current limit. Likewise, a capacitor coupled external clamp will provide soft
start. Switch duty cycle goes to zero if the VC pin is pulled
below the control pin threshold, placing the LT1371 in an
idle mode.
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APPLICATIO S I FOR ATIO
VOUT
Positive Output Voltage Setting
The LT1371 develops a 1.245V reference (VREF) from the
FB pin to ground. Output voltage is set by connecting the
FB pin to an output resistor divider (Figure 1). The FB pin
bias current represents a small error and can usually be
ignored for values of R2 up to 7k. The suggested value for
R2 is 6.19k. The NFB pin is normally left open for positive
output applications. Positive fixed voltage versions are
available (consult LTC Marketing).
Negative Output Voltage Setting
The LT1371 develops a – 2.49V reference (VNFR) from the
NFB pin to ground. Output voltage is set by connecting the
NFB pin to an output resistor divider (Figure 2). The
–30µA NFB pin bias current (INFB) can cause output
voltage errors and should not be ignored. This has been
accounted for in the formula in Figure 2. The suggested
value for R2 is 2.49k. The FB pin is normally left open for
negative output applications. See Dual Polarity Output
Voltage Sensing for limitations on FB pin loading when
using the NFB pin.
Dual Polarity Output Voltage Sensing
Certain applications benefit from sensing both positive
and negative output voltages. One example is the “Dual
Output Flyback Converter with Overvoltage Protection”
circuit shown in the Typical Applications section. Each
output voltage resistor divider is individually set as described above. When both the FB and NFB pins are used,
R1
FB
PIN
R2
( )
( )
VOUT = VREF 1 + R1
R2
R1 = R2
VOUT
–1
1.245
VREF
LT1371 • F01
Figure 1. Positive Output Resistor Divider
–VOUT
INFB
( )
R1
–VOUT = VNFB 1 + R1 + INFB (R1)
R2
R2
R1 =
NFB
PIN
VNFR
VOUT– 2.49
( )(
2.49 + 30 • 10– 6
R2
)
LT1371 • F02
Figure 2. Negative Output Resistor Divider
the LT1371 acts to prevent either output from going
beyond its set output voltage. For example, in this application if the positive output were more heavily loaded than
the negative, the negative output would be greater and
would regulate at the desired set-point voltage. The positive output would sag slightly below its set-point voltage.
This technique prevents either output from going unregulated high at no load. Please note that the load on the FB
pin should not exceed 250µA when the NFB pin is used.
This situation occurs when the resistor dividers are used
at both FB and NFB. True load on FB is not the full divider
current unless the positive output is shorted to ground.
See Dual Output Flyback Converter application.
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LT1371
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APPLICATIO S I FOR ATIO
Shutdown and Synchronization
The 7-pin R and T7 package devices have a dual function
S/S pin which is used for both shutdown and synchronization. The SW package device has both a Shutdown
(SHDN) pin and a Synchronization (SYNC) pin which can
be used separately or tied together. These pins are logic
level compatible and can be pulled high, tied to VIN or left
floating for normal operation. A logic low on the S/S pin or
SHDN pin activates shutdown, reducing the part’s supply
current to 12µA. Typical synchronization range is from
1.05 to 1.8 times the part’s natural switching frequency,
but is only guaranteed between 600kHz and 800kHz. A
12µs resetable shutdown delay network guarantees the
part will not go into shutdown while receiving a synchronization signal when the functions are combined.
Caution should be used when synchronizing above 700kHz
because at higher sync frequencies the amplitude of the
internal slope compensation used to prevent subharmonic
switching is reduced. This type of subharmonic switching
only occurs when the duty cycle of the switch is above 50%.
Higher inductor values will tend to eliminate problems.
Thermal Considerations
Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause excessive die temperatures. Typical thermal resistance is
30°C/W for the R package and 50°C/W for the SW and T7
packages but these numbers will vary depending on the
mounting techniques (copper area, air flow, etc.). Heat is
transferred from the R and T7 packages via the tab and
from the SW package via pins 4 to 7 and 14 to 17.
Average supply current (including driver current) is:
IIN = 4mA + DC [ISW/60 + ISW (0.004)]
ISW = switch current
DC = switch duty cycle
Switch power dissipation is given by:
PSW = (ISW)2 (RSW)(DC)
RSW = output switch ON resistance
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Total power dissipation of the die is the sum of supply
current times supply voltage, plus switch power:
PD(TOTAL) = (IIN)(VIN) + PSW
Surface mount heat sinks are also becoming available
which can lower package thermal resistance by 2 or 3
times. One manufacturer is Wakefield Engineering who
offers surface mount heat sinks for both the R package
(DD) and SW package (SW20) and can be reached at (617)
245-5900.
Choosing the Inductor
For most applications the inductor will fall in the range of
2.2µH to 22µH. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output
current because they reduce peak current seen by the
power switch, which has a 3A limit. Higher values also
reduce input ripple voltage and reduce core loss.
When choosing an inductor you might have to consider
maximum load current, core and copper losses, allowable
component height, output voltage ripple, EMI, fault
current in the inductor, saturation and, of course, cost.
The following procedure is suggested as a way of handling
these somewhat complicated and conflicting requirements.
1. Assume that the average inductor current for a boost
converter is equal to load current times VOUT / VIN and
decide whether or not the inductor must withstand
continuous overload conditions. If average inductor
current at maximum load current is 1A, for instance, a
1A inductor may not survive a continuous 3A overload
condition. Also be aware that boost converters are not
short-circuit protected and that, under output short
conditions, inductor current is limited only by the
available current of the input supply.
2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t
omit this step. Powdered iron cores are forgiving
because they saturate softly, whereas ferrite cores
LT1371
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saturate abruptly and other core materials fall in between. The following formula assumes continuous
mode operation but it errs only slightly on the high side
for discontinuous mode, so it can be used for all
conditions.
) )
V (V
–V )
V
IPEAK = (IOUT) OUT + IN OUT IN
VIN
2(f)(L)(VOUT)
VIN = Minimum Input Voltage
f = 500kHz Switching Frequency
3. Decide if the design can tolerate an “open” core geometry, like a rod or barrel, which has high magnetic field
radiation, or whether it needs a closed core, like a
toroid, to prevent EMI problems. One would not want an
open core next to a magnetic storage media, for instance! This is a tough decision because the rods or
barrels are temptingly cheap and small and there are no
helpful guidelines to calculate when the magnetic field
radiation will be a problem.
4. Start shopping for an inductor which meets the requirements of core shape, peak current (to avoid
saturation), average current (to limit heating) and fault
current. If the inductor gets too hot, wire insulation will
melt and cause turn-to-turn shorts. Keep in mind that
all good things like high efficiency, low profile and high
temperature operation will increase cost, sometimes
dramatically.
5. After making an initial choice, consider the secondary
things like output voltage ripple, second sourcing, etc.
Use the experts in the LTC Applications Department if
you feel uncertain about the final choice. They have
experience with a wide range of inductor types and can
tell you about the latest developments in low profile,
surface mounting, etc.
Output Capacitor
The output capacitor is normally chosen by its effective
series resistance (ESR), because this is what determines
output ripple voltage. At 500kHz any polarized capacitor
is essentially resistive. To get low ESR takes volume, so
physically smaller capacitors have high ESR. The ESR
range needed for typical LT1371 applications is 0.025Ω
to 0.2Ω. A typical output capacitor is an AVX type TPS,
22µF at 25V (2 each), with a guaranteed ESR less than
0.2Ω. This is a “D” size surface mount solid tantalum
capacitor. TPS capacitors are specially constructed and
tested for low ESR, so they give the lowest ESR for a given
volume. To further reduce ESR, multiple output capacitors can be used in parallel. The value in microfarads is
not particularly critical, and values from 22µF to greater
than 500µF work well, but you cannot cheat mother
nature on ESR. If you find a tiny 22µF solid tantalum
capacitor, it will have high ESR and output ripple voltage
will be terrible. Table 1 shows some typical solid tantalum
surface mount capacitors.
Table 1. Surface Mount Solid Tantalum Capacitor
ESR and Ripple Current
E CASE SIZE
AVX TPS, Sprague 593D
AVX TAJ
ESR (MAX Ω)
RIPPLE CURRENT (A)
0.1 to 0.3
0.7 to 0.9
0.7 to 1.1
0.4
0.1 to 0.3
0.9 to 2.0
0.7 to 1.1
0.36 to 0.24
0.2 (Typ)
1.8 to 3.0
0.5 (Typ)
0.22 to 0.17
2.5 to 10
0.16 to 0.08
D CASE SIZE
AVX TPS, Sprague 593D
AVX TAJ
C CASE SIZE
AVX TPS
AVX TAJ
B CASE SIZE
AVX TAJ
Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true and AVX type TPS capacitors are
specially tested for surge capability, but surge ruggedness
is not a critical issue with the output capacitor. Solid
tantalum capacitors fail during very high turn-on surges,
which do not occur at the output of regulators. High
discharge surges, such as when the regulator output is
dead-shorted, do not harm the capacitors.
Single inductor boost regulators have large RMS ripple
current in the output capacitor, which must be rated to
handle the current. The formula to calculate this is:
9
LT1371
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APPLICATIO S I FOR ATIO
Output Capacitor Ripple Current (RMS)
DC
IRIPPLE (RMS) = IOUT 1 – DC
= IOUT
VOUT – VIN
VIN
DC = Switch Duty Cycle
Input Capacitors
The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular and
does not contain large squarewave currents as is found in
the output capacitor. Capacitors in the range of 10µF to
100µF, with an ESR of 0.2Ω or less, work well up to full 3A
switch current. Higher ESR capacitors may be acceptable
at low switch currents. Input capacitor ripple current for a
boost converter is :
0.3(VIN)(VOUT – VIN)
IRIPPLE =
(f)(L)(VOUT)
f = 500kHz Switching Frequency
The input capacitor can see a very high surge current when
a battery or high capacitance source is connected “live”
and solid tantalum capacitors can fail under this condition.
Several manufacturers have developed tantalum capacitors specially tested for surge capability (AVX TPS series,
for instance) but even these units may fail if the input
voltage approaches the maximum voltage rating of the
capacitor during a high surge. AVX recommends derating
capacitor voltage by 2:1 for high surge applications.
Ceramic, OS-CON and aluminum electrolytic capacitors
may also be used and have a high tolerance to turn-on
surges.
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
10
generates a loop “zero” at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually
resonate with their ESL before ESR becomes effective.
They are appropriate for input bypassing because of their
high ripple current ratings and tolerance of turn-on surges.
Output Diode
The suggested output diode (D1) is a 1N5821 Schottky or
its Motorola equivalent MBR330. It is rated at 3A average
forward current and 30V reverse voltage. Typical forward
voltage is 0.6V at 3A. The diode conducts current only
during switch OFF time. Peak reverse voltage for boost
converters is equal to regulator output voltage. Average
forward current in normal operation is equal to output
current.
Frequency Compensation
Loop frequency compensation is performed on the output
of the error amplifier (VC pin) with a series RC network.
The main pole is formed by the series capacitor and the
output impedance (≈500kΩ) of the error amplifier. The
pole falls in the range of 2Hz to 20Hz. The series resistor
creates a “zero” at 1kHz to 5kHz, which improves loop
stability and transient response. A second capacitor, typically one-tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency
ripple on the VC pin. VC pin ripple is caused by output
voltage ripple attenuated by the output divider and multiplied by the error amplifier. Without the second capacitor,
VC pin ripple is:
VC Pin Ripple =
1.245(VRIPPLE)(gm)(RC)
(VOUT)
VRIPPLE = Output ripple (VP–P)
gm = Error amplifier transconductance
( ≈1500µmho)
RC = Series resistor on VC pin
VOUT = DC output voltage
To prevent irregular switching, VC pin ripple should be
kept below 50mVP–P. Worst-case VC pin ripple occurs at
LT1371
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APPLICATIO S I FOR ATIO
maximum output load current and will also be increased if
poor quality (high ESR) output capacitors are used. The
addition of a 0.0047µF capacitor on the VC pin reduces
switching frequency ripple to only a few millivolts. A low
value for RC will also reduce VC pin ripple, but loop phase
margin may be inadequate.
FB
Layout Considerations
VC
For maximum efficiency, LT1371 switch rise and fall times
are made as short as possible. To prevent radiation and
high frequency resonance problems, proper layout of the
components connected to the switch node is essential. B
field (magnetic) radiation is minimized by keeping output
diode, Switch pin and output bypass capacitor leads as
short as possible. Figures 3, 4 and 5 show recommended
positions for these components. E field radiation is kept
low by minimizing the length and area of all traces connected to the Switch pin. A ground plane should always be
used under the switcher circuitry to prevent interplane
coupling.
GND S/S
NFB VSW VIN
C
CONNECT
GROUND PIN
AND TAB DIRECTLY
TO GROUND PLANE.
TAB MAY BE
SOLDERED OR
BOLTED TO
GROUND PLANE*
D
C
KEEP PATH FROM
VSW, OUTPUT DIODE,
OUTPUT CAPACITORS
AND GROUND RETURN
AS SHORT AS POSSIBLE
*SEE T7 PACKAGE LAYOUT CONSIDERATIONS FOR VERTICAL MOUNTING
OF THE T7 PACKAGE
Figure 4. Layout Considerations— T7 Package
The high speed switching current path is shown schematically in Figure 6. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, output diode and output capacitor is
the only one containing nanosecond rise and fall times.
Keep this path as short as possible.
VC
VSW
FB
NC
NFB
VSW
GND
GND
GND
GND
GND
GND
GND
GND
SHDN
D
C
KEEP PATH FROM
VSW, OUTPUT DIODE,
OUTPUT CAPACITORS
AND GROUND RETURN
AS SHORT AS POSSIBLE
C
NC
SYNC
NC
VIN
GND
LT1371 • F05
CONNECT ALL GROUND PINS TO GROUND PLANE
FB
VC
GND S/S
NFB VSW VIN
Figure 5. Layout Considerations— SW Package
C
CONNECT
GROUND PIN
AND TAB DIRECTLY
TO GROUND PLANE
C
L1
D
LT1371 • F04
KEEP PATH FROM
VSW, OUTPUT DIODE,
OUTPUT CAPACITORS
AND GROUND RETURN
AS SHORT AS POSSIBLE
LT1371 • F03
SWITCH
NODE
VOUT
VIN
HIGH
FREQUENCY
CIRCULATING
PATH
LOAD
Figure 3. Layout Considerations— R Package
LT1371 • F06
Figure 6
11
LT1371
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APPLICATIO S I FOR ATIO
VIN
T7 Package Layout Considerations
Electrical connection to the tab of a T7 package is required
for proper device operation. If the tab is tied directly to the
ground plane (Figure 4) no other considerations are necessary. If the tab is not connected directly to the ground
plane, as in a vertically mounted application, a separate
electrical connection from the tab to a “floating node” is
required. Ground returns for the VIN capacitor, VC components and output feedback resistor divider are then connected to the floating node. This is shown schematically in
Figure 7. All other system ground connections are made to
Pin 4.
The electrical connection from the T7 package tab to the
floating node must be a low resistance (< 0.1Ω), low
inductance (< 20nH) path which can be accomplished with
a jumper wire or an electrically conductive heat sink.
Bolt the jumper wire directly to the tab using a solder tail
to maintain low resistance. The jumper wire length should
not exceed 3/4 inch of 24 AWG gauge wire or larger to
minimize the inductance.
Vertically mounted electrically conductive heat sinks are
available from many heat sink manufacturers. These heat
sinks also have tabs that solder directly to the board
creating the required low resistance, low inductance path
from the tab to the floating node. The tab should be bolted
12
VOUT
7
VIN
VSW
1
LT1371T7
VC
FB
GND
TAB
FLOATING NODE
(TAB TIES INTERNALLY
TO PIN 4 GROUND)
5
2
GND
4
LT1371 • F07
SYSTEM GROUND
Figure 7. Tab Connections for Vertically Mounted T7 Package
or soldered directly to the heat sink to maintain low
resistance. Heat sinks are available in clip-on styles but are
only recommended if the tab to heat sink contact resistance can be maintained below 0.1Ω for the life of the
product.
More Help
For more detailed information on switching regulator
circuits, please see Application Note 19. Linear Technology also offers a computer software program,
SwitcherCAD, to assist in designing switching converters.
In addition, our Applications Department is always ready
to lend a helping hand.
LT1371
U
TYPICAL APPLICATIONS N
Positive-to-Negative Converter with Direct Feedback
VIN
2.7V TO 13V
+
2
D2
P6KE-15A
D3
1N4148 1 •
VIN
OFF
VSW
S/S
R2
6.19k
1%
T1*
C1
100µF
ON
4
+
•
R2
2.49k
1%
NFB
+
–VOUT†
–5V
C1
22µF
ON
1N4148
VIN
VSW
FB
OFF
MBRS360T3
T1*
2, 3
7
+
P6KE-20A •
S/S
8, 9
•4
10
•
LT1371
NFB
C2
0.047µF
R1
2k
C3
0.0047µF
VIN
2.7V TO 10V
R3
2.49k
1%
GND
R1
68.1k
1%
C4
100µF
×2
3
D1
MBRS330T3
LT1371
VC
Dual Output Flyback Converter with Overvoltage Protection
*COILTRONICS CTX10-4
†
MAX IOUT
IOUT VIN
0.6A 3V
1.0A 5V
1.5A 9V
VC
1
C3
0.0047µF
C5
47µF
–VOUT
–15V
R4
12.1k
1%
R5
2.49k
1%
C2
0.047µF
R3
2k
LT1371 • TA03
C4
47µF
+
MBRS360T3
GND
VOUT
15V
*DALE LPE-5047-100MB
Single Li-Ion Cell to 5V
2 Li-Ion Cells to 5V SEPIC Converter**
VIN
4V TO 9V
L1*
L1A*
10µH
VIN
OFF
ON
VSW
S/S
•
LT1371
+
C1
33µF
20V
FB
GND
LT1371 • TA04
R2
18.7k
1%
•
+
L1B*
10µH
R1
2k
C4
0.047µF
R3
6.19k
1%
C5
0.0047µF
R1
18.7k
1%
FB
+
LT1371
+
C3
100µF
10V
×2
S/S
SINGLE
Li-Ion
CELL
+
C1**
100µF
10V
GND
C4**
100µF
10V
×2
VC
R2
6.19k
1%
C2
0.047µF
R3
2k
C3
0.0047µF
LT1371 • TA06
LT1371 • TA05
C1 = AVX TPSD 336M020R0200
C2 = TOKIN 1E475ZY5U-C304
C3 = AVX TPSD107M010R0100
* SINGLE INDUCTOR WITH TWO WINDINGS
COILTRONICS CTX10-4
** INPUT VOLTAGE MAY BE GREATER OR
LESS THAN OUTPUT VOLTAGE
OFF
VOUT†
5V
ON
VOUT†
5V
VSW
VIN
MBRS330T3
C2
4.7µF
VC
D1
MBRS320T3
†MAX I
OUT
IOUT
0.85A
1A
1.3A
1.5A
VIN
4V
5V
7V
9V
*COILCRAFT DO3316P-103
**AVX TPSD107M010R0100
†MAX I
IOUT
1.2A
1.6A
1.8A
OUT
VIN
2.7V
3.3V
3.6V
13
LT1371
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TYPICAL APPLICATIONS N
20W CCFL Supply
47pF
LAMP
1N4148
11
L1
5
8
1
4
3
2
+
0.47µF
22µF
Q2
Q1
150Ω
INTENSITY
CONTROL
L2
15µH
MUR405
1N4148
1N4148
VIN
9V
TO
15V
22k
VSW
VIN
+
10k
FB
LT1371
140Ω
1µF
2.2µF
GND
VC
L1=COILTRONICS CTX02-11128
L2=COILCRAFT DO3316P-153
Q1, Q2=ZETEX ZTX849, ZDT1048 OR ROHM 2SC5001
0.47 µF=WIMA 3X 0.15 µF TYPE MKP-20
COILTRONICS (407) 241-7876
+
LT1371 • TA07
2.2µF
Laser Power Supply
1800pF
10kV
0.01µF
5kV
47k
5W
1800pF
10kV
8
11
L1
1
4
5
HV DIODES
3
2
0.47µF
LASER
+
2.2µF
Q2
Q1
150Ω
L2
82µH
MUR405
VIN
12V TO 25V
VSW
10k
VIN
+
10k
FB
LT1371
2.2µF
VC
GND
0.1µF
VIN
1N4002
(ALL)
190Ω
1%
+
10µF
LT1371 • TA08
14
L1 = COILTRONICS CTX02-11128
L2 = GOWANDA GA40-822K
Q1, Q2 = ZETEX ZTX849
0.47µF = WIMA 3X 0.15µF TYPE MKP-20
HV DIODES = SEMTECH-FM-50
LASER = HUGHES 3121H-P
COILTRONICS (407) 241-7876
LT1371
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
R Package
7-Lead Plastic DD Pak
(LTC DWG # 05-08-1462)
0.256
(6.502)
0.060
(1.524)
TYP
0.060
(1.524)
0.390 – 0.415
(9.906 – 10.541)
0.165 – 0.180
(4.191 – 4.572)
0.045 – 0.055
(1.143 – 1.397)
15° TYP
0.060
(1.524)
0.183
(4.648)
0.059
(1.499)
TYP
0.330 – 0.370
(8.382 – 9.398)
(
+0.008
0.004 –0.004
+0.203
0.102 –0.102
)
0.095 – 0.115
(2.413 – 2.921)
0.075
(1.905)
0.300
(7.620)
(
+0.012
0.143 –0.020
+0.305
3.632 –0.508
BOTTOM VIEW OF DD PAK
HATCHED AREA IS SOLDER PLATED
COPPER HEAK SINK
0.040 – 0.060
(1.016 – 1.524)
0.026 – 0.036
(0.660 – 0.914)
)
0.050 ± 0.012
(1.270 ± 0.305)
0.013 – 0.023
(0.330 – 0.584)
R (DD7) 0695
SW Package
20-Lead Plastic Small Outline (Wide 0.300)
(LTC DWG # 05-08-1620)
0.496 – 0.512*
(12.598 – 13.005)
20
19
18
17
16
15
14
13
12
11
0.394 – 0.419
(10.007 – 10.643)
NOTE 1
0.291 – 0.299**
(7.391 – 7.595)
0.010 – 0.029 × 45°
(0.254 – 0.737)
1
2
3
4
5
6
7
8
9
0.093 – 0.104
(2.362 – 2.642)
10
0.037 – 0.045
(0.940 – 1.143)
0° – 8° TYP
0.009 – 0.013
(0.229 – 0.330)
NOTE 1
0.016 – 0.050
(0.406 – 1.270)
0.050
(1.270)
TYP
0.014 – 0.019
(0.356 – 0.482)
TYP
0.004 – 0.012
(0.102 – 0.305)
S20 (WIDE) 0695
NOTE:
1. PIN 1 IDENT, NOTCH ON TOP AND CAVITIES ON THE BOTTOM OF PACKAGES ARE THE MANUFACTURING OPTIONS.
THE PART MAY BE SUPPLIED WITH OR WITHOUT ANY OF THE OPTIONS
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LT1371
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
T7 Package
7-Lead Plastic TO-220 (Standard)
(LTC DWG # 05-08-1422)
0.165 – 0.180
(4.293 – 4.572)
0.147 – 0.155
(3.734 – 3.937)
DIA
0.390 – 0.415
(9.906 – 10.541)
0.045 – 0.055
(1.143 – 1.397)
0.230 – 0.270
(5.842 – 6.858)
0.570 – 0.620
(14.478 – 15.748)
0.460 – 0.500
(11.684 – 12.700)
0.330 – 0.370
(8.382 – 9.398)
0.620
(15.75)
TYP
0.700 – 0.728
(17.780 – 18.491)
0.152 – 0.202
0.260 – 0.320 (3.860 – 5.130)
(6.604 – 8.128)
0.040 – 0.060
(1.016 – 1.524)
0.095 – 0.115
(2.413 – 2.921)
0.013 – 0.023
(0.330 – 0.584)
0.026 – 0.036
(0.660 – 0.914)
0.135 – 0.165
(3.429 – 4.191)
0.155 – 0.195
(3.937 – 4.953)
T7 (TO-220) (FORMED) 0695
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1171
100kHz 2.5A Boost Switching Regulator
Good for Up to VIN = 40V
LTC 1265
12V 1.2A Monolithic Buck Converter
Converts 5V to 3.3V at 1A with 90% Efficiency
LT1302
Micropower 2A Boost Converter
Converts 2V to 5V at 600mA in SO-8 Packages
LT1372
500kHz 1.5A Boost Switching Regulator
Also Regulates Negative Flyback Outputs
LT1373
Low Supply Current 250kHz 1.5A Boost Switching Regulator
90% Efficient Boost Converter with Constant Frequency
LT1376
500kHz 1.5A Buck Switching Regulator
Steps Down from Up to 25V Using 4.7µH Inductors
LT1512
500kHz 1.5A SEPIC Battery Charger
Input Voltage May Be Greater or Less Than Battery Voltage
LT1513
500kHz 3A SEPIC Battery Charger
Input Voltage May Be Greater or Less Than Battery Voltage
®
16
Linear Technology Corporation
LT/GP 0996 5K REV A • PRINTED IN THE USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977
LINEAR TECHNOLOGY CORPORATION 1995
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