LT1372/LT1377
500kHz and 1MHz
High Efficiency
1.5A Switching Regulators
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FEATURES
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DESCRIPTIO
Faster Switching with Increased Efficiency
Uses Small Inductors: 4.7µH
All Surface Mount Components
Only 0.5 Square Inch of Board Space
Low Minimum Supply Voltage: 2.7V
Quiescent Current: 4mA Typ
Current Limited Power Switch: 1.5A
Regulates Positive or Negative Outputs
Shutdown Supply Current: 12µA Typ
Easy External Synchronization
8-Pin SO or PDIP Packages
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APPLICATIO S
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The LT ®1372/LT1377 are monolithic high frequency
switching regulators. They can be operated in all standard
switching configurations including boost, buck, flyback,
forward, inverting and “Cuk.” A 1.5A high efficiency switch
is included on the die, along with all oscillator, control and
protection circuitry. All functions of the LT1372/LT1377
are integrated into 8-pin SO/PDIP packages.
The LT1372/LT1377 typically consumes only 4mA quiescent current and has higher efficiency than previous parts.
High frequency switching allows for very small inductors
to be used. All surface mount components consume less
than 0.5 square inch of board space.
New design techniques increase flexibility and maintain
ease of use. Switching is easily synchronized to an external logic level source. A logic low on the shutdown pin
reduces supply current to 12µA. Unique error amplifier
circuitry can regulate positive or negative output voltage
while maintaining simple frequency compensation techniques. Nonlinear error amplifier transconductance reduces output overshoot on start-up or overload recovery.
Oscillator frequency shifting protects external components during overload conditions.
Boost Regulators
CCFL Backlight Driver
Laptop Computer Supplies
Multiple Output Flyback Supplies
Inverting Supplies
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATIO
12V Output Efficiency
5V-to-12V Boost Converter
OFF
5
VIN
ON 4 S/S
VSW
LT1372/LT1377
+
C1**
22µF
FB
GND
6, 7
C2
0.047µF
R3
2k
100
D1
MBRS120T3
L1*
4.7µH
VOUT†
12V
R1
53.6k
1%
8
2
+
VC
R2
6.19k
1%
1
C3
0.0047µF
C4**
22µF
VIN = 5V
90
*FOR LT1372 USE 10µH
COILCRAFT DO1608-472 (4.7µH) OR
COILCRAFT DT3316-103 (10µH) OR
SUMIDA CD43-4R7 (4.7µH) OR
SUMIDA CD73-100KC (10µH) OR
**AVX TPSD226M025R0200
†
MAX IOUT
L1 IOUT (LT1377) IOUT (LT1372)
0.25A
NA
4.7µH
0.35A
0.29A
10µH
LT1372 • TA01
EFFICIENCY (%)
5V
80
70
60
50
0.01
0.1
OUTPUT CURRENT (A)
1
LT1372 • TA02
1
LT1372/LT1377
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ABSOLUTE
RATI GS
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PACKAGE/ORDER I FOR ATIO
(Note 1)
Supply Voltage ....................................................... 30V
Switch Voltage
LT1372/LT1377 .................................................. 35V
LT1372HV .......................................................... 42V
S/S Pin Voltage ....................................................... 30V
Feedback Pin Voltage (Transient, 10ms) .............. ±10V
Feedback Pin Current ........................................... 10mA
Negative Feedback Pin Voltage
(Transient, 10ms) ............................................. ±10V
Operating Junction Temperature Range
Commercial ........................................ 0°C to 125°C*
Industrial ......................................... – 40°C to 125°C
Short Circuit ......................................... 0°C to 150°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
ORDER PART NUMBER
TOP VIEW
VC 1
8
VSW
FB 2
7
GND
NFB 3
6
GND S
S/S 4
5
VIN
N8 PACKAGE
8-LEAD PDIP
S8 PACKAGE
8-LEAD PLASTIC SO
LT1372CN8
LT1372HVCN8
LT1372CS8
LT1372HVCS8
LT1372IN8
LT1372HVIN8
LT1372IS8
LT1372HVIS8
LT1377CS8
LT1377IS8
S8 PART MARKING
TJMAX = 125°C, θJA = 100°C/ W (N8)
TJMAX = 125°C, θJA = 120°C/ W (S8)
1372H
1372HI
1372
1372I
1377
1377I
Consult factory for parts specified with wider operating temperature ranges.
*Units shipped prior to Date Code 9552 are rated at 100°C maximum
operating temperature.
ELECTRICAL CHARACTERISTICS
The ● denotes specifcatons which appy over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VREF
Reference Voltage
Measured at Feedback Pin
VC = 0.8V
IFB
Feedback Input Current
●
MIN
TYP
MAX
UNITS
1.230
1.225
1.245
1.245
1.260
1.265
V
V
250
550
900
nA
nA
0.01
0.03
%/V
– 2.490
– 2.490
– 2.440
– 2.410
VFB = VREF
●
VNFB
INFB
gm
AV
f
2
Reference Voltage Line Regulation
2.7V ≤ VIN ≤ 25V, VC = 0.8V
●
Negative Feedback Reference Voltage
Measured at Negative Feedback Pin
Feedback Pin Open, VC = 0.8V
●
– 2.540
– 2.570
Negative Feedback Input Current
VNFB = VNFR
●
– 45
Negative Feedback Reference Voltage
Line Regulation
2.7V ≤ VIN ≤ 25V, VC = 0.8V
●
Error Amplifier Transconductance
∆IC = ±25µA
V
V
– 30
– 15
µA
0.01
0.05
%/V
1100
700
1500
●
1900
2300
µmho
µmho
120
200
350
µA
1400
2400
µA
1.95
0.40
2.30
0.52
V
V
Error Amplifier Source Current
VFB = VREF – 150mV, VC = 1.5V
●
Error Amplifier Sink Current
VFB = VREF + 150mV, VC = 1.5V
●
Error Amplifier Clamp Voltage
High Clamp, VFB = 1V
Low Clamp, VFB = 1.5V
1.70
0.25
VC Pin Threshold
Duty Cycle = 0%
0.8
1
1.25
V
Switching Frequency
2.7V ≤ VIN ≤ 25V
LT1372
0°C ≤ TJ ≤ 125°C
– 40°C ≤ TJ < 0°C (I Grade)
LT1377
0°C ≤ TJ ≤ 125°C
– 40°C ≤ TJ < 0°C (I Grade)
450
430
400
0.90
0.86
0.80
500
500
550
580
580
1.10
1.16
1.16
kHz
kHz
kHz
MHz
MHz
MHz
Error Amplifier Voltage Gain
500
●
●
1
1
V/ V
LT1372/LT1377
ELECTRICAL CHARACTERISTICS
The ● denotes specifcatons which appy over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
Maximum Switch Duty Cycle
TYP
85
●
130
Output Switch Breakdown Voltage
260
ns
%
LT1372/LT1377
LT1372HV
0°C ≤ TJ ≤ 125°C
– 40°C ≤ TJ < 0°C (I Grade)
●
35
47
V
●
42
40
47
V
V
0.5
0.8
Ω
1.9
1.7
2.7
2.5
A
A
Supply Current Increase During Switch On-Time
15
25
mA/A
Control Voltage to Switch Current
Transconductance
2
VSAT
Output Switch “On” Resistance
ISW = 1A
●
ILIM
Switch Current Limit
Duty Cycle = 50%
Duty Cycle = 80% (Note 2)
●
●
∆IIN
∆ISW
Minimum Input Voltage
IQ
UNITS
95
Switch Current Limit Blanking Time
BV
MAX
1.5
1.3
A/V
●
2.4
2.7
V
Supply Current
2.7V ≤ VIN ≤ 25V
●
4
5.5
mA
Shutdown Supply Current
2.7V ≤ VIN ≤ 25V, VS/S ≤ 0.6V
0°C ≤ TJ ≤ 125°C
– 40°C ≤ TJ < 0°C (I Grade)
●
12
30
50
µA
µA
2.7V ≤ VIN ≤ 25V
●
0.6
1.3
2
V
●
5
12
25
µs
Shutdown Threshold
Shutdown Delay
S/S Pin Input Current
0V ≤ VS/S ≤ 5V
●
– 10
15
µA
Synchronization Frequency Range
LT1372
LT1377
●
●
600
1.2
800
1.6
kHz
MHz
Note 1: Absolute Maximum Ratings are those values beyond which the life
of the device may be impaired.
Note 2: For duty cycles (DC) between 50% and 90%, minimum
guaranteed switch current is given by ILIM = 0.667 (2.75 – DC).
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TYPICAL PERFORMANCE CHARACTERISTICS
Switch Saturation Voltage
vs Switch Current
Switch Current Limit
vs Duty Cycle
3.0
0.9
25°C
SWITCH CURRENT LIMIT (A)
150°C
100°C
0.8
0.7
0.6
0.5
–55°C
0.4
0.3
0.2
3.0
2.5
2.8
25°C AND
125°C
2.0
–55°C
1.5
1.0
0.5
INPUT VOLTAGE (V)
1.0
SWITCH SATURATION VOLTAGE (V)
Minimum Input Voltage
vs Temperature
2.6
2.4
2.2
2.0
0.1
0
0
0
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
SWITCH CURRENT (A)
LT1372 • G01
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
LT1372 • G02
1.8
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1372 • G03
3
LT1372/LT1377
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TYPICAL PERFOR A CE CHARACTERISTICS
2.0
18
1.8
SHUTDOWN DELAY (µs)
14
1.4
12
1.2
10
1.0
SHUTDOWN DELAY
8
0.8
6
0.6
4
0.4
2
0.2
0
–50 –25
0
SHUTDOWN THRESHOLD (V)
1.6
SHUTDOWN THRESHOLD
0
25 50 75 100 125 150
TEMPERATURE (°C)
3.0
2.5
400
fSYNC = 700kHz (LT1372)
fSYNC = 1.4MHz (LT1377)
2.0
LT1377
1.5
LT1372
1.0
0.5
0
–50 –25
2
1
0
–1
–2
–3
–4
–5
–1
0
1
2 3 4 5 6
S/S PIN VOLTAGE (V)
7
8
90
80
70
60
50
40
30
VC THRESHOLD
0.8
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1372 • G10
∆I (VC)
∆V (FB)
1400
1200
1000
800
600
400
200
20
0
0
–50 –25
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
FEEDBACK PIN VOLTAGE (V)
0
LT1372 • G09
0
VFB =VREF
700
600
500
400
300
200
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
Negative Feedback Input Current
vs Temperature
100
0.6
gm =
1600
NEGATIVE FEEDBACK INPUT CURRENT (µA)
1.4
4
Error Amplifier Transconductance
vs Temperature
Feedback Input Current
vs Temperature
FEEDBACK INPUT CURRENT (nA)
VC PIN VOLTAGE (V)
1.8
1.6
0.1
LT1372 • G06
2000
800
VC HIGH CLAMP
VREF
–0.2
–0.1
FEEDBACK PIN VOLTAGE (V)
LT1372 • G08
2.4
0.4
–50 –25
–0.3
1800
10
9
2.2
1.0
–200
100
VC Pin Threshold and High
Clamp Voltage vs Temperature
1.2
–100
110
LT1372 • G07
2.0
0
25 50 75 100 125 150
TEMPERATURE (°C)
TRANSCONDUCTANCE (µmho)
SWITCHING FREQUENCY (% OF TYPICAL)
S/S PIN INPUT CURRENT (µA)
3
125°C
100
Switching Frequency
vs Feedback Pin Voltage
VIN = 5V
25°C
–55°C
200
LT1372 • G05
S/S Pin Input Current
vs Voltage
4
300
–300
0
LT1372 • G04
5
ERROR AMPLIFIER OUTPUT CURRENT (µA)
20
16
Error Amplifier Output Current
vs Feedback Pin Voltage
Minimum Synchronization
Voltage vs Temperature
MINIMUM SYNCHRONIZATION VOLTAGE (VP-P)
Shutdown Delay and Threshold
vs Temperature
25 50 75 100 125 150
TEMPERATURE (°C)
LT1372 • G11
VNFB =VNFR
–10
–20
–30
–40
–50
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1372 • G12
LT1372/LT1377
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PI FU CTIO S
VIN (Pin 5): Bypass input supply pin with 10µF or more. The
part goes into undervoltage lockout when VIN drops below
2.5V. Undervoltage lockout stops switching and pulls the
VC pin low.
VC (Pin 1): The compensation pin is used for frequency
compensation, current limiting and soft start. It is the
output of the error amplifier and the input of the current
comparator. Loop frequency compensation can be performed with an RC network connected from the VC pin to
ground.
GND S (Pin 6): The ground sense pin is a “clean” ground.
The internal reference, error amplifier and negative feedback amplifier are referred to the ground sense pin. Connect it to ground. Keep the ground path connection to the
output resistor divider and the VC compensation network
free of large ground currents.
FB (Pin 2): The feedback pin is used for positive output
voltage sensing and oscillator frequency shifting. It is the
inverting input to the error amplifier. The noninverting
input of this amplifier is internally tied to a 1.245V
reference. Load on the FB pin should not exceed 250µA
when the NFB pin is used. See Applications Information.
GND (Pin 7): The ground pin is the emitter connection of
the power switch and has large currents flowing through it.
It should be connected directly to a good quality ground
plane.
NFB (Pin 3): The negative feedback pin is used for negative
output voltage sensing. It is connected to the inverting
input of the negative feedback amplifier through a 100k
source resistor.
VSW (Pin 8): The switch pin is the collector of the power
switch and has large currents flowing through it. Keep the
traces to the switching components as short as possible to
minimize radiation and voltage spikes.
S/S (Pin 4): Shutdown and Synchronization Pin. The S/S
pin is logic level compatible. Shutdown is active low and
the shutdown threshold is typically 1.3V. For normal
operation, pull the S/S pin high, tie it to VIN or leave it
floating. To synchronize switching, drive the S/S pin between 600kHz and 800kHz (LT1372) or 1.2MHz to 1.6MHz
(LT1377).
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BLOCK DIAGRA
VIN
SHUTDOWN
DELAY AND RESET
S/S
SYNC
SW
LOW DROPOUT
2.3V REG
ANTI-SAT
LOGIC
OSC
DRIVER
SWITCH
5:1 FREQUENCY
SHIFT
+
100k
NFB
NFBA
–
COMP
50k
–
FB
+
1.245V
REF
GND SENSE
+
EA
IA
VC
AV ≈ 6
0.08Ω
–
GND
LT1372 • BD
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LT1372/LT1377
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OPERATIO
The LT1372/LT1377 are current mode switchers. This
means that switch duty cycle is directly controlled by
switch current rather than by output voltage. Referring to
the block diagram, the switch is turned “On” at the start of
each oscillator cycle. It is turned “Off” when switch current
reaches a predetermined level. Control of output voltage is
obtained by using the output of a voltage sensing error
amplifier to set current trip level. This technique has
several advantages. First, it has immediate response to
input voltage variations, unlike voltage mode switchers
which have notoriously poor line transient response.
Second, it reduces the 90° phase shift at mid-frequencies
in the energy storage inductor. This greatly simplifies
closed-loop frequency compensation under widely varying input voltage or output load conditions. Finally, it
allows simple pulse-by-pulse current limiting to provide
maximum switch protection under output overload or
short conditions. A low dropout internal regulator provides a 2.3V supply for all internal circuitry. This low
dropout design allows input voltage to vary from 2.7V to
25V with virtually no change in device performance. A
500kHz (LT1372) or 1MHz (LT1377) oscillator is the basic
clock for all internal timing. It turns “On” the output switch
via the logic and driver circuitry. Special adaptive anti-sat
circuitry detects onset of saturation in the power switch
and adjusts driver current instantaneously to limit switch
saturation. This minimizes driver dissipation and provides
very rapid turn-off of the switch.
A 1.245V bandgap reference biases the positive input of
the error amplifier. The negative input of the amplifier is
brought out for positive output voltage sensing. The error
amplifier has nonlinear transconductance to reduce out-
put overshoot on start-up or overload recovery. When
the feedback voltage exceeds the reference by 40mV,
error amplifier transconductance increases ten times,
which reduces output overshoot. The feedback input also
invokes oscillator frequency shifting, which helps protect components during overload conditions. When the
feedback voltage drops below 0.6V, the oscillator frequency is reduced 5:1. Lower switching frequency allows
full control of switch current limit by reducing minimum
switch duty cycle.
Unique error amplifier circuitry allows the LT1372/LT1377
to directly regulate negative output voltages. The negative
feedback amplifier’s 100k source resistor is brought out
for negative output voltage sensing. The NFB pin regulates
at – 2.49V while the amplifier output internally drives the
FB pin to 1.245V. This architecture, which uses the same
main error amplifier, prevents duplicating functions and
maintains ease of use. Consult Linear Technology Marketing for units that can regulate down to – 1.25V.
The error signal developed at the amplifier output is
brought out externally. This pin (VC) has three different
functions. It is used for frequency compensation, current
limit adjustment and soft starting. During normal regulator operation this pin sits at a voltage between 1V (low
output current) and 1.9V (high output current). The error
amplifier is a current output (gm) type, so this voltage can
be externally clamped for lowering current limit. Likewise, a capacitor coupled external clamp will provide soft
start. Switch duty cycle goes to zero if the VC pin is pulled
below the control pin threshold, placing the LT1372/
LT1377 in an idle mode.
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APPLICATIO S I FOR ATIO
Positive Output Voltage Setting
The LT1372/LT1377 develops a 1.245V reference (VREF)
from the FB pin to ground. Output voltage is set by
connecting the FB pin to an output resistor divider
(Figure 1). The FB pin bias current represents a small
error and can usually be ignored for values of R2 up to 7k.
The suggested value for R2 is 6.19k. The NFB pin is
normally left open for positive output applications.
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VOUT
R1
FB
PIN
R2
( )
( )
VOUT = VREF 1 + R1
R2
R1 = R2
VOUT
–1
1.245
VREF
LT1372 • F01
Figure 1. Positive Output Resistor Divider
LT1372/LT1377
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APPLICATIO S I FOR ATIO
Positive fixed voltage versions are available (consult
Linear Technology marketing).
Negative Output Voltage Setting
The LT1372/LT1377 develops a – 2.49V reference (VNFR)
from the NFB pin to ground. Output voltage is set by
connecting the NFB pin to an output resistor divider
(Figure 2). The – 30µA NFB pin bias current (INFB) can
cause output voltage errors and should not be ignored.
This has been accounted for in the formula in Figure 2. The
suggested value for R2 is 2.49k. The FB pin is normally left
open for negative output application. See Dual Polarity
Output Voltage Sensing for limitatins on FB pin loading
when using the NFB pin.
–VOUT
INFB
( )
R1
–VOUT = VNFB 1 + R1 + INFB (R1)
R2
R2
R1 =
NFB
PIN
VNFR
VOUT– 2.49
( )(
2.49 + 30 × 10–6
R2
)
LT1372 • F02
Figure 2. Negative Output Resistor Divider
Dual Polarity Output Voltage Sensing
Certain applications benefit from sensing both positive
and negative output voltages. One example is the “Dual
Output Flyback Converter with Overvoltage Protection”
circuit shown in the Typical Applications section. Each
output voltage resistor divider is individually set as described above. When both the FB and NFB pins are used,
the LT1372/LT1377 acts to prevent either output from
going beyond its set output voltage. For example in this
application, if the positive output were more heavily loaded
than the negative, the negative output would be greater
and would regulate at the desired set-point voltage. The
positive output would sag slightly below its set-point
voltage. This technique prevents either output from going
unregulated high at no load. Please note that the load on
the FB pin should not exceed 250µA when the NFB pin is
used. This situation occurs when the resistor dividers are
used at both FB and NFB. True load on FB is not the full
divider current unless the positive output is shorted to
ground. See Dual Output Flyback Converter application.
Shutdown and Synchronization
The dual function S/S pin provides easy shutdown and
synchronization. It is logic level compatible and can be
pulled high, tied to VIN or left floating for normal operation.
A logic low on the S/S pin activates shutdown, reducing
the part’s supply current to 12µA. Typical synchronization
range is from 1.05 to 1.8 times the part’s natural switching
frequency, but is only guaranteed between 600kHz and
800kHz (LT1372) or 1.2MHz and 1.6MHz (LT1377). At
start-up, the synchronization signal should not be applied
until the feedback pin is above the frequency shift voltage
of 0.7V. If the NFB pin is used, synchronization should not
be applied until the NFB pin is more negative than – 1.4V.
A 12µs resetable shutdown delay network guarantees the
part will not go into shutdown while receiving a synchronization signal.
Caution should be used when synchronizing above 700kHz
(LT1372) or 1.4MHz (LT1377) because at higher sync
frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced.
This type of subharmonic switching only occurs when the
duty cycle of the switch is above 50%. Higher inductor
values will tend to eliminate problems.
Thermal Considerations
Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause excessive die temperatures. The packages are rated at 120°C/W
for SO (S8) and 130°C/W for PDIP (N8).
Average supply current (including driver current) is:
IIN = 4mA + DC (ISW/60 + ISW × 0.004)
ISW = switch current
DC = switch duty cycle
Switch power dissipation is given by:
PSW = (ISW)2 × RSW × DC
RSW = output switch “On” resistance
Total power dissipation of the die is the sum of supply
current times supply voltage plus switch power:
PD(TOTAL) = (IIN × VIN) + PSW
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LT1372/LT1377
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APPLICATIO S I FOR ATIO
Choosing the Inductor
For most applications the inductor will fall in the range of
2.2µH to 22µH. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output
current because they reduce peak current seen by the
power switch, which has a 1.5A limit. Higher values also
reduce input ripple voltage and reduce core loss.
When choosing an inductor you might have to consider
maximum load current, core and copper losses, allowable
component height, output voltage ripple, EMI, fault
current in the inductor, saturation, and of course, cost.
The following procedure is suggested as a way of handling
these somewhat complicated and conflicting requirements.
1. Assume that the average inductor current for a boost
converter is equal to load current times VOUT / VIN and
decide whether or not the inductor must withstand
continuous overload conditions. If average inductor
current at maximum load current is 0.5A, for instance,
a 0.5A inductor may not survive a continuous 1.5A
overload condition. Also be aware that boost converters are not short circuit protected, and that under
output short conditions, inductor current is limited only
by the available current of the input supply.
2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t
omit this step. Powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core materials fall in between
somewhere. The following formula assumes continuous mode operation but it errors only slightly on the
high side for discontinuous mode, so it can be used for
all conditions.
V (V
–V )
V
IPEAK = IOUT × OUT + IN OUT IN
VIN
2(f)(L)(VOUT)
VIN = Minimum Input Voltage
f = 500kHz Switching Frequency (LT1372) or
1MHz Switching Frequency (LT1377)
3. Decide if the design can tolerate an “open” core geometry like a rod or barrel, which have high magnetic field
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radiation, or whether it needs a closed core like a toroid
to prevent EMI problems. One would not want an open
core next to a magnetic storage media for instance!
This is a tough decision because the rods or barrels are
temptingly cheap and small, and there are no helpful
guidelines to calculate when the magnetic field radiation will be a problem.
4. Start shopping for an inductor which meets the requirements of core shape, peak current (to avoid
saturation), average current (to limit heating) and fault
current. If the inductor gets too hot, wire insulation will
melt and cause turn-to-turn shorts. Keep in mind that
all good things like high efficiency, low profile and high
temperature operation will increase cost, sometimes
dramatically.
5. After making an initial choice, consider the secondary
things like output voltage ripple, second sourcing, etc.
Use the experts in the Linear Technology application
department if you feel uncertain about the final choice.
They have experience with a wide range of inductor
types and can tell you about the latest developments in
low profile, surface mounting, etc.
Output Capacitor
The output capacitor is normally chosen by its effective
series resistance, (ESR), because this is what determines
output ripple voltage. At 500kHz, any polarized capacitor
is essentially resistive. To get low ESR takes volume, so
physically smaller capacitors have high ESR. The ESR
range for typical LT1372 and LT1377 applications is
0.05Ω to 0.5Ω. A typical output capacitor is an AVX type
TPS, 22µF at 25V, with a guaranteed ESR less than 0.2Ω.
This is a “D” size surface mount solid tantalum capacitor.
TPS capacitors are specially constructed and tested for
low ESR, so they give the lowest ESR for a given volume.
To further reduce ESR, multiple output capacitors can be
used in parallel. The value in microfarads is not particularly critical, and values from 22µF to greater than 500µF
work well, but you cannot cheat mother nature on ESR.
If you find a tiny 22µF solid tantalum capacitor, it will have
high ESR, and output ripple voltage will be terrible. Table
1 shows some typical solid tantalum surface mount
capacitors.
LT1372/LT1377
U
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APPLICATIO S I FOR ATIO
Table 1. Surface Mount Solid Tantalum Capacitor
ESR and Ripple Current
E CASE SIZE
IRIPPLE =
ESR (MAX Ω)
RIPPLE CURRENT (A)
0.1 to 0.3
0.7 to 0.9
0.7 to 1.1
0.4
0.1 to 0.3
0.9 to 2.0
0.7 to 1.1
0.36 to 0.24
0.2 (Typ)
1.8 to 3.0
0.5 (Typ)
0.22 to 0.17
2.5 to 10
0.16 to 0.08
AVX TPS, Sprague 593D
AVX TAJ
0.3(VIN)(VOUT – VIN)
(f)(L)(VOUT)
f = 500kHz Switching frequency (LT1372) or,
1MHz Switching frequency (LT1377)
D CASE SIZE
AVX TPS, Sprague 593D
AVX TAJ
C CASE SIZE
AVX TPS
AVX TAJ
B CASE SIZE
AVX TAJ
Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true and type TPS capacitors are
specially tested for surge capability, but surge ruggedness
is not a critical issue with the output capacitor. Solid
tantalum capacitors fail during very high turn-on surges,
which do not occur at the output of regulators. High
discharge surges, such as when the regulator output is
dead shorted, do not harm the capacitors.
Single inductor boost regulators have large RMS ripple
current in the output capacitor, which must be rated to
handle the current. The formula to calculate this is:
Output Capacitor Ripple Current (RMS)
DC
IRIPPLE (RMS) = IOUT 1 – DC
= IOUT
VOUT – VIN
VIN
Input Capacitors
The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular and
does not contain large squarewave currents as is found in
the output capacitor. Capacitors in the range of 10µF to
100µF with an ESR of 0.3Ω or less work well up to full 1.5A
switch current. Higher ESR capacitors may be acceptable
at low switch currents. Input capacitor ripple current for
boost converter is :
The input capacitor can see a very high surge current when
a battery or high capacitance source is connected “live”
and solid tantalum capacitors can fail under this condition.
Several manufacturers have developed a line of solid
tantalum capacitors specially tested for surge capability
(AVX TPS series, for instance), but even these units may
fail if the input voltage approaches the maximum voltage
rating of the capacitor. AVX recommends derating capacitor voltage by 2:1 for high surge applications. Ceramic and
aluminum electrolytic capacitors may also be used and
have a high tolerance to turn-on surges.
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generates a loop “zero” at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually
resonate with their ESL before ESR becomes effective.
They are appropriate for input bypassing because of their
high ripple current ratings and tolerance of turn-on surges.
Linear Technology plans to issue a Design Note on the use
of ceramic capacitors in the near future.
Output Diode
The suggested output diode (D1) is a 1N5818 Schottky or
its Motorola equivalent, MBR130. It is rated at 1A average
forward current and 30V reverse voltage. Typical forward
voltage is 0.42V at 1A. The diode conducts current only
during switch off time. Peak reverse voltage for boost
converters is equal to regulator output voltage. Average
forward current in normal operation is equal to output
current.
9
LT1372/LT1377
U
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APPLICATIO S I FOR ATIO
Frequency Compensation
Loop frequency compensation is performed on the output
of the error amplifier (VC pin) with a series RC network.
The main pole is formed by the series capacitor and the
output impedance (≈500kΩ) of the error amplifier. The
pole falls in the range of 2Hz to 20Hz. The series resistor
creates a “zero” at 1kHz to 5kHz, which improves loop
stability and transient response. A second capacitor,
typically one-tenth the size of the main compensation
capacitor, is sometimes used to reduce the switching
frequency ripple on the VC pin. VC pin ripple is caused by
output voltage ripple attenuated by the output divider and
multiplied by the error amplifier. Without the second
capacitor, VC pin ripple is:
VC Pin Ripple =
1.245(VRIPPLE)(gm)(RC)
(VOUT)
VRIPPLE = Output ripple (VP–P)
gm = Error amplifier transconductance
(≈1500µmho)
RC = Series resistor on VC pin
VOUT = DC output voltage
To prevent irregular switching, VC pin ripple should be
kept below 50mVP–P. Worst-case VC pin ripple occurs at
maximum output load current and will also be increased
if poor quality (high ESR) output capacitors are used. The
addition of a 0.0047µF capacitor on the VC pin reduces
switching frequency ripple to only a few millivolts. A low
value for RC will also reduce VC pin ripple, but loop phase
margin may be inadequate.
Switch Node Considerations
For maximum efficiency, switch rise and fall time are
made as short as possible. To prevent radiation and high
frequency resonance problems, proper layout of the components connected to the switch node is essential. B field
10
(magnetic) radiation is minimized by keeping output diode, switch pin, and output bypass capacitor leads as
short as possible. E field radiation is kept low by minimizing the length and area of all traces connected to the switch
pin. A ground plane should always be used under the
switcher circuitry to prevent interplane coupling.
The high speed switching current path is shown schematically in Figure 3. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, output diode, and output capacitor is
the only one containing nanosecond rise and fall times.
Keep this path as short as possible.
L1
SWITCH
NODE
VOUT
VIN
HIGH
FREQUENCY
CIRCULATING
PATH
LOAD
LT1372 • F03
Figure 3
More Help
For more detailed information on switching regulator
circuits, please see Application Note 19. Linear Technology also offers a computer software program, SwitcherCAD,
to assist in designing switching converters. In addition,
our applications department is always ready to lend a
helping hand.
LT1372/LT1377
U
TYPICAL APPLICATIONS N
Positive-to-Negative Converter with Direct Feedback
VIN
2.7V TO 16V
+
C1
22µF
OFF
VIN
VSW
LT1372/LT1377
NFB
VC
2
D2
P6KE-15A
D3
1N4148 1 •
8
4
+
•
R2
2.49k
1%
D1
MBRS130LT3
3
VIN
2.7V TO 13V
–VOUT†
–5V
R3
2.49k
1%
6, 7
R1
13k
1%
C4
47µF
3
GND
1
+
OFF
C1
22µF
2
*COILTRONICS CTX10-2 (407) 241-7876
†
MAX IOUT
IOUT VIN
0.3A 3V
0.5A 5V
0.75A 9V
MBRS140T3
T1*
2, 3
5
+
P6KE-20A •
5
LT1372/LT1377
VC
•4
8
1N4148
VIN
8
VSW
FB
ON 4 S/S
6, 7
•
3
NFB
C2
0.047µF
R1
2k
C3
0.0047µF
R2
1.21k
1%
T1*
5
ON 4 S/S
Dual Output Flyback Converter with Overvoltage Protection
1
MBRS140T3
GND
1
+
6, 7
C3
0.0047µF
C4
47µF
C5
47µF
–VOUT
–15V
R4
12.1k
1%
R5
2.49k
1%
C2
0.047µF
R3
2k
LT1372 • TA03
VOUT
15V
*DALE LPE-4841-100MB (605) 665-9301
Low Ripple 5V to – 3V “Cuk”† Converter
2
VOUT
–3V
250mA
3
1•
5
C1
22µF
10V
90% Efficient CCFL Supply
L1*
VIN
5V
+
4
7
6
LT1372/LT1377
VSW
VIN
NFB
GND S
VC
4
3
Q2
Q1
1
C4
0.047µF
1
C1
0.1µF
C6
0.1µF
3
2
+
10µF
+
*SUMIDA CLS62-100L
**MOTOROLA MBR0520LT3
†
PATENTS MAY APPLY
5
+
R4
2k
D1
1N4148
10
8
D1**
C5
0.0047µF
C2
27pF
VIN
4.5V
TO 30V
R1
1k
1%
S/S
GND
5mA MAX
LAMP
T1
•4
C2
47µF
16V
LT1372 • TA04
330Ω
C3
47µF
16V
2.7V TO
5.5V
R2
4.99k
1%
L1
33µH
+
2.2µF
OFF
ON
4
1N5818
5
VIN
S/S
VSW
562Ω*
8
LT1372/LT1377
VFB
6, 7
2
VC
+
10k
20k
DIMMING
LT1372 • TA05
GND
D2
1N4148
22k
0.1µF
1
1N4148
2µF
OPTIONAL REMOTE
DIMMING
C1 = WIMA MKP-20
L1 = COILCRAFT DT3316-333
Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001
T1 = COILTRONICS CTX 110609
* = 1% FILM RESISTOR
LT1372 • TA06
CCFL BACKLIGHT APPLICATION CIRCUITS
CONTAINED IN THIS DATA SHEET ARE
COVERED BY U.S. PATENT NUMBER 5408162
AND OTHER PATENTS PENDING
DO NOT SUBSTITUTE COMPONENTS
COILTRONICS (407) 241-7876
COILCRAFT (708) 639-6400
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of circuits as described herein will not infringe on existing patent rights.
11
LT1372/LT1377
U
TYPICAL APPLICATIONS N
2 Li-Ion Cell to 5V SEPIC Converter
VIN
4V TO 9V
OFF
+
ON 4 S/S
C1
33µF
20V
L1A*
10µH
5
VIN
8
VSW
•
VC
+
L1B*
10µH
1
6, 7
VOUT†
5V
R2
18.7k
1%
C2
1µF
LT1372/LT1377
FB 2
GND
MBRS130LT3
•
R1
2k
C5
0.0047µF
C4
0.047µF
U
PACKAGE DESCRIPTION
C3
100µF
10V
C1 = AVX TPSD 336M020R0200
C2 = TOKIN 1E105ZY5U-C103-F
C3 = AVX TPSD107M010R0100
*SINGLE INDUCTOR WITH TWO WINDINGS
COILTRONICS CTX10-1
†
MAX IOUT
IOUT
0.45A
0.55A
0.65A
0.72A
R3
6.19k
1%
VIN
4V
5V
7V
9V
LT1372 • TA07
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead PDIP (Narrow 0.300)
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1510)
(LTC DWG # 05-08-1610)
0.400*
(10.160)
MAX
0.189 – 0.197*
(4.801 – 5.004)
8
8
7
6
0.255 ± 0.015*
(6.477 ± 0.381)
0.009 – 0.015
(0.229 – 0.381)
(
2
3
0.045 – 0.065
(1.143 – 1.651)
)
5
0.150 – 0.157**
(3.810 – 3.988)
4
0.130 ± 0.005
(3.302 ± 0.127)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.100
(2.54)
BSC
0.125
(3.175) 0.020
MIN
(0.508)
MIN
0.018 ± 0.003
(0.457 ± 0.076)
N8 1098
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
2
3
4
0.053 – 0.069
(1.346 – 1.752)
0.008 – 0.010
(0.203 – 0.254)
0.065
(1.651)
TYP
+0.035
0.325 –0.015
+0.889
8.255
–0.381
6
0.228 – 0.244
(5.791 – 6.197)
1
0.300 – 0.325
(7.620 – 8.255)
7
5
0.004 – 0.010
(0.101 – 0.254)
0°– 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.014 – 0.019
(0.355 – 0.483)
TYP
0.050
(1.270)
BSC
SO8 1298
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
RELATED PARTS
PART NUMBER DESCRIPTION
COMMENTS
LT1370
High Efficiency DC/DC Converter
42V, 6A, 500kHz Switch
LT1767
1.5A, 1.25MHz Step-Down Switching Regulator
3V to 25V Input, VREF = 1.2V, Synchronizable Up to 2MHz, MSOP Package
LT1374
High Efficiency Step-Down Switching Regulator
25V, 4.5A, 500kHz Switch
LTC1735-1
High Efficiency Step-Down Controller with Power Good
Output Fault Protection, 16-Pin SSOP and SO-8
Single Cell, High Current (2A), Micropower, Synchronous
3MHz Step-Up DC/DC Converter
VIN = 0.7V to 5V, Up to 95% Efficiency Synchronizable Oscillator
from 100kHz to 3MHz
®
LTC 3402
12
Linear Technology Corporation
sn13727 13727fbs LT/TP 0401 2K REV B • PRINTED IN THE USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
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