LT3475/LT3475-1
Dual Step-Down
1.5A LED Driver
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DESCRIPTIO
FEATURES
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True Color PWMTM Delivers Constant Color with
3000:1 Dimming Range
Wide Input Range: 4V to 36V Operating, 40V
Maximum
Accurate and Adjustable Control of LED Current
from 50mA to 1.5A
High Side Current Sense Allows Grounded Cathode
LED Operation
Open LED (LT3475) and Short Circuit Protection
LT3475-1 Drives LED Strings Up to 25V
Accurate and Adjustable 200kHz to 2MHz
Switching Frequency
Anti-Phase Switching Reduces Ripple
Uses Small Inductors and Ceramic Capacitors
Available in the Compact 20-Lead TSSOP Thermally
Enhanced Surface Mount Package
The LT®3475/LT3475-1 are dual step-down DC/DC
converters designed to operate as a constant-current
source. An internal sense resistor monitors the output
current allowing accurate current regulation ideal for
driving high current LEDs. The high side current sense allows grounded cathode LED operation. High output current
accuracy is maintained over a wide current range, from
50mA to 1.5A, allowing a wide dimming range. Unique
PWM circuitry allows a dimming range of 3000:1, avoiding the color shift normally associated with LED current
dimming.
The high switching frequency offers several advantages,
permitting the use of small inductors and ceramic capacitors. Small inductors combined with the 20 lead TSSOP
surface mount package save space and cost versus
alternative solutions. The constant switching frequency
combined with low-impedance ceramic capacitors result
in low, predictable output ripple.
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APPLICATIO S
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Automotive and Avionic Lighting
Architectural Detail Lighting
Display Backlighting
Constant-Current Sources
With its wide input range of 4V to 36V, the LT3475/LT3475-1
regulate a broad array of power sources. A current mode
PWM architecture provides fast transient response and
cycle-by-cycle current limiting. Frequency foldback and
thermal shutdown provide additional protection.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners. Patents Pending.
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TYPICAL APPLICATIO
Dual Step-Down 1.5A LED Driver
Efficiency
VIN
5V TO 36V
4.7μF
BOOST2
SW1
*DIMMING
CONTROL
2.2μF
0.1μF
1.5A LED
CURRENT
10μH
85
SW2
OUT1
LED1
OUT2
LED2
PWM1
PWM2
VC1
VC2
REF
RT
VADJ1
DIMMING*
CONTROL
24.3k
1.5A LED
CURRENT
SINGLE WHITE 1.5A LED
75
70
fSW = 600kHz
60
55
3475 TA01
*SEE APPLICATIONS SECTION FOR DETAILS
80
65
2.2μF
0.1μF
VADJ2
GND
TWO SERIES CONNECTED
WHITE 1.5A LEDS
0.22μF
LT3475
10μH
VIN = 12V
90
EFFICIENCY (%)
0.22μF
95
SHDN
VIN
BOOST1
0
0.5
1
1.5
LED CURRENT (A)
3475 TA01b
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LT3475/LT3475-1
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ABSOLUTE
AXI U RATI GS
PIN CONFIGURATION
(Note 1)
TOP VIEW
VIN Pin .........................................................(-0.3V), 40V
BOOST Pin Voltage ...................................................60V
BOOST Above SW Pin ...............................................30V
OUT, LED, Pins (LT3475) ...........................................15V
OUT, LED Pins (LT3475-1).........................................25V
PWM Pin ...................................................................15V
VADJ Pin ......................................................................6V
VC, RT, REF Pins ..........................................................3V
SHDN Pin ...................................................................VIN
Maximum Junction Temperature (Note 2)............. 125°C
Operating Temperature Range (Note 3)
LT3475E/LT3475E-1 ............................. –40°C to 85°C
LT3475I/LT3475I-1 ............................. –40°C to 125°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature Range (Soldering, 10 sec) ....... 300°C
OUT1
1
20 PWM1
LED1
2
19 VADJ1
BOOST1
3
18 VC1
SW1
4
17 REF
VIN
5
VIN
6
SW2
7
14 RT
BOOST2
8
13 VC2
LED2
9
12 VADJ2
16 SHDN
21
OUT2 10
15 GND
11 PWM2
FE PACKAGE
20-LEAD PLASTIC TSSOP
TJMAX = 125°C, θJA = 30°C/W, θJC = 8°C/W
EXPOSED PAD (PIN 21) IS GROUND AND MUST
BE ELECTRICALLY CONNECTED TO THE PCB.
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3475EFE#PBF
LT3475EFE#TRPBF
LT3475EFE
20-Lead Plastic TSSOP
–40°C to 85°C
LT3475IFE#PBF
LT3475IFE#TRPBF
LT3475IFE
20-Lead Plastic TSSOP
–40°C to 125°C
LT3475EFE-1#PBF
LT3475EFE-1#TRPBF
LT3475FE-1
20-Lead Plastic TSSOP
–40°C to 85°C
LT3475IFE-1#PBF
LT3475IFE-1#TRPBF
LT3475FE-1
20-Lead Plastic TSSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3)
PARAMETER
CONDITIONS
MIN
●
Minimum Input Voltage
Input Quiescent Current
Not Switching
Shutdown Current
SHDN = 0.3V, VBOOST = VOUT = 0V
TYP
MAX
UNITS
3.7
4
V
6
8
mA
0.01
2
μA
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LT3475/LT3475-1
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3)
PARAMETER
CONDITIONS
LED Pin Current
VADJ Tied to VREF • 2/3
MIN
TYP
MAX
UNITS
1.00
0.350
●
0.97
0.94
0.336
0.325
0.31
1.03
1.04
0.364
0.375
0.385
A
A
A
A
A
●
1.22
1.25
1.27
V
●
VADJ Tied to VREF • 7/30
LT3475E/LT3475E-1 0°C to 85°C
REF Voltage
Reference Voltage Line Regulation
4V < VIN < 40V
Reference Voltage Load Regulation
0 < IREF < 500μA
●
VADJ Pin Bias Current (Note 4)
0.05
%/V
0.0002
%/μA
40
400
nA
Switching Frequency
RT = 24.3k
●
530
600
640
kHz
Maximum Duty Cycle
RT = 24.3k
RT = 4.32k
RT = 100k
●
90
95
80
98
Switching Phase
RT = 24.3k
150
180
Foldback Frequency
RT = 24.3k, VOUT = 0V
210
80
SHDN Threshold (to Switch)
SHDN Pin Current (Note 5)
%
%
%
VSHDN = 2.6V
PWM Threshold
Deg
kHz
2.5
2.6
2.74
V
7
9
11
μA
0.3
0.8
1.2
V
0.8
V
VC Source Current
VC Switching Threshold
VC = 1V
50
μA
VC Sink Current
VC = 1V
50
μA
LED to VC Transresistance
500
LED to VC Current Gain
V/A
1
mA/μA
VC to Switch Current Gain
2.6
A/V
VC Clamp Voltage
1.8
V
VC Pin Current in PWM Mode
VC = 1V, VPWM = 0.3V
●
VOUT = 4V, VPWM = 0.3V
●
OUT Pin Clamp Voltage (LT3475)
OUT Pin Current in PWM Mode
13.5
Switch Current Limit (Note 6)
2.3
10
400
nA
14
14.5
V
25
50
μA
2.7
3.2
A
Switch VCESAT
ISW =1.5A
350
500
mV
BOOST Pin Current
ISW =1.5A
25
40
mA
Switch Leakage Current
0.1
10
μA
Minimum Boost Voltage Above SW
1.8
2.5
V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
Note 3: The LT3475E and LT3475E-1 are guaranteed to meet performance
specifications from 0°C to 85°C. Specifications over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls. The LT3475I and LT3475I-1
are guaranteed to meet performance specifications over the –40°C to
125°C operating temperature range.
Note 4: Current flows out of pin.
Note 5: Current flows into pin.
Note 6: Current limit is guaranteed by design and/or correlation to static
test. Slope compensation reduces current limit at higher duty cycles.
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LT3475/LT3475-1
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TYPICAL PERFOR A CE CHARACTERISTICS
LED Current vs VADJ
1.50
LED Current vs Temperature
TA = 25°C
600
VADJ = VREF • 2/3
1.25
0.75
0.50
SWITCH ON VOLTAGE (mV)
1.00
0.8
0.6
VADJ = VREF • 7/30
0.4
0
–50 –25
0
0
0.25
0.5
0.75
VADJ (V)
1.25
1
100
MINIMUM
1.5
1.0
TA = 25°C
0
20
40
60
DUTY CYCLE (%)
Current Limit vs Output Voltage
3.5
3.0
3.0
2.5
2.5
2.0
1.5
1.0
50
25
75
0
TEMPERATURE (°C)
100
Oscillator Frequency
vs Temperature
1.0
0
125
0
550
500
450
125
1.5 2.0 2.5
VOUT (V)
3.0
3.5
4.0
3475 G06
TA = 25°C
RT = 24.3kΩ
600
TA = 25°C
1000
500
400
300
200
100
10
0
0.5
1.0
1.5
2.0
2.5
VOUT (V)
3475 G07
1.0
Oscillator Frequency vs RT
0
100
0.5
OSCILLATOR FREQUENCY (kHz)
OSCILLATOR FREQUENCY (kHz)
600
50
25
75
0
TEMPERATURE (˚C)
1.5
Oscillator Frequency Foldback
700
RT = 24.3kΩ
400
–50 –25
2.0
3475 G05
3475 G04
650
TA = 25°C
0.5
0
–50 –25
100
80
2.0
3475 G03
0.5
0
1.5
0.5
1.0
SWITCH CURRENT (A)
0
CURRENT LIMIT (A)
CURRENT LIMIT (A)
CURRENT LIMIT (A)
TYPICAL
0.5
OSCILLATOR FREQUENCY (kHz)
125
Switch Current Limit vs
Temperature
3.0
700
200
3475 G02
Switch Current Limit
vs Duty Cycle
2.0
300
0
50
25
75
0
TEMPERATURE (˚C)
3475 G01
2.5
400
100
0.2
0.25
TA = 25°C
500
1.0
LED CURRENT (A)
LED CURRENT (A)
Switch On Voltage
1.2
3475 G08
1
10
RT (kΩ)
100
3475 G09
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LT3475/LT3475-1
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TYPICAL PERFOR A CE CHARACTERISTICS
Boost Pin Current
35
7
TA = 25°C
50
TA = 25°C
14
INPUT CURRENT
LT3475-1
12
20
15
10
OUTPUT VOLTAGE (V)
25
5
4
3
2
1.0
LT3475-1
10
20
OUTPUT VOLTAGE
15
0
40
30
2
0
5
1.26
4
VIN (V)
1.27
10
TA = 25°C
TA = 25°C
TO START
9
TO RUN
LED VOLTAGE
8
TO START
LED VOLTAGE
7
2
1.23
1
100
125
0
0
Minimum Input Voltage, Two Series
Connected 1.5A White LEDs
3
1.24
40
3475 G12
VIN (V)
6
50
25
75
0
TEMPERATURE (˚C)
30
VIN (V)
Minimum Input Voltage, Single
1.5A White LED
1.28
1.22
–50 –25
20
10
3475 G11
Reference Voltage
4
LT3475
VIN (V)
3475 G10
1.25
6
20
5
0
2.0
1.5
SWITCH CURRENT (A)
8
25
0
0.5
0
LT3475
30
10
1
0
10
35
TO RUN
6
5
0
0.5
1
LED CURRENT (A)
3475 G13
1.5
3475 G14
0
0.5
1
LED CURRENT (A)
1.5
3475 G15
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PI FU CTIO S
OUT1, OUT2 (Pins 1, 10): The OUT pin is the input to the
current sense resistor. Connect this pin to the inductor
and the output capacitor.
BOOST1, BOOST2 (Pins 3, 8): The BOOST pin is used to
provide a drive voltage, higher than the input voltage, to
the internal bipolar NPN power switch.
LED1, LED2 (Pins 2, 9): The LED pin is the output of
the current sense resistor. Connect the anode of the LED
here.
GND (Pins 15, Exposed Pad Pin 21): Ground. Tie the GND
pin and the exposed pad directly to the ground plane. The
exposed pad metal of the package provides both electrical
contact to ground and good thermal contact to the printed
circuit board. The exposed pad must be soldered to the
circuit board for proper operation. Use a large ground plane
and thermal vias to optimize thermal performance.
VIN (Pins 5, 6): The VIN pins supply current to the internal
circuitry and to the internal power switches and must be
locally bypassed.
SW1, SW2 (Pins 4, 7): The SW pin is the output of the
internal power switch. Connect this pin to the inductor,
switching diode and boost capacitor.
3475fb
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INPUT CURRENT (mA)
INPUT CURRENT (mA)
40
5
VREF (V)
TA = 25°C
45
6
30
BOOST PIN CURRENT (mA)
Open-Circuit Output Voltage and
Input Current
Quiescent Current
LT3475/LT3475-1
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PI FU CTIO S
RT (Pin 14): The RT pin is used to set the internal
oscillator frequency. Tie a 24.3k resistor from RT to GND
for a 600kHz switching frequency.
VC1, VC2 (Pins 18, 13): The VC pin is the output of the
internal error amp. The voltage on this pin controls the
peak switch current. Use this pin to compensate the
control loop.
SHDN (Pin 16): The SHDN pin is used to shut down the
switching regulator and the internal bias circuits. The
2.6V switching threshold can function as an accurate
undervoltage lockout. Pull below 0.3V to shut down the
LT3475/LT3475-1. Pull above 2.6V to enable the LT3475/
LT3475-1. Tie to VIN if the SHDN function is unused.
VADJ1, VADJ2 (Pins 19, 12): The VADJ pin is the input to
the internal voltage-to-current amplifier. Connect the VADJ
pin to the REF pin for a 1.5A output current. For lower
output currents, program the VADJ pin using the following
formula: ILED = 1.5A • VADJ/1.25V.
REF (Pin 17): The REF pin is the buffered output of the
internal reference. Either tie the REF pin to the VADJ pin
for a 1.5A output current, or use a resistor divider to
generate a lower voltage at the VADJ pin. Leave this pin
unconnected if unused.
PWM1, PWM2 (Pins 20, 11): The PWM pin controls the
connection of the VC pin to the internal circuitry. When
the PWM pin is low, the VC pin is disconnected from the
internal circuitry and draws minimal current. If the PWM
feature is unused, leave this pin unconnected.
BLOCK DIAGRAM
VIN
RT
CIN
VIN
SHDN
RT
INT REG
AND
UVLO
D1
VIN
MASTER
OSC
C1
C1
Q
R
∑
SLOPE COMP
SLOPE COMP
MOSC 1
SW1
L1
Q
Q1
SLAVE
OSC
D3
∑
C2
C2
R
Q
S
Q
MOSC 2
S
DRIVER
Q2
SLAVE
OSC
FREQUENCY
FOLDBACK
L2
D4
FREQUENCY
FOLDBACK
OUT2
+
–
0.067Ω
100Ω
LED1
2V
2V
–
+
100Ω
COUT2
0.067Ω
LED2
gm1
DLED1
SW2
DRIVER
OUT1
COUT1
D2
BOOST2
BOOST1
gm2
DLED 2
1.25V
PWM 1
PWM2
Q3
Q4
VC1
VC2
1.25k
1.25k
CC1
CC2
VADJ1
REF
VADJ2 EXPOSED
PAD
GND
3475 BD
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LT3475/LT3475-1
OPERATION
The LT3475 is a dual constant frequency, current mode
regulator with internal power switches capable of generating constant 1.5A outputs. Operation can be best
understood by referring to the Block Diagram.
If the SHDN pin is tied to ground, the LT3475 is shut
down and draws minimal current from the input source
tied to VIN. If the SHDN pin exceeds 1V, the internal bias
circuits turn on, including the internal regulator, reference
and oscillator. The switching regulators will only begin to
operate when the SHDN pin exceeds 2.6V.
The switcher is a current mode regulator. Instead of directly
modulating the duty cycle of the power switch, the feedback
loop controls the peak current in the switch during each
cycle. Compared to voltage mode control, current mode
control improves loop dynamics and provides cycle-bycycle current limit.
A pulse from the oscillator sets the RS flip-flop and turns
on the internal NPN bipolar power switch. Current in the
switch and the external inductor begins to increase. When
this current exceeds a level determined by the voltage at
VC, current comparator C1 resets the flip-flop, turning
off the switch. The current in the inductor flows through
the external Schottky diode and begins to decrease. The
cycle begins again at the next pulse from the oscillator.
In this way, the voltage on the VC pin controls the current
through the inductor to the output. The internal error
amplifier regulates the output current by continually
adjusting the VC pin voltage. The threshold for switching
on the VC pin is 0.8V, and an active clamp of 1.8V limits
the output current.
The voltage on the VADJ pin sets the current through the
LED pin. The NPN, Q3, pulls a current proportional to the
voltage on the VADJ pin through the 100Ω resistor. The gm
amplifier servos the VC pin to set the current through the
0.067Ω resistor and the LED pin. When the voltage drop
across the 0.067Ω resistor is equal to the voltage drop
across the 100Ω resistor, the servo loop is balanced.
Tying the REF pin to the VADJ pin sets the LED pin current
to 1.5A. Tying a resistor divider to the REF pin allows the
programming of LED pin currents of less than 1.5A. LED
pin current can also be programmed by tying the VADJ pin
directly to a voltage source.
An LED can be dimmed with pulse width modulation
using the PWM pin and an external NFET. If the PWM
pin is unconnected or is pulled high, the part operates
nominally. If the PWM pin is pulled low, the VC pin is disconnected from the internal circuitry and draws minimal
current from the compensation capacitor. Circuitry drawing current from the OUT pin is also disabled. This way,
the VC pin and the output capacitor store the state of
the LED pin current until the PWM is pulled high again.
This leads to a highly linear relationship between pulse
width and output light, allowing for a large and accurate
dimming range.
The RT pin allows programming of the switching frequency.
For applications requiring the smallest external components
possible, a fast switching frequency can be used. If low
dropout or very high input voltages are required, a slower
switching frequency can be programmed.
During startup VOUT will be at a low voltage. The NPN,
Q3, can only operate correctly with sufficient voltage
of ≈1.7V at VOUT, A comparator senses VOUT and forces
the VC pin high until VOUT rises above 2V, and Q3 is operating correctly.
The switching regulator performs frequency foldback
during overload conditions. An amplifier senses when
VOUT is less than 2V and begins decreasing the oscillator
frequency down from full frequency to 15% of the nominal
frequency when VOUT = 0V. The OUT pin is less than 2V
during startup, short circuit, and overload conditions.
Frequency foldback helps limit switch current under these
conditions.
The switch driver operates either from VIN or from the
BOOST pin. An external capacitor and Schottky diode
are used to generate a voltage at the BOOST pin that
is higher than the input supply. This allows the driver
to saturate the internal bipolar NPN power switch for
efficient operation.
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LT3475/LT3475-1
APPLICATIONS INFORMATION
Open Circuit Protection
The LT3475 has internal open-circuit protection. If the LED
is absent or is open circuit, the LT3475 clamps the voltage
on the LED pin at 14V. The switching regulator then operates at a very low frequency to limit the input current. The
LT3475-1 has no internal open circuit protection. With the
LT3475-1, be careful not to violate the ABSMAX voltage of
th BOOST pin; if VIN > 25V, external open circuit protection
circuitry (as shown in Figure 1) may be necessary.The
output voltage during an open LED condition is shown in
the Typical Performance Characteristics section.
OUT
10k
100k
3475 F01
Figure 1. External Overvoltage Protection
Circuitry for the LT3475-1
LT3475
VIN
VIN
2.6V
Undervoltage Lockout
Undervoltage lockout (UVLO) is typically used in situations
where the input supply is current limited, or has high source
resistance. A switching regulator draws constant power
from the source, so the source current increases as the
source voltage drops. This looks like a negative resistance
load to the source and can cause the source to current limit
or latch low under low source voltage conditions. UVLO
prevents the regulator from operating at source voltages
where these problems might occur.
An internal comparator will force the part into shutdown when VIN falls below 3.7V. If an adjustable UVLO
threshold is required, the SHDN pin can be used. The
threshold voltage of the SHDN pin comparator is 2.6V. An
internal resistor pulls 9μA to ground from the SHDN pin
at the UVLO threshold.
Choose resistors according to the following formula:
R2 =
2.6V
VTH – 2.6V
– 9μA
R1
VTH = UVLO Threshold
Example: Switching should not start until the input is
above 8V.
VTH = 8V
R1=100k
2.6V
R2 =
= 57.6k
8V – 2.6V
– 9μA
100k
VC
22V
R1
VC
SHDN
9μA
C1
R2
GND
3475 F02
Figure 2. Undervoltage Lockout
Keep the connections from the resistors to the SHDN pin
short and make sure the coupling to the SW and BOOST
pins is minimized. If high resistance values are used, the
SHDN pin should be bypassed with a 1nF capacitor to
prevent coupling problems from switching nodes.
Setting the Switching Frequency
The LT3475 uses a constant frequency architecture that
can be programmed over a 200kHz to 2MHz range with a
single external timing resistor from the RT pin to ground.
A graph for selecting the value of RT for a given operating
frequency is shown in the Typical Applications section.
Table 1. Switching Frequencies
SWITCHING FREQUENCY (MHz)
RT (kΩ)
2
4.32
1.5
6.81
1.2
9.09
1
11.8
0.8
16.9
0.6
24.3
0.4
40.2
0.3
57.6
0.2
100
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LT3475/LT3475-1
APPLICATIONS INFORMATION
Table 1 shows suggested RT selections for a variety of
switching frequencies.
Operating Frequency Selection
The choice of operating frequency is determined by
several factors. There is a tradeoff between efficiency and
component size. A higher switching frequency allows the
use of smaller inductors at the cost of increased switching
losses and decreased efficiency.
Another consideration is the maximum duty cycle. In certain
applications, the converter needs to operate at a high duty
cycle in order to work at the lowest input voltage possible.
The LT3475 has a fixed oscillator off time and a variable
on time. As a result, the maximum duty cycle increases
as the switching frequency is decreased.
Input Voltage Range
The minimum operating voltage is determined either by the
LT3475’s undervoltage lockout of 4V, or by its maximum
duty cycle. The duty cycle is the fraction of time that the
internal switch is on and is determined by the input and
output voltages:
( VOUT + VF )
DC =
( VIN – VSW + VF )
where VF is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.4V at maximum load). This leads to a minimum input
voltage of:
V +V
VIN(MIN ) = OUT F – VF + VSW
DCMAX
with DCMAX = 1–tOFF(MIN) • f
where t0FF(MIN) is equal to 167ns and f is the switching
frequency.
The maximum operating voltage is determined by the
absolute maximum ratings of the VIN and BOOST pins,
and by the minimum duty cycle.
V +V
VIN(MAX ) = OUT F – VF + VSW
DCMIN
with DCMIN = tON(MIN) • f
where tON(MIN) is equal to 140ns and f is the switching
frequency.
Example: f = 750kHz, VOUT = 3.4V
DCMIN = 140ns • 750kHz = 0.105
3.4V + 0.4V
VIN(MAX ) =
– 0.4V + 0.4V = 36V
0.105
The minimum duty cycle depends on the switching frequency. Running at a lower switching frequency might
allow a higher maximum operating voltage. Note that
this is a restriction on the operating input voltage; the
circuit will tolerate transient inputs up to the Absolute
Maximum Ratings of the VIN and BOOST pins. The input
voltage should be limited to the VIN operating range (36V)
during overload conditions (short circuit or start up).
Minimum On Time
The LT3475 will regulate the output current at input voltages greater than VIN(MAX). For example, an application
with an output voltage of 3V and switching frequency of
1.2MHz has a VIN(MAX) of 20V, as shown in Figure 3. Figure
4 shows operation at 35V. Output ripple and peak inductor
VOUT
500mV/DIV
(AC COUPLED)
IL
1A/DIV
Example: f = 600kHz, VOUT = 4V
DCMAX = 1− 167ns • 600kHz = 0.90
4V + 0.4V
VIN(MIN ) =
– 0.4V + 0.4V = 4.9V
0.9
VSW
20V/DIV
3475 F03
Figure 3. Operation at VIN(MAX) = 20V.
VOUT = 3V and fSW = 1.2MHHz
3475fb
9
LT3475/LT3475-1
APPLICATIONS INFORMATION
current have significantly increased. Exceeding VIN(MAX)
is safe if the external components have adequate ratings
to handle the peak conditions and if the peak inductor
current does not exceed 3.2A. A saturating inductor may
further reduce performance.
VOUT
500mV/DIV
(AC COUPLED)
IL
1A/DIV
VSW
20V/DIV
3475 F04
Figure 4. Operation above VIN(MAX). Output
Ripple and Peak Inductor Current Increases
Table 2. Inductors
VALUE
(μH)
IRMS
(A)
DCR
( )
HEIGHT
(mm)
CR43-3R3
3.3
1.44
0.086
3.5
CR43-4R7
4.7
1.15
0.109
3.5
CDRH4D16-3R3
3.3
1.10
0.063
1.8
CDRH4D28-3R3
3.3
1.57
0.049
3.0
CDRH4D28-4R7
4.7
1.32
0.072
3.0
CDRH6D26-5R0
5.0
2.20
0.032
2.8
CDRH6D26-5R6
5.6
2.0
0.036
2.8
CDRH5D28-100
10
1.30
0.048
3.0
CDRH5D28-150
15
1.10
0.076
3.0
CDRH73-100
10
1.68
0.072
3.4
CDRH73-150
15
1.33
0.130
3.4
CDRH104R-150
15
3.1
0.050
4.0
DO1606T-332
3.3
1.30
0.100
2.0
DO1606T-472
4.7
1.10
0.120
2.0
DO1608C-332
3.3
2.00
0.080
2.9
DO1608C-472
4.7
1.50
0.090
2.9
MOS6020-332
3.3
1.80
0.046
2.0
MOS6020-472
10
1.50
0.050
2.0
DO3316P-103
10
3.9
0.038
5.2
DO3316P-153
15
3.1
0.046
5.2
PART NUMBER
Sumida
Coilcraft
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is:
L = (VOUT + VF ) •
1.2MHz
f
where VF is the voltage drop of the catch diode (~0.4V),
f is the switching frequency and L is in μH. With this value
the maximum load current will be above 1.6A at all duty
cycles. The inductor’s RMS current rating must be greater
than the maximum load current and its saturation current
should be at least 30% higher. For highest efficiency,
the series resistance (DCR) should be less than 0.15Ω.
Table 2 lists several vendors and types that are suitable.
For robust operation at full load and high input voltages
(VIN > 30V), use an inductor with a saturation current
higher than 3.2A.
The optimum inductor for a given application may differ
from the one indicated by this simple design guide. A larger
value inductor provides a higher maximum load current, and
reduces the output voltage ripple. If your load is lower than
the maximum load current, then you can relax the value of the
inductor and operate with higher ripple current. This allows
you to use a physically smaller inductor, or one with a lower
DCR resulting in higher efficiency. In addition, low inductance
may result in discontinuous mode operation, which further
reduces maximum load current. For details of maximum
output current and discontinuous mode operation, see Linear
Technology’s Application Note 44. Finally, for duty cycles
greater than 50% (VOUT/VIN > 0.5), a minimum inductance
is required to avoid sub-harmonic oscillations:
L MIN = (VOUT + VF ) •
800kHz
f
3475fb
10
LT3475/LT3475-1
APPLICATIONS INFORMATION
The current in the inductor is a triangle wave with an average
value equal to the load current. The peak switch current
is equal to the output current plus half the peak-to-peak
inductor ripple current. The LT3475 limits its switch current in order to protect itself and the system from overload
faults. Therefore, the maximum output current that the
LT3475 will deliver depends on the switch current limit,
the inductor value, and the input and output voltages.
When the switch is off, the potential across the inductor
is the output voltage plus the catch diode drop. This gives
the peak-to-peak ripple current in the inductor
ΔIL =
(1– DC)( VOUT + VF )
(L • f )
where f is the switching frequency of the LT3475 and L
is the value of the inductor. The peak inductor and switch
current is
ΔI
ISW (PK ) = IL (PK ) = IOUT + L
2
To maintain output regulation, this peak current must be
less than the LT3475’s switch current limit ILIM. ILIM is at
least 2.3A at low duty cycles and decreases linearly to 1.8A
at DC = 0.9. The maximum output current is a function of
the chosen inductor value:
IOUT (MAX ) = ILIM –
ΔIL
2
= 2.3A• (1–0.25•DC) –
ΔIL
2
Choosing an inductor value so that the ripple current is
small will allow a maximum output current near the switch
current limit.
One approach to choosing the inductor is to start with the
simple rule given above, look at the available inductors,
and choose one to meet cost or space goals. Then use
these equations to check that the LT3475 will be able to
deliver the required output current. Note again that these
equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT is less
than ΔIL/2.
Input Capacitor Selection
Bypass the input of the LT3475 circuit with a 4.7μF or
higher ceramic capacitor of X7R or X5R type. A lower
value or a less expensive Y5V type will work if there is
additional bypassing provided by bulk electrolytic capacitors or if the input source impedance is low. The following
paragraphs describe the input capacitor considerations in
more detail.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at
the LT3475 input and to force this switching current into a
tight local loop, minimizing EMI. The input capacitor must
have low impedance at the switching frequency to do this
effectively, and it must have an adequate ripple current rating. With two switchers operating at the same frequency
but with different phases and duty cycles, calculating the
input capacitor RMS current is not simple. However, a
conservative value is the RMS input current for the channel
that is delivering most power (VOUT • IOUT):
CINRMS = IOUT •
VOUT (VIN – VOUT ) IOUT
<
VIN
2
and is largest when VIN = 2VOUT (50% duty cycle). As the
second, lower power channel draws input current, the
input capacitor’s RMS current actually decreases as the
out-of-phase current cancels the current drawn by the
higher power channel. Considering that the maximum
load current from a single channel is ~1.5A, RMS ripple
current will always be less than 0.75A.
The high frequency of the LT3475 reduces the energy
storage requirements of the input capacitor, so that the
capacitance required is less than 10μF. The combination
of small size and low impedance (low equivalent series
resistance or ESR) of ceramic capacitors makes them the
preferred choice. The low ESR results in very low voltage
ripple. Ceramic capacitors can handle larger magnitudes
of ripple current than other capacitor types of the same
value. Use X5R and X7R types.
3475fb
11
LT3475/LT3475-1
APPLICATIONS INFORMATION
An alternative to a high value ceramic capacitor is a
lower value ceramic along with a larger electrolytic
capacitor. The electrolytic capacitor likely needs to be greater
than 10μF in order to meet the ESR and ripple current
requirements. The input capacitor is likely to see high
surge currents when the input source is applied. Tantalum capacitors can fail due to an over-surge of current.
Only use tantalum capacitors with the appropriate surge
current rating. The manufacturer may also recommend
operation below the rated voltage of the capacitor.
A final caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plugging the circuit into a live power source) this tank can ring,
doubling the input voltage and damaging the LT3475. The
solution is to either clamp the input voltage or dampen the
tank circuit by adding a lossy capacitor in parallel with the
ceramic capacitor. For details, see Application Note 88.
Output Capacitor Selection
For most LEDs, a 2.2μF, 6.3V ceramic capacitor (X5R or
X7R) at the output results in very low output voltage ripple
and good transient response. Other types and values will
also work. The following discusses tradeoffs in output
ripple and transient performance.
The output capacitor filters the inductor current to
generate an output with low voltage ripple. It also stores
energy in order to satisfy transient loads and stabilizes the
LT3475’s control loop. Because the LT3475 operates at a
high frequency, minimal output capacitance is necessary.
In addition, the control loop operates well with or without
the presence of output capacitor series resistance (ESR).
Ceramic capacitors, which achieve very low output ripple
and small circuit size, are therefore an option.
You can estimate output ripple with the following
equation:
VRIPPLE = ΔIL / (8 • f • COUT) for ceramic capacitors
where ΔIL is the peak-to-peak ripple current in the
inductor. The RMS content of this ripple is very low so the
RMS current rating of the output capacitor is usually not
of concern. It can be estimated with the formula:
IC(RMS) = ΔIL / 12
The low ESR and small size of ceramic capacitors make
them the preferred type for LT3475 applications. Not all
ceramic capacitors are the same, however. Many of the
higher value capacitors use poor dielectrics with high
temperature and voltage coefficients. In particular Y5V
and Z5U types lose a large fraction of their capacitance
with applied voltage and at temperature extremes.
Because loop stability and transient response depend on
the value of COUT, this loss may be unacceptable. Use X7R
and X5R types. Table 3 lists several capacitor vendors.
Table 3. Low ESR Surface Mount Capacitors.
VENDOR
TYPE
SERIES
Taiyo-Yuden
Ceramic
X5R, X7R
AVX
Ceramic
X5R, X7R
TDK
Ceramic
X5R, X7R
Diode Selection
The catch diode (D3 from the Block Diagram) conducts
current only during switch off time. Average forward current in normal operation can be calculated from:
ID(AVG) = IOUT (VIN – VOUT)/VIN
The only reason to consider a diode with a larger current
rating than necessary for nominal operation is for the
worst-case condition of shorted output. The diode current will then increase to one half the typical peak switch
current limit.
Peak reverse voltage is equal to the regulator input
voltage. Use a diode with a reverse voltage rating greater
than the input voltage. Table 4 lists several Schottky
diodes and their manufacturers.
Diode reverse leakage can discharge the output capacitor
during LED off times while PWM dimming. If operating at
high ambient temperatures, use a low leakage Schottky
for the widest PWM dimming range.
3475fb
12
LT3475/LT3475-1
APPLICATIONS INFORMATION
Table 4. Schottky Diodes
VF at 1A
(mV)
VR
(V)
IAVE(A)
(A)
MBR0540
40
0.5
620
MBRM120E
20
1
530
MBRM140
40
1
550
B120
20
1
500
B130
30
1
500
B140HB
40
1
530
DFLS140
40
1.1
510
B240
40
2
30
1
VF at 2A
(mV)
On Semiconductor
Diodes Inc
500
International Rectifier
10BQ030
420
BOOST Pin Considerations
The capacitor and diode tied to the BOOST pin generate a voltage that is higher than the input voltage. In
most cases, a 0.22μF capacitor and fast switching diode
(such as the CMDSH-3 or MMSD914LT1) will work well.
Figure 5 shows three ways to arrange the boost circuit.
The BOOST pin must be more than 2.5V above the SW
pin for full efficiency. For outputs of 3.3V and higher, the
standard circuit (Figure 5a) is best. For outputs between
2.8V and 3.3V, use a small Schottky diode (such as the
BAT-54). For lower output voltages, the boost diode can be
tied to the input (Figure 5b). The circuit in Figure 5a is more
efficient because the BOOST pin current comes from a
lower voltage source. The anode of the boost diode can
be tied to another source that is at least 3V. For example, if
you are generating a 3.3V output, and the 3.3V output is on
whenever the LED is on, the BOOST pin can be
connected to the 3.3V output. For LT3475-1 applications
with higher output voltages, an additional Zener diode
may be necessary (Figure 5d) to maintain pin voltage
below the absolute maximum. In any case, be sure that
the maximum voltage at the BOOST pin is both less than
60V and the voltage difference between the BOOST and
SW pins is less than 30V.
The minimum operating voltage of an LT3475 application
is limited by the undervoltage lockout (~3.7V) and by the
maximum duty cycle. The boost circuit also limits the
minimum input voltage for proper start up. If the input
voltage ramps slowly, or the LT3475 turns on when the
output is already in regulation, the boost capacitor may
not be fully charged. Because the boost capacitor charges
D2
D2
C3
BOOST
VIN
VIN
C3
BOOST
LT3475
LT3475
VOUT
SW
VIN
VIN
VOUT
SW
GND
GND
VBOOST – VSW ≅ VIN
MAX VBOOST ≅ 2VIN
VBOOST – VSW ≅ VOUT
MAX VBOOST ≅ VIN + VOUT
(5a)
(5b)
D2
D2
VIN2 > 3V
BOOST
BOOST
C3
LT3475
VIN
VIN
C3
LT3475
SW
VOUT
VIN
VIN
GND
VOUT
SW
GND
3475 F05
VBOOST – VSW ≅ VIN2
MAX VBOOST ≅ VIN2 + VIN
MINIMUM VALUE FOR VIN2 = 3V
VBOOST – VSW – VZ
MAX VBOOST ≅ VIN + VOUT – VZ
(5c)
3475 F05
(5d)
Figure 5. Generating the Boost Voltage
3475fb
13
LT3475/LT3475-1
APPLICATIONS INFORMATION
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The minimum load current generally goes to zero
once the circuit has started. The typical performance characteristics section shows a plot of minimum load to start
and to run as a function of input voltage. Even without an
output load current, in many cases the discharged output
capacitor will present a load to the switcher that will allow
it to start. The plots show the worst case, where VIN is
ramping very slowly.
Programming LED Current
The LED current can be set by adjusting the voltage on the
VADJ pin. For a 1.5A LED current, either tie VADJ to REF or
to a 1.25V source. For lower output currents, program the
VADJ using the following formula: ILED = 1.5A • VADJ/1.25V.
Voltages less than 1.25V can be generated with a voltage
divider from the REF pin, as shown in Figure 6. In order
to have accurate LED current, precision resistors are
preferred (1% or better is recommended). Note that the
VADJ pin sources a small amount of bias current, so use
the following formula to choose resistors:
R2 =
the voltage on the VADJ pin by tying a low on resistance
FET to the resistor divider string. This allows the selection of two different LED currents. For reliable operation program an LED current of no less than 50mA.
The maximum current dimming ratio (IRATIO) can be
calculated from the maximum LED current (IMAX) and the
minimum LED current (IMIN) as follows:
IMAX/IMIN = IRATIO
Another dimming control circuit (Figure 8) uses the PWM
pin and an external NFET tied to the cathode of the LED.
An external PWM signal is applied to the PWM pin and the
gate of the NFET (For PWM dimming ratios of 20 to 1 or
less, the NFET can be omitted). The average LED current is
proportional to the duty cycle of the PWM signal. When the
PWM signal goes low, the NFET turns off, turning off the
LED and leaving the output capacitor charged. The PWM
pin is pulled low as well, which disconnects the VC pin,
storing the voltage in the capacitor tied there. Use the C-RC
string shown in Figure 8 and Figure 9 tied to the VC pin for
proper operation during startup. When the PWM pin goes
high again, the LED current returns rapidly to its previous
on state since the compensation and output capacitors are
at the correct voltage. This fast settling time allows the
VADJ
1.25V – VADJ
+ 50nA
R1
REF
R1
To minimize the error from variations in VADJ pin current,
use resistors with a parallel resistance of less than 4k. Use
resistor strings with a high enough series resistance so as not
to exceed the 500μA current compliance of the REF pin.
Dimming Control
LT3475
VADJ
GND
R2
3475 F07
DIM
Figure 7. Dimming with a MOSFET and Resistor Divider
There are several different types of dimming control
circuits. One dimming control circuit (Figure 7) changes
PWM
100Hz TO
10kHz
VC
PWM
10k
LT3475
REF
3.3nF
LED
R1
LT3475
VADJ
R2
0.1μF
GND
3475 F08
GND
3475 F06
Figure 6. Setting VADJ with a Resistor Divider
Figure 8. Dimming Using PWM Signal
3475fb
14
LT3475/LT3475-1
APPLICATIONS INFORMATION
LT3475 to maintain diode current regulation with PWM
pulse widths as short as 7.5 switching cycles (12.5μs for
fSW = 600kHz). Maximum PWM period is determined by
the system and is unlikely to be longer than 12ms. Using
PWM periods shorter than 100μs is not recommended.
The maximum PWM dimming ratio (PWMRATIO) can be
calculated from the maximum PWM period (tMAX) and
minimum PWM pulse width (tMIN) as follows:
tMAX/tMIN = PWMRATIO
Total dimming ratio (DIMRATIO) is the product of the PWM
dimming ratio and the current dimming ratio.
Example:
IMAX = 1A, IMIN = 0.1A, tMAX = 9.9ms
tMIN = 3.3μs (fSW = 1.4MHz)
IRATIO = 1A/0.1A =10:1
PWMRATIO = 9.9ms/3.3μs = 3000:1
DIMRATIO = 10 • 3000 = 30000:1
Layout Hints
As with all switching regulators, careful attention must
be paid to the PCB layout and component placement. To
maximize efficiency, switch rise and fall times are made
as short as possible. To prevent electromagnetic interference (EMI) problems, proper layout of the high frequency
switching path is essential. The voltage signal of the SW
and BOOST pins have sharp rise and fall edges. Minimize
the area of all traces connected to the BOOST and SW
pins and always use a ground plane under the switching
regulator to minimize interplane coupling. In addition, the
ground connection for frequency setting resistor RT and
capacitors at VC1, VC2 pins (refer to the Block Diagram)
should be tied directly to the GND pin and not shared
with the power ground path, ensuring a clean, noise-free
connection.
20
19
18
17
16
15
14
13
12
11
3
4
5
6
7
8
9
10
PWM2
2
To achieve the maximum PWM dimming ratio, use the
circuit shown in Figure 9. This allows PWM pulse widths
as short as 4.5 switching cycles (7.5μs for fSW = 600kHz).
Note that if you use the circuit in Figure 9, the rising edge
of the two PWM signals must align within 100ns.
SHDN
1
PWM1
VIN
220pF
RT
VC
10k
LT3475
1M
3.3nF
0.1μF
PWM1
RT
GND
3475 F09
3475 F10
VIA TO LOCAL GND PLANE
Figure 9. Extending the PWM Dimming Range
Figure 10. Recommended Component Placement
3475fb
15
LT3475/LT3475-1
TYPICAL APPLICATIONS
Dual Step-Down 1A LED Driver
VIN
5V TO 36V
C1
4.7μF
50V
D3
VIN
SHDN
BOOST1
C4
0.22μF
6.3V
L2
10μH
D4
BOOST2
C3
0.22μF
6.3V
LT3475
SW1
L1
10μH
SW2
D1
D2
C5
2.2μF
6.3V
OUT1
OUT2
LED1
LED2
C2
2.2μF
6.3V
LED 1
LED 2
C6
0.1μF
R2
1k
VC1
VC2
REF
RT
VADJ1
C7
0.1μF
VADJ2
GND
R3
2k
R1
24.3k
3475 TA02
C1 TO C5: X5R OR X7R
D1, D2: DFLS140
D3, D4: MBR0540
LED CURRENT = 1A
fSW = 600kHz
Dual Step-Down 1.5A LED Driver with 1200 : 1 True Color PWM Dimming
VIN
6V TO 36V
C1
4.7μF
50V
D3
VIN
SHDN
BOOST1
C2
0.22μF
6.3V
L2
10μH
1.5A LED
CURRENT
C3
0.22μF
6.3V
LT3475
SW1
C4
2.2μF
6.3V
D4
BOOST2
L1
10μH
SW2
C5
2.2μF
6.3V
D2
D1
OUT1
OUT2
LED1
LED2
PWM1
PWM2
LED 1
LED 2
R3
10k
C6
3.3nF
VC1
VC2
REF
RT
VADJ1
C7
3.3nF
R4
10k
VADJ2
M1
M2
GND
C8
0.1μF
C9
0.1μF
M3
C8
220p
1M
R2
R1
24.3k
fSW = 600kHz
PWM1
1.5A LED
CURRENT
3475 TA03
PWM2
D1, D2: B260
D3, D4: MBR0540
C1 TO C5: X5R OR X7R
M1, M2: Si2302ADS
M3: 2n7002L
3475fb
16
LT3475/LT3475-1
TYPICAL APPLICATIONS
Step-Down 3A LED Driver
VIN
5V TO 36V
C1
4.7μF
50V
D3
VIN
SHDN
BOOST1
C2
0.22μF
6.3V
L2
10μH
D4
BOOST2
C3
0.22μF
6.3V
LT3475
L1
10μH
SW2
SW1
C5
2.2μF
6.3V
D2
D1
OUT1
OUT2
C4
2.2μF
6.3V
LED1
LED2
C6
0.1μF
VC1
VC2
REF
RT
VADJ1
C7
0.1μF
VADJ2
R1
24.3k
GND
3A LED
CURRENT
LED 1
fSW = 600kHz
D1, D2: B240A
D3, D4: MBR0540
C1 TO C5: X5R OR X7R
3475 TA04
Dual Step-Down LED Driver with Series Connected LEDs
VIN
10V TO 36V
D3
C1
4.7μF
50V
VIN
SHDN
BOOST1
C2
0.22μF
10V
L2
15μH
D4
BOOST2
C3
0.22μF
10V
LT3475
SW2
SW1
D2
D1
C4
2.2μF
10V
1.5A LED
CURRENT
L1
15μH
OUT1
OUT2
LED1
LED2
C5
2.2μF
10V
LED 1
VC1
VC2
LED 2
C6
0.1μF
REF
RT
C7
0.1μF
LED 3
D1, D2: B240A
D3, D4: MMSD4148T1
C1 TO C5: X5R OR X7R
VADJ1
VADJ2
GND
R1
24.3k
1.5A LED
CURRENT
LED 4
fSW = 600kHz
3475 TA05
3475fb
17
LT3475/LT3475-1
TYPICAL APPLICATIONS
Dual Step-Down 1.5A Red LED Driver
VIN
5V TO 28V
C1
4.7μF
35V
D3
VIN
BOOST1
C2
0.22μF
35V
L2
10μH
D4
SHDN
BOOST2
C3
0.22μF
35V
LT3475
SW1
SW2
D2
D1
C4
2.2μF
6.3V
C6
0.1μF
1.5A LED
CURRENT
LED 1
D1, D2: B240A
D3, D4: MMSD4148T1
C1 TO C5: X5R OR X7R
L1
10μH
OUT1
OUT2
LED1
LED2
VC1
VC2
REF
RT
VADJ1
VADJ2
GND
C5
2.2μF
6.3V
C7
0.1μF
R1
24.3k
LED 2
1.5A LED
CURRENT
fSW = 600kHz
3475 TA06
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18
LT3475/LT3475-1
PACKAGE DESCRIPTION
FE Package
20-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation CB
6.40 – 6.60*
(.252 – .260)
3.86
(.152)
3.86
(.152)
20 1918 17 16 15 14 13 12 11
6.60 ±0.10
2.74
(.108)
4.50 ±0.10
6.40
2.74 (.252)
(.108) BSC
SEE NOTE 4
0.45 ±0.05
1.05 ±0.10
0.65 BSC
1 2 3 4 5 6 7 8 9 10
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.09 – 0.20
(.0035 – .0079)
0.25
REF
0.50 – 0.75
(.020 – .030)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
3. DRAWING NOT TO SCALE
1.20
(.047)
MAX
0° – 8°
0.65
(.0256)
BSC
0.195 – 0.30
(.0077 – .0118)
TYP
0.05 – 0.15
(.002 – .006)
FE20 (CB) TSSOP 0204
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
3475fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However,
no responsibility is assumed for its use. Linear Technology Corporation makes no representation that
the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LT3475/LT3475-1
TYPICAL APPLICATION
Dual Step-Down 1.5A LED Driver with Four Series Connected LED Output
VIN
21V TO 36V
D1
C1
4.7μF
50V
D2
C2
0.22μF
16V
L1
33μH
SHDN
VIN
BOOST1
LT3475-1
SW1
R1
1k
D7
C4
2.2μF
25V
D5
D6
Q1
L2
33μH
R2
1k
12V TO 18V LED VOLTAGE
VC1
VC2
REF
RT
VADJ1
R6
100k
D4
OUT2
LED2
12V TO 18V LED VOLTAGE
C6
0.1μF
C3
0.22μF
16V
SW2
OUT1
LED1
R4
10k
D3
BOOST2
VADJ2
GND
1.5A LED
CURRENT*
R5
10k
C7
0.1μF
R3
24.3k
1.5A LED
CURRENT*
C5
2.2μF
25V
D8
R7
100k
Q2
fSW = 600kHz
3475 TA08
D1, D4: 7.5V ZENER DIODE
D2, D3: MMSD4148
D5, D6: B240A
D7, D8: 22V ZENER DIODE
R1, R2: USE 0.5W RESISTOR OF TWO 2k 0.25W RESISTORS IN PARALLEL
Q1, Q2: MMBT3904
C1 TO C5: X5R or X7R
*DERATE LED CURRENT AT ELEVATED AMBIENT TEMPERATURES TO MAINTAIN LT3475-1 JUNCTION TEMPERATURE BELOW 125 °C.
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1618
Constant-Current, 1.4MHz, 1.5A Boost
Converter
VIN(MIN) = 1.6V, VIN(MAX) = 18V, VOUT(MAX) = 35V, Analog/PWM, ISD < 1μA,
MS10 Package
LT3466
Dual Full Function Step-Up LED Driver
Drivers Up to 20 LEDs, VIN: 2.7V to 24V, VOUT(MAX) = 40V, DFN, TSSOP16E Packages
LT3474
36V, 1A (ILED), 2MHz Step-Down
LED Driver
VIN(MIN) = 4V, VIN(MAX) = 36V, 400:1 True Color PWM, ISD < 1μA,
TSSOP16E Package
LT3477
42V, 3A, 3.5MHz Boost, Buck-Boost,
Buck LED Driver
VIN(MIN) = 2.5V, VIN(MAX) = 25V, VOUT(MAX) = 40V, Analog/PWM, ISD < 1μA,
QFN, TSSOP20E Packages
LT3479
3A, Full-Featured DC/DC Converter with VIN(MIN) = 2.5V, VIN(MAX) = 24V, VOUT(MAX) = 40V, Analog/PWM, ISD < 1μA,
Soft-Start and Inrush Current Protection DFN, TSSOP Packages
LT3846
Dual 1.3A, 2MHz, LED Driver
VIN: 2.5V to 24V, VOUT(MAX) = 36V, 1000:1 True Color PWMTM Dimmin,
DFN, TSSOP16E Packages
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20
Linear Technology Corporation
LT 1007 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2006
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