LT3481
36V, 2A, 2.8MHz Step-Down
Switching Regulator with
50µA Quiescent Current
FEATURES
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DESCRIPTION
Wide Input Range: 3.6V to 34V Operating,
36V Maximum
2A Maximum Output Current
Low Ripple Burst Mode® Operation
50μA IQ at 12VIN to 3.3VOUT
Output Ripple < 15mV
Adjustable Switching Frequency: 300kHz to 2.8MHz
Low Shutdown Current: IQ < 1μA
Integrated Boost Diode
Power Good Flag
Saturating Switch Design: 0.18Ω On-Resistance
1.265V Feedback Reference Voltage
Output Voltage: 1.265V to 20V
Soft-Start Capability
Synchronizable Between 275kHz to 475kHz
Small 10-Pin Thermally Enhanced MSOP and
(3mm x 3mm) DFN Packages
APPLICATIONS
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The LT®3481 is an adjustable frequency (300kHz to 2.8MHz)
monolithic buck switching regulator that accepts input
voltages up to 34V (36V maximum). A high efficiency
0.18Ω switch is included on the die along with a boost
Schottky diode and the necessary oscillator, control, and
logic circuitry. Current mode topology is used for fast
transient response and good loop stability. Low ripple
Burst Mode operation maintains high efficiency at low
output currents while keeping output ripple below 15mV
in a typical application. In addition, the LT3481 can further enhance low output current efficiency by drawing
bias current from the output when VOUT is above 3V.
Shutdown reduces input supply current to less than 1μA
while a resistor and capacitor on the RUN/SS pin provide a
controlled output voltage ramp (soft-start). A power good
flag signals when VOUT reaches 90% of the programmed
output voltage. The LT3481 is available in 10-Pin MSOP
and 3mm x 3mm DFN packages with exposed pads for
low thermal resistance.
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered
trademarks of Linear Technology Corporation. All other trademarks are the property of their
respective owners.
Automotive Battery Regulation
Power for Portable Products
Distributed Supply Regulation
Industrial Supplies
Wall Transformer Regulation
TYPICAL APPLICATION
3.3V Step-Down Converter
VIN
4.5V TO
34V
VIN
BD
RUN/SS
EFFICIENCY (%)
0.47μF
4.7μH
VC
LT3481
SW
RT
330pF
80
1000.0
70
100.0
60
10.0
50
1.0
BIAS
PG
60.4k
10000.0
BOOST
16.2k
4.7μF
90
324k
GND
40
FB
22μF
200k
3481 TA01
VIN = 12V
VOUT = 3.3V
L = 4.7μ
F = 800 kHz
30
0.0001
0.001
0.01
0.1
LOAD CURRENT (A)
1
POWER LOSS (mW)
OFF ON
Efficiency
VOUT
3.3V
2A
0.1
0.01
10
3481 TA01b
3481fc
1
LT3481
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VIN, RUN/SS Voltage .................................................36V
BOOST Pin Voltage ...................................................56V
BOOST Pin Above SW Pin.........................................30V
FB, RT, VC Voltage .......................................................5V
BIAS, PG, BD Voltage ................................................30V
Maximum Junction Temperature........................... 125°C
LT3481E, LT3481I ............................................. 125°C
LT3481H ........................................................... 150°C
Operating Temperature Range (Note 2)
LT3481E............................................... –40°C to 85°C
LT3481I.............................................. –40°C to 125°C
LT3481H ............................................ –40°C to 150°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
MSE Only .......................................................... 300°C
PIN CONFIGURATION
TOP VIEW
TOP VIEW
BD
1
10 RT
BOOST
2
9 VC
SW
3
VIN
4
7 BIAS
RUN/SS
5
6 PG
11
BD
BOOST
SW
VIN
RUN/SS
8 FB
1
2
3
4
5
11
10
9
8
7
6
RT
VC
FB
BIAS
PG
MSE PACKAGE
10-LEAD PLASTIC MSOP
DD PACKAGE
10-LEAD (3mm s 3mm) PLASTIC DFN
EXPOSED PAD (PIN 11) IS GND
MUST BE CONNECTED TO GND
EXPOSED PAD (PIN 11) IS GND
MUST BE CONNECTED TO GND
θJA = 40°C/W
θJA = 43°C/W
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3481EDD#PBF
LT3481EDD#TRPBF
LBVS
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LT3481IDD#PBF
LT3481IDD#TRPBF
LBVV
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3481HDD#PBF
LT3481HDD#TRPBF
LBVS
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 150°C
LT3481EMSE#PBF
LT3481EMSE#TRPBF
LTBVT
10-Lead Plastic MSOP
–40°C to 85°C
LT3481IMSE#PBF
LT3481IMSE#TRPBF
LTBVW
10-Lead Plastic MSOP
–40°C to 125°C
LT3481HMSE#PBF
LT3481HMSE#TRPBF
LTBVT
10-Lead Plastic MSOP
–40°C to 150°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3481EDD
LT3481EDD#TR
LBVS
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LT3481IDD
LT3481IDD#TR
LBVV
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3481HDD
LT3481HDD#TR
LBVS
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 150°C
LT3481EMSE
LT3481EMSE#TR
LTBVT
10-Lead Plastic MSOP
–40°C to 85°C
LT3481IMSE
LT3481IMSE#TR
LTBVW
10-Lead Plastic MSOP
–40°C to 125°C
LT3481HMSE
LT3481HMSE#TR
LTBVT
10-Lead Plastic MSOP
–40°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3481fc
2
LT3481
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUNS/SS = 10V VBOOST = 15V, VBIAS = 3.3V unless otherwise
noted. (Note 2)
PARAMETER
CONDITIONS
MIN
●
Minimum Input Voltage
Quiescent Current from VIN
VRUN/SS = 0.2V
VBIAS = 3V, Not Switching
●
VBIAS = 0, Not Switching
Quiescent Current from BIAS
VRUN/SS = 0.2V
VBIAS = 3V, Not Switching
●
VBIAS = 0, Not Switching
Minimum Bias Voltage
Feedback Voltage
●
FB Pin Bias Current (Note 3)
VFB = 1.25V, VC = 0.4V
FB Voltage Line Regulation
4V < VIN < 34V
1.25
1.24
●
TYP
MAX
UNITS
3
3.6
V
0.01
0.5
μA
22
60
μA
75
120
μA
0.01
0.5
μA
50
120
μA
0
5
μA
2.7
3
V
1.265
1.265
1.29
1.3
V
V
30
100
nA
0.002
0.02
%/V
Error Amp gm
330
Error Amp Gain
800
VC Source Current
65
μMho
μA
VC Sink Current
85
μA
VC Pin to Switch Current Gain
3.5
A/V
VC Clamp Voltage
Switching Frequency
2
RT = 8.66k
RT = 29.4k
RT = 187k
2.5
1.25
250
●
Minimum Switch Off-Time
3.2
V
2.8
1.4
300
3.1
1.55
350
MHz
MHz
kHz
130
200
nS
3.8
4.4
A
Switch Current Limit
Duty Cycle = 5%
Switch VCESAT
ISW = 2A
360
Boost Schottky Reverse Leakage
VSW = 10V, VBIAS = 0V
0.02
2
μA
1.5
2.1
V
●
Minimum Boost Voltage (Note 4)
mV
BOOST Pin Current
ISW = 1A
18
35
mA
RUN/SS Pin Current
VRUN/SS = 2.5V
5
10
μA
0.2
V
RUN/SS Input Voltage High
2.5
V
RUN/SS Input Voltage Low
PG Threshold Offset from Feedback Voltage
VFB Rising
122
PG Leakage
VPG = 5V
0.1
PG Sink Current
VPG = 3V
PG Hysteresis
mV
5
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3481E is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LT3481I specifications are
●
100
mV
1
μA
600
μA
guaranteed over the –40°C to 125°C temperature range. The LT3481H
specifications are guaranteed over the –40°C to 150°C operating
temperature range. High junction temperatures degrade operating
lifetimes. Operating lifetime is derated at junction temperatures greater
than 125°C.
Note 3: Bias current flows into the FB pin.
Note 4: This is the minimum voltage across the boost capacitor needed
to guarantee full saturation of the switch.
3481fc
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LT3481
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency (VOUT = 3.3V)
Efficiency (VOUT = 5.0V)
90
VIN = 12V
80
EFFICIENCY (%)
60
50
40
60
20
0
0.0001
0.001
0.01
0.1
LOAD CURRENT (A)
1
VIN = 24V
40
20
L: NEC PLC-0745-4R7
f: 800kHz
0
0.0001
0.001
0.01
0.1
LOAD CURRENT (A)
1
30
20
10
VIN = 12V
VOUT = 3.3V
3.5
250
200
INCREASED SUPPLY
CURRENT DUE TO CATCH
DIODE LEAKAGE AT
HIGH TEMPERATURE
150
100
25
20
15
INPUT VOLTAGE (V)
30
0
25 50 75 100
TEMPERATURE (°C)
3481 G04
3.5
LOAD CURRENT (A)
3.0
2.5
MINIMUM
2.0
VOUT = 5.0V
TA = 25°C
L = 4.7μH
f = 800kHz
20
15
INPUT VOLTAGE (V)
25
30
3481 G07
VOUT = 3.3V
TA = 25°C
L = 4.7μH
f = 800kHz
5
10
20
15
INPUT VOLTAGE (V)
25
30
3481 G06
Switch Current Limit
DUTY CYCLE = 10 %
4.0
3.0
2.5
2.0
3.0
2.5
DUTY CYCLE = 90 %
2.0
1.5
1.0
0.5
1.0
10
2.0
4.5
1.5
1.0
MINIMUM
1.0
125 150
SWITCH CURRENT LIMIT (A)
3.5
SWITCH CURRENT LIMIT(A)
4.0
5
2.5
Switch Current Limit
Maximum Load Current
1.5
3.0
3481 G05
4.0
TYPICAL
TYPICAL
1.5
0
–50 –25
35
3
Maximum Load Current
50
FRONT PAGE APPLICATION
10
1
2
2.5
1.5
SWITCHING FREQUENCY (MHz)
4.0
LOAD CURRENT (A)
300
SUPPLY CURRENT (μA)
SUPPLY CURRENT (μA)
60
40
0.5
3481 G03
CATCH DIODE: DIODES, INC. PDS360
350
50
VOUT = 3.3V
L = 10μH
LOAD = 1A
0
No Load Supply Current
TA = 25°C
5
65
50
10
400
70
0
70
3481 G02
No Load Supply Current
0
75
55
L: NEC PLC-0745-4R7
f: 800kHz
3481 G01
80
VIN = 24V
60
10
10
VIN = 12V
50
30
10
80
70
30
VIN = 12V
85
VIN = 7V
80
VIN = 24V
70
90
EFFICIENCY (%)
90
EFFICIENCY (%)
Efficiency
100
100
0
20
60
40
DUTY CYCLE (%)
80
100
3481 G08
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
3481 G09
3481fc
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LT3481
TYPICAL PERFORMANCE CHARACTERISTICS
Switch Voltage Drop
500
400
300
200
100
90
1.290
80
1.285
70
FEEDBACK VOLTAGE (V)
BOOST PIN CURRENT (mA)
VOLTAGE DROP (mV)
600
60
50
40
30
20
500
0
1000 1500 2000 2500 3000 3500
SWITCH CURRENT (mA)
500 1000 1500 2000 2500 3000 3500
SWITCH CURRENT (mA)
RT = 45.3k
1.00
0.95
0.90
125 150
Minimum Switch On-Time
140
1000
800
600
400
200
0.85
0
200
Soft-Start
400 600 800 1000 1200 1400
FB PIN VOLTAGE (mV)
80
60
40
20
RUN/SS Pin Current
2.5
2.0
1.5
1.0
Boost Diode
1.4
10
8
6
4
3
3.5
3481 G16
1.0
0.8
0.6
0.2
0
0
1.2
0.4
2
0.5
25 50 75 100 125 150
TEMPERATURE (˚C)
1.6
BOOST DIODE Vf (V)
RUN/SS PIN CURRENT (μA)
3.5
3.0
0
3481 G15
12
2.5
2
1.5
RUN/SS PIN VOLTAGE (V)
100
3481 G14
4.0
1
120
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
4381 G13
0.5
25 50 75 100
TEMPERATURE (°C)
4381 G12
MINIMUM SWITCH ON TIME (ns)
SWITCHING FREQUENCY (kHz)
1.05
0
0
RT = 45.3k
1.10
SWITCH CURRENT LIMIT (A)
1.260
1200
1.15
0
1.265
Frequency Foldback
Switching Frequency
0.80
–50 –25
1.270
3481 G11
3481 G10
1.20
1.275
1.250
–50 –25
0
0
1.280
1.255
10
0
FREQUENCY (MHz)
Feedback Voltage
BOOST Pin Current
700
0
5
20
30
15
25
10
RUN/SS PIN VOLTAGE (V)
35
3481 G17
0
0
1.0
0.5
1.5
BOOST DIODE CURRENT (A)
2.0
3481 G18
3481fc
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LT3481
TYPICAL PERFORMANCE CHARACTERISTICS
Error Amp Output Current
Minimum Input Voltage
Minimum Input Voltage
100
4.5
6.5
4.0
6.0
INPUT VOLTAGE (V)
VC PIN CURRENT (μA)
60
40
20
0
–20
–40
INPUT VOLTAGE (V)
80
3.5
3.0
VOUT = 3.3V
TA = 25°C
L = 4.7μH
f = 800kHz
2.5
–60
–80
1.065
1?.265
1.165
1.365
FB PIN VOLTAGE (V)
2.0
0.001
1.465
5.0
4.5
0.1
0.01
1
LOAD CURRENT (A)
4.0
0.001
10
2.50
THRESHOLD VOLTAGE (V)
1.50
1.00
SWITCHING THRESHOLD
0.50
10
Switching Waveforms;
Burst Mode Operation
1.200
CURRENT LIMIT CLAMP
0.1
0.01
1
LOAD CURRENT (A)
3481 G21
Power Good Threshold
VC Voltages
2.00
VOUT = 5.0V
TA = 25°C
L = 4.7μH
f = 800kHz
3481 G20
3481 G19
THRESHOLD VOLTAGE (V)
5.5
VIN = 12V; FRONT PAGE APPLICATION
ILOAD = 10mA
IL
0.5A/DIV
1.180
1.160
VSW
5V/DIV
1.140
VOUT
10mV/DIV
1.120
PG RISING
0
–50 –25
0
25
50 75 100 125 150
TEMPERATURE (°C)
1.100
–50 –25
3481 G22
0
25 50 75 100 125 150
TEMPERATURE (°C)
5μs/DIV
3481 G24
3481 G23
Switching Waveforms; Transition
from Burst Mode Operation to
Full Frequency
Switching Waveforms; Full
Frequency Continuous Operation
IL
0.5A/DIV
IL
0.5A/DIV
VRUN/SS
5V/DIV
VRUN/SS
5V/DIV
VOUT
10mV/DIV
VOUT
10mV/DIV
VIN = 12V; FRONT PAGE APPLICATION
ILOAD = 1A
VIN = 12V; FRONT PAGE APPLICATION
ILOAD = 140mA
1μs/DIV
3481 G25
1μs/DIV
3481 G26
3481fc
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LT3481
PIN FUNCTIONS
BD (Pin 1): This pin connects to the anode of the boost
Schottky diode.
BOOST (Pin 2): This pin is used to provide a drive
voltage, higher than the input voltage, to the internal bipolar
NPN power switch.
SW (Pin 3): The SW pin is the output of the internal power
switch. Connect this pin to the inductor, catch diode and
boost capacitor.
VIN (Pin 4): The VIN pin supplies current to the LT3481’s
internal regulator and to the internal power switch. This
pin must be locally bypassed.
RUN/SS (Pin 5): The RUN/SS pin is used to put the
LT3481 in shutdown mode. Tie to ground to shut down
the LT3481. Tie to 2.3V or more for normal operation. If
the shutdown feature is not used, tie this pin to the VIN
pin. RUN/SS also provides a soft-start function; see the
Applications Information section.
PG (Pin 6): The PG pin is the open-collector output of an
internal comparator. PG remains low until the FB pin is
within 10% of the final regulation voltage. PG output is
valid when VIN is above 3.5V and RUN/SS is high.
BIAS (Pin 7): The BIAS pin supplies the current to the
LT3481’s internal regulator. Tie this pin to the lowest
available voltage source above 3V (typically VOUT). This
architecture increases efficiency especially when the input
voltage is much higher than the output.
FB (Pin 8): The LT3481 regulates the FB pin to 1.265V.
Connect the feedback resistor divider tap to this pin.
VC (Pin 9): The VC pin is the output of the internal error
amplifier. The voltage on this pin controls the peak switch
current. Tie an RC network from this pin to ground to
compensate the control loop.
RT (Pin 10): Oscillator Resistor Input. Connecting a resistor
to ground from this pin sets the switching frequency.
Exposed Pad (Pin 11): Ground. The Exposed Pad must
be soldered to PCB.
BLOCK DIAGRAM
VIN
4
VIN
C1
7
5
10
BIAS
–
+
INTERNAL 1.265V REF
RUN/SS
3
SLOPE COMP
BD
SWITCH
LATCH
BOOST
1
2
C3
R
RT
OSCILLATOR
300kHz TO 2.8MHz
Q
S
SW
RT
DISABLE
L1
VOUT
3
C2
D1
SOFT-START
6
Burst Mode
DETECT
PG
ERROR AMP
+
–
+
–
1.12V
VC CLAMP
VC
9
CC
GND
11
FB
RC
CF
8
R2
R1
3481 BD
3481fc
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LT3481
OPERATION
The LT3481 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT,
enables an RS flip-flop, turning on the internal power
switch. An amplifier and comparator monitor the current
flowing between the VIN and SW pins, turning the switch
off when this current reaches a level determined by the
voltage at VC. An error amplifier measures the output
voltage through an external resistor divider tied to the FB
pin and servos the VC pin. If the error amplifier’s output
increases, more current is delivered to the output; if it
decreases, less current is delivered. An active clamp on the
VC pin provides current limit. The VC pin is also clamped to
the voltage on the RUN/SS pin; soft-start is implemented
by generating a voltage ramp at the RUN/SS pin using an
external resistor and capacitor.
An internal regulator provides power to the control circuitry. The bias regulator normally draws power from the
VIN pin, but if the BIAS pin is connected to an external
voltage higher than 3V bias power will be drawn from the
external source (typically the regulated output voltage).
This improves efficiency. The RUN/SS pin is used to place
the LT3481 in shutdown, disconnecting the output and
reducing the input current to less than 1μA.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate
the internal bipolar NPN power switch for efficient
operation.
To further optimize efficiency, the LT3481 automatically
switches to Burst Mode operation in light load situations.
Between bursts, all circuitry associated with controlling
the output switch is shut down reducing the input supply
current to 50μA in a typical application.
The oscillator reduces the LT3481’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the output current during startup
and overload.
The LT3481 contains a power good comparator which trips
when the FB pin is at 91% of its regulated value. The PG
output is an open-collector transistor that is off when the
output is in regulation, allowing an external resistor to pull
the PG pin high. Power good is valid when the LT3481 is
enabled and VIN is above 3.6V.
3481fc
8
LT3481
APPLICATIONS INFORMATION
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
⎛V
⎞
R1= R2 ⎜ OUT – 1⎟
⎝ 1.265 ⎠
Reference designators refer to the Block Diagram.
Setting the Switching Frequency
The LT3481 uses a constant frequency PWM architecture
that can be programmed to switch from 300kHz to 2.8MHz
by using a resistor tied from the RT pin to ground. A table
showing the necessary RT value for a desired switching
frequency is in Figure 1.
SWITCHING FREQUENCY (MHz)
RT VALUE (kΩ)
0.2
0.3
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
2.2
2.4
2.6
2.8
267
187
133
84.5
60.4
45.3
36.5
29.4
23.7
20.5
16.9
14.3
12.1
10.2
8.66
Figure 1. Switching Frequency vs RT Value
Operating Frequency Trade-Offs
Selection of the operating frequency is a trade-off between
efficiency, component size, minimum dropout voltage, and
maximum input voltage. The advantage of high frequency
operation is that smaller inductor and capacitor values may
be used. The disadvantages are lower efficiency, lower
maximum input voltage, and higher dropout voltage. The
highest acceptable switching frequency (fSW(MAX)) for a
given application can be calculated as follows:
fSW(MAX ) =
VD + VOUT
tON(MIN) ( VD + VIN – VSW )
where VIN is the typical input voltage, VOUT is the output
voltage, is the catch diode drop (~0.5V), VSW is the internal
switch drop (~0.5V at max load). This equation shows
that slower switching frequency is necessary to safely
accommodate high VIN/VOUT ratio. Also, as shown in
the next section, lower frequency allows a lower dropout
voltage. The reason input voltage range depends on the
switching frequency is because the LT3481 switch has
finite minimum on and off times. The switch can turn on
for a minimum of ~150ns and turn off for a minimum of
~150ns. This means that the minimum and maximum
duty cycles are:
DCMIN = fSW tON(MIN)
DCMAX = 1– fSW tOFF(MIN)
where fSW is the switching frequency, the tON(MIN) is the
minimum switch on-time (~150ns), and the tOFF(MIN) is
the minimum switch off-time (~150ns). These equations
show that duty cycle range increases when switching
frequency is decreased.
A good choice of switching frequency should allow adequate input voltage range (see next section) and keep
the inductor and capacitor values small.
Input Voltage Range
The maximum input voltage for LT3481 applications
depends on switching frequency, the absolute maximum
ratings on VIN and BOOST pins, and on operating mode.
If the output is in start-up or short-circuit operating modes,
then VIN must be below 34V and below the result of the
following equation:
VIN(MAX ) =
VOUT + VD
–V +V
fSW tON(MIN) D SW
where VIN(MAX) is the maximum operating input voltage,
VOUT is the output voltage, VD is the catch diode drop
(~0.5V), VSW is the internal switch drop (~0.5V at max
load), fSW is the switching frequency (set by RT), and
tON(MIN) is the minimum switch on-time (~150ns). Note that
a higher switching frequency will depress the maximum
operating input voltage. Conversely, a lower switching
3481fc
9
LT3481
APPLICATIONS INFORMATION
frequency will be necessary to achieve safe operation at
high input voltages.
If the output is in regulation and no short-circuit or start-up
events are expected, then input voltage transients of up to
36V are acceptable regardless of the switching frequency.
In this mode, the LT3481 may enter pulse-skipping operation where some switching pulses are skipped to maintain
output regulation. In this mode the output voltage ripple
and inductor current ripple will be higher than in normal
operation.
The minimum input voltage is determined by either the
LT3481’s minimum operating voltage of ~3.6V or by its
maximum duty cycle (see equation in previous section).
The minimum input voltage due to duty cycle is:
VIN(MIN) =
VOUT + VD
–V +V
1– fSW tOFF(MIN) D SW
where VIN(MIN) is the minimum input voltage, and tOFF(MIN)
is the minimum switch off-time (150ns). Note that higher
switching frequency will increase the minimum input
voltage. If a lower dropout voltage is desired, a lower
switching frequency should be used.
Inductor Selection
For a given input and output voltage, the inductor value
and switching frequency will determine the ripple current.
The ripple current ΔIL increases with higher VIN or VOUT
and decreases with higher inductance and faster switching frequency. A reasonable starting point for selecting
the ripple current is:
ΔIL = 0.4((IOUT(MAX))
where IOUT(MAX) is the maximum output load current. To
guarantee sufficient output current, peak inductor current
must be lower than the LT3481’s switch current limit (ILIM).
The peak inductor current is:
IL(PEAK) = IOUT(MAX) + ΔIL/2
where IL(PEAK) is the peak inductor current, IOUT(MAX) is
the maximum output load current, and ΔIL is the inductor
ripple current. The LT3481’s switch current limit (ILIM) is
at least 3.5A at low duty cycles and decreases linearly to
2.5A at DC = 0.8. The maximum output current is a function of the inductor ripple current:
IOUT(MAX) = ILIM – ΔIL/2
Be sure to pick an inductor ripple current that provides
sufficient maximum output current (IOUT(MAX)).
The largest inductor ripple current occurs at the highest
VIN. To guarantee that the ripple current stays below the
specified maximum, the inductor value should be chosen
according to the following equation:
⎛ VOUT + VD ⎞ ⎛ VOUT + VD ⎞
L=⎜
⎜ 1–
⎟
VIN(MAX ) ⎟⎠
⎝ fΔIL ⎟⎠ ⎜⎝
where VD is the voltage drop of the catch diode (~0.4V),
VIN(MAX) is the maximum input voltage, VOUT is the output
voltage, fSW is the switching frequency (set by RT), and
L is in the inductor value.
The inductor’s RMS current rating must be greater than the
maximum load current and its saturation current should be
about 30% higher. For robust operation in fault conditions
(start-up or short circuit) and high input voltage (>30V),
the saturation current should be above 3.5A. To keep the
efficiency high, the series resistance (DCR) should be less
than 0.1Ω, and the core material should be intended for
high frequency applications. Table 1 lists several vendors
and suitable types.
Table 1. Inductor Vendors
VENDOR
URL
PART SERIES
Murata
www.murata.com
LQH55D
Open
TDK
www.componenttdk.com
SLF7045
SLF10145
Shielded
Shielded
Toko
www.toko.com
D62CB
Shielded
D63CB
Shielded
D75C
Shielded
D75F
Open
CR54
Open
CDRH74
Shielded
CDRH6D38
Shielded
CR75
Open
Sumida
www.sumida.com
TYPE
3481fc
10
LT3481
APPLICATIONS INFORMATION
Of course, such a simple design guide will not always
result in the optimum inductor for your application. A larger
value inductor provides a slightly higher maximum load
current and will reduce the output voltage ripple. If your
load is lower than 2A, then you can decrease the value of
the inductor and operate with higher ripple current. This
allows you to use a physically smaller inductor, or one
with a lower DCR resulting in higher efficiency. There are
several graphs in the Typical Performance Characteristics
section of this data sheet that show the maximum load
current as a function of input voltage and inductor value
for several popular output voltages. Low inductance may
result in discontinuous mode operation, which is okay
but further reduces maximum load current. For details
of maximum output current and discontinuous mode
operation, see Linear Technology Application Note 44.
Finally, for duty cycles greater than 50% (VOUT/VIN >
0.5), there is a minimum inductance required to avoid
subharmonic oscillations. See AN19.
Input Capacitor
Bypass the input of the LT3481 circuit with a ceramic capacitor of X7R or X5R type. Y5V types have poor performance
over temperature and applied voltage, and should not be
used. A 4.7μF to 10μF ceramic capacitor is adequate to
bypass the LT3481 and will easily handle the ripple current.
Note that larger input capacitance is required when a lower
switching frequency is used. If the input power source has
high impedance, or there is significant inductance due to
long wires or cables, additional bulk capacitance may be
necessary. This can be provided with a low performance
electrolytic capacitor.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT3481 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 4.7μF capacitor is capable of this task, but only if it is
placed close to the LT3481 and the catch diode (see the
PCB Layout section). A second precaution regarding the
ceramic input capacitor concerns the maximum input
voltage rating of the LT3481. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (under damped) tank circuit. If the LT3481 circuit
is plugged into a live supply, the input voltage can ring to
twice its nominal value, possibly exceeding the LT3481’s
voltage rating. This situation is easily avoided (see the Hot
Plugging Safety section).
For space sensitive applications, a 2.2μF ceramic capacitor can be used for local bypassing of the LT3481 input.
However, the lower input capacitance will result in increased input current ripple and input voltage ripple, and
may couple noise into other circuitry. Also, the increased
voltage ripple will raise the minimum operating voltage
of the LT3481 to ~3.7V.
Output Capacitor and Output Ripple
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by the
LT3481 to produce the DC output. In this role it determines
the output ripple, and low impedance at the switching
frequency is important. The second function is to store
energy in order to satisfy transient loads and stabilize the
LT3481’s control loop. Ceramic capacitors have very low
equivalent series resistance (ESR) and provide the best
ripple performance. A good starting value is:
COUT =
100
VOUT fSW
where fSW is in MHz, and COUT is the recommended
output capacitance in μF. Use X5R or X7R types. This
choice will provide low output ripple and good transient
response. Transient performance can be improved with
a higher value capacitor if the compensation network is
also adjusted to maintain the loop bandwidth. A lower
value of output capacitor can be used to save space
and cost but transient performance will suffer. See the
Frequency Compensation section to choose an appropriate
compensation network.
3481fc
11
LT3481
APPLICATIONS INFORMATION
Table 2. Capacitor Vendors
VENDOR
PHONE
URL
PART SERIES
Panasonic
(714) 373-7366
www.panasonic.com
Ceramic,
COMMANDS
Polymer,
EEF Series
Tantalum
Kemet
(864) 963-6300
www.kemet.com
Ceramic,
Tantalum
Sanyo
(408) 749-9714
www.sanyovideo.com
T494, T495
Ceramic,
Polymer,
POSCAP
Tantalum
Murata
(408) 436-1300
AVX
www.murata.com
Ceramic
www.avxcorp.com
Ceramic,
Tantalum
Taiyo Yuden
(864) 963-6300
www.taiyo-yuden.com
When choosing a capacitor, look carefully through the
data sheet to find out what the actual capacitance is under
operating conditions (applied voltage and temperature).
A physically larger capacitor, or one with a higher voltage
rating, may be required. High performance tantalum or
electrolytic capacitors can be used for the output capacitor.
Low ESR is important, so choose one that is intended for
use in switching regulators. The ESR should be specified
by the supplier, and should be 0.05Ω or less. Such a
capacitor will be larger than a ceramic capacitor and will
have a larger capacitance, because the capacitor must be
large to achieve low ESR. Table 2 lists several capacitor
vendors.
TPS Series
Ceramic
Table 3. Diode Vendors
PART NUMBER
VR
(V)
IAVE
(A)
VF AT 1A
(mV)
VF AT 2A
(mV)
On Semicnductor
MBRM120E
MBRM140
20
40
1
1
530
550
595
Diodes Inc.
B120
B130
B220
B230
DFLS240L
20
30
20
30
40
1
1
2
2
2
500
500
International Rectifier
10BQ030
20BQ030
30
30
1
2
420
500
500
500
470
470
Catch Diode
Ceramic Capacitors
The catch diode conducts current only during switch off
time. Average forward current in normal operation can
be calculated from:
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT3481 due to their piezoelectric nature.
When in Burst Mode operation, the LT3481’s switching
frequency depends on the load current, and at very light
loads the LT3481 can excite the ceramic capacitor at audio
frequencies, generating audible noise. Since the LT3481
operates at a lower current limit during Burst Mode operation, the noise is typically very quiet to a casual ear.
If this is unacceptable, use a high performance tantalum
or electrolytic capacitor at the output.
ID(AVG) = IOUT (VIN – VOUT)/VIN
where IOUT is the output load current. The only reason to
consider a diode with a larger current rating than necessary
for nominal operation is for the worst-case condition of
shorted output. The diode current will then increase to the
typical peak switch current. Peak reverse voltage is equal
to the regulator input voltage. Use a diode with a reverse
voltage rating greater than the input voltage. Table 3 lists
several Schottky diodes and their manufacturers.
3481fc
12
LT3481
APPLICATIONS INFORMATION
The LT3481 uses current mode control to regulate the
output. This simplifies loop compensation. In particular, the
LT3481 does not require the ESR of the output capacitor
for stability, so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size. Frequency
compensation is provided by the components tied to the
VC pin, as shown in Figure 2. Generally a capacitor (CC)
and a resistor (RC) in series to ground are used. In addition, there may be lower value capacitor in parallel. This
capacitor (CF) is not part of the loop compensation but
is used to filter noise at the switching frequency, and is
required only if a phase-lead capacitor is used or if the
output capacitor has high ESR.
Loop compensation determines the stability and transient
performance. Designing the compensation network is
a bit complicated and the best values depend on the
application and in particular the type of output capacitor.
A practical approach is to start with one of the circuits in
this data sheet that is similar to your application and tune
the compensation network to optimize the performance.
Stability should then be checked across all operating
conditions, including load current, input voltage and
temperature. The LT1375 data sheet contains a more
thorough discussion of loop compensation and describes
how to test the stability using a transient load. Figure 2
shows an equivalent circuit for the LT3481 control loop.
The error amplifier is a transconductance amplifier with
finite output impedance. The power section, consisting
of the modulator, power switch and inductor, is modeled
as a transconductance amplifier generating an output
current proportional to the voltage at the VC pin. Note that
the output capacitor integrates this current, and that the
LT3481
CURRENT MODE
POWER STAGE
gm = 3.5mho
SW
ERROR
AMPLIFIER
OUTPUT
R1
FB
+
Frequency Compensation
capacitor on the VC pin (CC) integrates the error amplifier
output current, resulting in two poles in the loop. In most
cases a zero is required and comes from either the output
capacitor ESR or from a resistor RC in series with CC.
This simple model works well as long as the value of the
inductor is not too high and the loop crossover frequency
is much lower than the switching frequency. A phase lead
capacitor (CPL) across the feedback divider may improve
the transient response. Figure 3 shows the transient
response when the load current is stepped from 500mA
to 1500mA and back to 500mA.
–
A final precaution regarding ceramic capacitors concerns
the maximum input voltage rating of the LT3481. A ceramic
input capacitor combined with trace or cable inductance
forms a high quality (under damped) tank circuit. If the
LT3481 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding
the LT3481’s rating. This situation is easily avoided (see
the Hot Plugging Safely section).
1.265V
CPL
gm =
330μmho
ESR
C1
+
3Meg
C1
VC
CF
POLYMER
OR
TANTALUM
GND
RC
CERAMIC
R2
CC
3481 F02
Figure 2. Model for Loop Response
VOUT = 12V; FRONT PAGE APPLICATION
IL
1A/DIV
VOUT
100mV/DIV
10μs/DIV
3481 F03
Figure 3. Transient Load Response of the LT3481 Front Page
Application as the Load Current is Stepped from 500mA to
1500mA. VOUT = 3.3V
3481fc
13
LT3481
APPLICATIONS INFORMATION
Burst Mode Operation
To enhance efficiency at light loads, the LT3481 automatically switches to Burst Mode operation which keeps
the output capacitor charged to the proper voltage while
minimizing the input quiescent current. During Burst Mode
operation, the LT3481 delivers single cycle bursts of current
to the output capacitor followed by sleep periods where
the output power is delivered to the load by the output
capacitor. In addition, VIN and BIAS quiescent currents are
reduced to typically 20μA and 50μA respectively during
the sleep time. As the load current decreases towards a
no load condition, the percentage of time that the LT3481
operates in sleep mode increases and the average input
current is greatly reduced resulting in higher efficiency.
See Figure 4.
boost diode can be tied to the input (Figure 5c), or to
another supply greater than 2.8V. The circuit in Figure 5a
is more efficient because the BOOST pin current and BIAS
pin quiescent current comes from a lower voltage source.
You must also be sure that the maximum voltage ratings
of the BOOST and BIAS pins are not exceeded.
VOUT
BD
BOOST
VIN
VIN
LT3481
GND
4.7μF
C3
SW
(5a) For VOUT > 2.8V
VOUT
VIN = 12V; FRONT PAGE APPLICATION
ILOAD = 10mA
D2
BD
IL
0.5A/DIV
BOOST
VIN
VIN
GND
4.7μF
VSW
5V/DIV
VOUT
10mV/DIV
LT3481
C3
SW
(5b) For 2.5V < VOUT < 2.8V
5μs/DIV
VOUT
3481 F04
BD
Figure 4. Burst Mode Operation
BOOST
VIN
BOOST and BIAS Pin Considerations
Capacitor C3 and the internal boost Schottky diode (see
the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases
a 0.22μF capacitor will work well. Figure 2 shows three
ways to arrange the boost circuit. The BOOST pin must be
more than 2.3V above the SW pin for best efficiency. For
outputs of 3V and above, the standard circuit (Figure 5a)
is best. For outputs between 2.8V and 3V, use a 1μF boost
capacitor. A 2.5V output presents a special case because it
is marginally adequate to support the boosted drive stage
while using the internal boost diode. For reliable BOOST pin
operation with 2.5V outputs use a good external Schottky
diode (such as the ON Semi MBR0540), and a 1μF boost
capacitor (see Figure 5b). For lower output voltages the
4.7μF
VIN
LT3481
GND
C3
SW
(5c) For VOUT < 2.5V
3481 FO5
Figure 5. Three Circuits For Generating The Boost Voltage
The minimum operating voltage of an LT3481 application
is limited by the minimum input voltage (3.6V) and by the
maximum duty cycle as outlined in a previous section. For
proper startup, the minimum input voltage is also limited
by the boost circuit. If the input voltage is ramped slowly,
or the LT3481 is turned on with its RUN/SS pin when the
output is already in regulation, then the boost capacitor
3481fc
14
LT3481
APPLICATIONS INFORMATION
may not be fully charged. Because the boost capacitor is
charged with the energy stored in the inductor, the circuit
will rely on some minimum load current to get the boost
circuit running properly. This minimum load will depend
on input and output voltages, and on the arrangement of
the boost circuit. The minimum load generally goes to
zero once the circuit has started. Figure 6 shows a plot
of minimum load to start and to run as a function of input
voltage. In many cases the discharged output capacitor
will present a load to the switcher, which will allow it to
start. The plots show the worst-case situation where VIN
is ramping very slowly. For lower start-up voltage, the
boost diode can be tied to VIN; however, this restricts the
input range to one-half of the absolute maximum rating
of the BOOST pin.
At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
300mV above VOUT. At higher load currents, the inductor
current is continuous and the duty cycle is limited by the
maximum duty cycle of the LT3481, requiring a higher
input voltage to maintain regulation.
Soft-Start
The RUN/SS pin can be used to soft-start the LT3481,
reducing the maximum input current during start-up.
The RUN/SS pin is driven through an external RC filter to
create a voltage ramp at this pin. Figure 7 shows the startup and shutdown waveforms with the soft-start circuit.
By choosing a large RC time constant, the peak start-up
current can be reduced to the current that is required to
regulate the output, with no overshoot. Choose the value
of the resistor so that it can supply 20μA when the RUN/SS
pin reaches 2.3V.
IL
1A/DIV
RUN
15k
RUN/SS
0.22μF
VRUN/SS
2V/DIV
GND
VOUT
2V/DIV
2ms/DIV
3481 F07
Figure 7. To Soft-Start the LT3481, Add a Resisitor
and Capacitor to the RUN/SS Pin
6.0
TO START
Synchronization
INPUT VOLTAGE (V)
5.5
5.0
4.5
4.0
TO RUN
3.5
VOUT = 3.3V
TA = 25°C
L = 4.7μ
f = 800 kHz
3.0
2.5
2.0
0.001
0.1
0.01
1
LOAD CURRENT (A)
10
The LT3481 should not be synchronized until its output
is near regulation as indicated by the PG flag. This can be
done with the system microcontroller/microprocessor or
with a discrete circuit by using the PG output. If a sync
signal is applied while the PG is low, the LT3481 may
exhibit erratic operation. See Typical Applications.
8.0
TO START
INPUT VOLTAGE (V)
7.0
6.0
5.0
TO RUN
4.0
3.0
2.0
0.001
VOUT = 5.0V
TA = 25°C
L = 4.7μ
f = 800 kHz
0.1
0.01
1
LOAD CURRENT (A)
The internal oscillator of the LT3481 can be synchronized
to an external 275kHz to 475kHz clock by using a 5pF
to 20pF capacitor to connect the clock signal to the RT
pin. The resistor tying the RT pin to ground should be
chosen such that the LT3481 oscillates 20% lower than
the intended synchronization frequency (see Setting the
Switching Frequency section).
10
3481 F06
Figure 6. The Minimum Input Voltage Depends on
Output Voltage, Load Current and Boost Circuit
When applying a sync signal, positive clock transitions
reset LT3481’s internal clock and negative transitions
initiate a switch cycle. The amplitude of the sync signal
must be at least 2V. The sync signal duty cycle can range
3481fc
15
LT3481
APPLICATIONS INFORMATION
from 5% up to a maximum value given by the following
equation:
⎛
VOUT + VD ⎞
DCSYNC(MAX ) = ⎜ 1 –
– f • 600ns
VIN – VSW + VD ⎟⎠ SW
⎝
where VOUT is the programmed output voltage, VD is the
diode forward drop, VIN is the typical input voltage, VSW
is the switch drop, and fSW is the desired switching frequency. For example, a 24V input to 5V output at 300kHz
can be synchronized to a square wave with a maximum
duty cycle of 60%. For some applications, such as 12VIN
to 5VOUT at 350kHz, the maximum allowable sync duty
cycle will be less than 50%. If a low duty cycle clock cannot be obtained from the system, then a one-shot should
be used between the sync signal and the LT3481. See
Typical Applications.
The value of the coupling capacitor which connects the
clock signal to the RT pin should be chosen based on the
clock signal amplitude. Good starting values for 3.3V and
5V clock signals are 10pF and 5pF, respectively. These
values should be tested and adjusted for each individual
application to assure reliable operation.
Caution should be used when synchronizing more than
50% above the initial switching frequency (as set by
the RT resistor) because at higher clock frequencies the
amplitude of the internal slope compensation used to
prevent subharmonic switching is reduced. This type of
subharmonic switching only occurs at input voltages less
than twice output voltage. Higher inductor values will tend
to reduce this problem.
Shorted and Reversed Input Protection
If the inductor is chosen so that it won’t saturate excessively, an LT3481 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT3481 is absent. This may occur in battery charging applications or in battery backup systems where a battery
or some other supply is diode ORed with the LT3481’s
output. If the VIN pin is allowed to float and the RUN/SS
pin is held high (either by a logic signal or because it is
tied to VIN), then the LT3481’s internal circuitry will pull
its quiescent current through its SW pin. This is fine if
16
your system can tolerate a few mA in this state. If you
ground the RUN/SS pin, the SW pin current will drop to
essentially zero. However, if the VIN pin is grounded while
the output is held high, then parasitic diodes inside the
LT3481 can pull large currents from the output through
the SW pin and the VIN pin. Figure 8 shows a circuit that
will run only when the input voltage is present and that
protects against a shorted or reversed input.
D4
MBRS140
VIN
VIN
BOOST
LT3481
RUN/SS
VOUT
SW
VC
GND FB
BACKUP
3481 F08
Figure 8. Diode D4 Prevents a Shorted Input from
Discharging a Backup Battery Tied to the Output. It Also
Protects the Circuit from a Reversed Input. The LT3481
Runs Only When the Input is Present
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 9 shows
the recommended component placement with trace,
ground plane and via locations. Note that large, switched
currents flow in the LT3481’s VIN and SW pins, the catch
diode (D1) and the input capacitor (C1). The loop formed
by these components should be as small as possible. These
components, along with the inductor and output capacitor,
should be placed on the same side of the circuit board,
and their connections should be made on that layer. Place
a local, unbroken ground plane below these components.
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB and VC nodes small so that the ground
traces will shield them from the SW and BOOST nodes.
The Exposed Pad on the bottom of the package must be
soldered to ground so that the pad acts as a heat sink. To
keep thermal resistance low, extend the ground plane as
much as possible, and add thermal vias under and near
the LT3481 to additional ground planes within the circuit
board and on the bottom side.
3481fc
LT3481
APPLICATIONS INFORMATION
Hot Plugging Safely
L1
C2
VOUT
CC
RRT
RC
R2
R1
D1
C1
RPG
GND
3481 F09
VIAS TO LOCAL GROUND PLANE
VIAS TO RUN/SS
VIAS TO VOUT
VIAS TO PG
VIAS TO VIN
OUTLINE OF LOCAL
GROUND PLANE
Figure 9. A Good PCB Layout Ensures Proper, Low EMI Operation
CLOSING SWITCH
SIMULATES HOT PLUG
IIN
VIN
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT3481 circuits. However, these capacitors can cause problems if the LT3481 is plugged into a
live supply (see Linear Technology Application Note 88 for
a complete discussion). The low loss ceramic capacitor,
combined with stray inductance in series with the power
source, forms an under damped tank circuit, and the
voltage at the VIN pin of the LT3481 can ring to twice the
nominal input voltage, possibly exceeding the LT3481’s
rating and damaging the part. If the input supply is poorly
controlled or the user will be plugging the LT3481 into an
energized supply, the input network should be designed to
prevent this overshoot. Figure 10 shows the waveforms
that result when an LT3481 circuit is connected to a 24V
supply through six feet of 24-gauge twisted pair. The
DANGER
VIN
20V/DIV
RINGING VIN MAY EXCEED
ABSOLUTE MAXIMUM RATING
LT3481
+
4.7μF
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
IIN
10A/DIV
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
20μs/DIV
(10a)
0.7Ω
LT3481
VIN
20V/DIV
+
0.1μF
4.7μF
IIN
10A/DIV
(10b)
LT3481
+
22μF
35V
AI.EI.
20μs/DIV
VIN
20V/DIV
+
4.7μF
IIN
10A/DIV
(10c)
20μs/DIV
3481 F10
Figure 10. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation when the LT3481 is Connected to a Live Supply
3481fc
17
LT3481
APPLICATIONS INFORMATION
first plot is the response with a 4.7μF ceramic capacitor
at the input. The input voltage rings as high as 50V and
the input current peaks at 26A. A good solution is shown
in Figure 10b. A 0.7Ω resistor is added in series with the
input to eliminate the voltage overshoot (it also reduces
the peak input current). A 0.1μF capacitor improves high
frequency filtering. For high input voltages its impact on
efficiency is minor, reducing efficiency by 1.5 percent for
a 5V output at full load operating from 24V.
High Temperature Considerations
The PCB must provide heat sinking to keep the LT3481
cool. The Exposed Pad on the bottom of the package must
be soldered to a ground plane. This ground should be tied
to large copper layers below with thermal vias; these layers will spread the heat dissipated by the LT3481. Place
additional vias can reduce thermal resistance further. With
these steps, the thermal resistance from die (or junction)
to ambient can be reduced to θJA = 35°C/W or less. With
100LFPM airflow, this resistance can fall by another 25%.
Further increases in airflow will lead to lower thermal resistance. Because of the large output current capability of
the LT3481, it is possible to dissipate enough heat to raise
the junction temperature beyond the absolute maximum
of 125°C (150°C for the H grade). When operating at high
ambient temperatures, the maximum load current should
be derated as the ambient temperature approaches 125°C
(150°C for the H grade).
Power dissipation within the LT3481 can be estimated
by calculating the total power loss from an efficiency
measurement and subtracting the catch diode loss. The
die temperature is calculated by multiplying the LT3481
power dissipation by the thermal resistance from junction
to ambient.
Other Linear Technology Publications
Application Notes 19, 35 and 44 contain more detailed
descriptions and design information for buck regulators
and other switching regulators. The LT1376 data sheet
has a more extensive discussion of output ripple, loop
compensation and stability testing. Design Note 100
shows how to generate a bipolar output supply using a
buck regulator.
TYPICAL APPLICATIONS
5V Step-Down Converter
VOUT
5V
2A
VIN
6.3V TO 34V
BD
VIN
ON OFF
RUN/SS
BOOST
0.47μF
VC
4.7μF
LT3481
SW
D
RT
20k
L
6.8μH
BIAS
PG
590k
60.4k
GND
330pF
f = 800kHz
FB
22μF
200k
3481 TA02
D: DIODES INC. DFLS240L
L: TAIYO YUDEN NP06DZB6R8M
3481fc
18
LT3481
TYPICAL APPLICATIONS
3.3V Step-Down Converter
VOUT
3.3V
2A
VIN
4.4V TO 34V
VIN
BD
RUN/SS
ON OFF
BOOST
L
4.7μH
0.47μF
VC
4.7μF
SW
LT3481
D
RT
16.2k
PG
BIAS
324k
60.4k
FB
GND
330pF
22μF
200k
f = 800kHz
3481 TA03
D: DIODES INC. DFLS240L
L: TAIYO YUDEN NP06DZB4R7M
2.5V Step-Down Converter
VOUT
2.5V
2A
VIN
4V TO 34V
VIN
BD
RUN/SS
ON OFF
D2
BOOST
L
4.7μH
1μF
VC
4.7μF
SW
LT3481
D1
RT
22.1k
BIAS
PG
196k
84.5k
FB
GND
220pF
47μF
200k
f = 600kHz
3481 TA04
D1: DIODES INC. DFLS240L
D2: MBR0540
L: TAIYO YUDEN NP06DZB4R7M
5V, 2MHz Step-Down Converter
VOUT
5V
2A
VIN
8.6V TO 22V
TRANSIENT TO 36V
BD
VIN
ON OFF
RUN/SS
BOOST
0.47μF
VC
2.2μF
LT3481
SW
D
RT
20k
L
2.2μH
BIAS
PG
590k
16.9k
GND
330pF
f = 2MHz
FB
10μF
200k
3481 TA05
D: DIODES INC. DFLS240L
L: SUMIDA CDRM4D22/HP-2R2
3481fc
19
LT3481
TYPICAL APPLICATIONS
12V Step-Down Converter
VOUT
12V
2A
VIN
15V TO 34V
VIN
BD
RUN/SS
ON OFF
BOOST
0.47μF
VC
10μF
SW
LT3481
D
RT
30k
L
10μH
PG
BIAS
845k
60.4k
FB
GND
330pF
22μF
100k
f = 800kHz
3481 TA06
D: DIODES INC. DFLS240L
L: NEC/TOKIN PLC-0755-100
5V Step-Down Converter with Sync Input
VOUT
5V
2A
VIN
20V TO 34V
BD
VIN
4.7μF
NOTE: DO NOT APPLY SYNC
SIGNAL UNTIL PGOOD GOES HIGH
RUN/SS
ON OFF
BOOST
L
8.2μH
0.47μF
VC
8.2pF
RT
SYNC IN
SW
D
LT3481
3.3V SQ WAVE 300kHz TO 375kHz
PGOOD
BIAS
PG
11.8k
75pF
100k
29.4k
226k
1000pF
VOUT
f = 250kHz
GND
FB
47μF
10k
3481 TA07
D: DIODES INC. DFLS240L
L: NEC/TOKIN PLC-0755-8R2
3481fc
20
LT3481
TYPICAL APPLICATIONS
5V Step-Down Converter with Sync and One-Shot
VOUT
5V
2A
VIN
8V TO 34V
BD
VIN
4.7μF
RUN/SS
ON OFF
BOOST
L
15μH
0.47μF
VC
15pF
1k
SYNC IN
SW
RT
3V SQ WAVE
Hz TO 450kHz
LT3481
AND
BIAS
PG
25k
11.8k
75pF
100k
Q1
29.4k
133k
25k
D
50pF
VOUT
1000pF
FB
GND
47μF
10k
f = 300kHz
3481 TA08
D: DIODES INC. DFLS240L
L: NEC/TOKIN PLC-0755-150
Q1: ON SEMI MMBT3904
1.8V Step-Down Converter
VIN
3.5V TO 27V
VIN
ON OFF
BD
RUN/SS
BOOST
0.47μF
VC
4.7μF
LT3481
SW
D
RT
15.4k
L
3.3μH
PG
VOUT
1.8V
2A
BIAS
84.5k
105k
GND
330pF
f = 500kHz
FB
47μF
200k
3481 TA09
D: DIODES INC. DFLS240L
L: TAIYO YUDEN NP06DZB3R3M
3481fc
21
LT3481
PACKAGE DESCRIPTION
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1699 Rev B)
0.70 p0.05
3.55 p0.05
1.65 p0.05
2.15 p0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 p 0.05
0.50
BSC
2.38 p0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
3.00 p0.10
(4 SIDES)
R = 0.125
TYP
0.40 p 0.10
6
10
5
1
1.65 p 0.10
(2 SIDES)
PIN 1
TOP MARK
(SEE NOTE 6)
0.200 REF
(DD) DFN REV B 0309
0.25 p 0.05
0.50 BSC
0.75 p0.05
0.00 – 0.05
2.38 p0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
3481fc
22
LT3481
PACKAGE DESCRIPTION
MSE Package
10-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1664 Rev C)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.794 p 0.102
(.110 p .004)
5.23
(.206)
MIN
0.889 p 0.127
(.035 p .005)
1
0.05 REF
10
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
3.00 p 0.102
(.118 p .004)
(NOTE 3)
10 9 8 7 6
DETAIL “A”
0o – 6o TYP
1 2 3 4 5
GAUGE PLANE
0.53 p 0.152
(.021 p .006)
DETAIL “A”
0.18
(.007)
0.497 p 0.076
(.0196 p .003)
REF
3.00 p 0.102
(.118 p .004)
(NOTE 4)
4.90 p 0.152
(.193 p .006)
0.254
(.010)
0.29
REF
1.83 p 0.102
(.072 p .004)
2.083 p 0.102 3.20 – 3.45
(.082 p .004) (.126 – .136)
0.50
0.305 p 0.038
(.0197)
(.0120 p .0015)
BSC
TYP
RECOMMENDED SOLDER PAD LAYOUT
2.06 p 0.102
(.081 p .004)
SEATING
PLANE
1.10
(.043)
MAX
0.17 – 0.27
(.007 – .011)
TYP
0.50
(.0197)
NOTE:
BSC
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.86
(.034)
REF
0.1016 p 0.0508
(.004 p .002)
MSOP (MSE) 0908 REV C
3481fc
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT3481
TYPICAL APPLICATION
1.265V Step-Down Converter
VIN
3.6V TO 27V
BD
VIN
RUN/SS
ON OFF
BOOST
0.47μF
VC
4.7μF
LT3481
SW
D
RT
13k
VOUT
1.265V
2A
BIAS
PG
105k
L
3.3μH
GND
FB
330pF
47μF
f = 500kHz
3481 TA10
D: DIODES INC. DFLS240L
L: TAIYO YUDEN NP06DZB3R3M
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1933
500mA (IOUT), 500kHz Step-Down Switching
Regulator in SOT-23
VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.6mA, ISD
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