LT3493-3
1.2A, 750kHz Step-Down
Switching Regulator in
2mm × 3mm DFN
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FEATURES
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DESCRIPTIO
The LT®3493-3 is a current mode PWM step-down DC/DC
converter with an internal 1.75A power switch. The wide
operating input range of 6.8V to 36V (40V maximum) makes
the LT3493-3 ideal for regulating power from a wide variety of sources, including unregulated wall transformers,
24V industrial supplies and automotive batteries. Its high
operating frequency allows the use of tiny, low cost inductors and ceramic capacitors, resulting in low, predictable
output ripple.
Wide Input Range: 6.8V to 36V Operating,
40V Maximum
1.2A Output Current
Fixed Frequency Operation: 750kHz
Output Adjustable Down to 780mV
Short-Circuit Robust
Uses Tiny Capacitors and Inductors
Soft-Start
Internally Compensated
Low Shutdown Current: 8V
SW
SHDN
32.4k
GND
1µF
FB
22pF
10µF
10k
85
80
EFFICIENCY (%)
VIN
6.8V TO 36V
75
70
65
60
3493-3 TA01a
VIN = 12V
VOUT = 3.3V
L = 10µH
55
50
0
0.2
0.8
0.4
0.6
LOAD CURRENT (A)
1.0
1.2
3493-3 TA01b
3493-3f
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LT3493-3
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ABSOLUTE
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PACKAGE/ORDER I FOR ATIO
(Note 1)
Input Voltage (VIN) .................................................. 40V
BOOST Pin Voltage .................................................. 50V
BOOST Pin Above SW Pin ....................................... 25V
SHDN Pin ................................................................ 40V
FB Voltage ................................................................. 6V
Operating Temperature Range (Note 2)
LT3493E-3 ........................................... –40°C to 85°C
Maximum Junction Temperature .......................... 125°C
Storage Temperature Range ................. – 65°C to 150°C
TOP VIEW
6 SHDN
FB 1
7
GND 2
5 VIN
4 SW
BOOST 3
DCB PACKAGE
6-LEAD (2mm × 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 64°C/ W
EXPOSED PAD (PIN 7) IS GND, MUST BE SOLDERED TO PCB
ORDER PART NUMBER
DCB PART MARKING
LT3493EDCB-3
LCGJ
Order Options Tape and Reel: Add #TR Lead Free: Add #PBF
Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 12V, VBOOST = 17V, unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
MIN
VIN Operating Range
6.8
Undervoltage Lockout
6.2
765
●
Feedback Voltage
FB Pin Bias Current
VFB = Measured VREF (Note 4)
Quiescent Current
Not Switching
●
TYP
V
6.5
6.8
V
780
795
mV
50
150
nA
1.9
2.5
mA
2
VSHDN = 0V
0.01
Reference Line Regulation
VIN = 6.8V to 36V
0.007
Switching Frequency
VFB = 0.7V
VFB = 0V
●
TA = 25°C
UNITS
36
Quiescent Current in Shutdown
Maximum Duty Cycle
MAX
685
750
36
88
91
95
95
µA
%/V
815
kHz
kHz
%
%
3493-3f
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LT3493-3
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 12V, VBOOST = 17V, unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
MIN
TYP
MAX
Switch Current Limit
(Note 3)
1.4
1.75
2.2
Switch VCESAT
ISW = 1A
330
UNITS
A
mV
Switch Leakage Current
2
µA
Minimum Boost Voltage Above Switch
ISW = 1A
1.85
2.2
V
BOOST Pin Current
ISW = 1A
30
50
mA
SHDN Input Voltage High
2.3
V
SHDN Input Voltage Low
SHDN Bias Current
VSHDN = 2.3V (Note 5)
VSHDN = 0V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3493E-3 is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
6
0.01
V
15
0.1
µA
µA
Note 3: Current limit guaranteed by design and/or correlation to static test.
Slope compensation reduces current limit at higher duty cycle.
Note 4: Current flows out of pin.
Note 5: Current flows into pin.
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TYPICAL PERFOR A CE CHARACTERISTICS
TA = 25°C unless otherwise noted.
Efficiency (VOUT = 5V, L = 10µH)
Efficiency (VOUT = 3.3V, L = 10µH)
95
90
90
85
85
80
80
EFFICIENCY (%)
EFFICIENCY (%)
0.3
75
70
65
75
70
65
60
60
VIN = 8V
VIN = 12V
VIN = 24V
55
50
0
0.2
0.4
0.6
0.8
LOAD CURRENT (A)
1.0
1.2
3493-3 G01
VIN = 8V
VIN = 12V
VIN = 24V
55
50
0
0.2
0.4
0.6
0.8
LOAD CURRENT (A)
1.0
1.2
3493-3 G02
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TYPICAL PERFOR A CE CHARACTERISTICS
Maximum Load Current,
VOUT = 5V, L = 8.2µH
Maximum Load Current,
VOUT = 5V, L = 33µH
1.60
1.60
1.50
1.50
1.40
1.30
1.20
MINIMUM
1.10
1.00
OUTPUT CURRENT (A)
TYPICAL
OUTPUT CURRENT (A)
1.40
1.30
MINIMUM
1.20
1.10
0.90
8
12
16
20
VIN (V)
28
8
12
16
20
VIN (V)
24
5
28
1.50
TYPICAL
7.0
500
6.9
VCE(SW) (mV)
MINIMUM
1.10
1.00
350
300
TA = –40°C
250
200
6.2
50
6.1
0
6.0
–50
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8
SWITCH CURRENT (A)
700
740
720
700
680
660
640
1.8
600
500
400
300
200
0
75
100
3493-3 G09
100
Soft-Start
100
620
75
2.0
SWITCH CURRENT LIMIT (A)
780
SWITCHING FREQUENCY (kHz)
800
0
25
50
TEMPERATURE (°C)
3493-3 G07
Frequency Foldback
Switching Frequency
760
–25
3493-3 G06
800
0
25
50
TEMPERATURE (°C)
6.4
100
3493-3 G21
–25
6.5
6.3
VIN (V)
600
–50
6.6
150
30
25
30
25
6.7
TA = 25°C
0
0.90
20
20
6.8
TA = 85°C
400
1.40
1.20
15
Undervoltage Lockout
550
450
1.30
10
3493-3 G05
Switch Voltage Drop
1.60
15
MINIMUM
1.10
3493-3 G22
Maximum Load Current,
VOUT = 3.3V, L = 10µH
10
1.20
VIN (V)
3493-3 G04
5
1.30
0.90
0.90
24
TYPICAL
1.40
1.00
1.00
UVLO (V)
OUTPUT CURRENT (A)
TYPICAL
OUTPUT CURRENT (A)
Maximum Load Current,
VOUT = 3.3V, L = 4.7µH
1.60
1.50
FREQUENCY (kHz)
TA = 25°C unless otherwise noted.
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
0
100 200 300 400 500 600 700 800
FEEDBACK VOLTAGE (mV)
3493-3 G11
0
0.25 0.50 0.75 1 1.25 1.50 1.75
SHDN PIN VOLTAGE (V)
2
3493-3 G13
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TYPICAL PERFOR A CE CHARACTERISTICS
SHDN Pin Current
Switch Current Limit
Switch Current Limit
2.0
45
1.9
1.8
40
1.8
1.6
30
25
20
15
10
1.7
1.6
1.5
1.4
1.3
1.2
5
1.1
0
1.0
–50
0
2
4
6
8 10 12 14 16 18 20
VSHDN (V)
SWITCH CURRENT LIMIT (A)
2.0
SWITCH CURRENT LIMIT (A)
50
35
ISHDN (µA)
TA = 25°C unless otherwise noted.
–25
0
25
50
TEMPERATURE (°C)
1.0
0.8
0.6
0.4
75
100
0
0
20
60
40
DUTY CYCLE (%)
3493-3 G17
80
100
3493-3 G18
Operating Waveforms,
Discontinuous Mode
Operating Waveforms
VSW
5V/DIV
VSW
5V/DIV
IL
0.5A/DIV
0
VOUT
20mV/DIV
IL
0.5A/DIV
0
VOUT
20mV/DIV
1µs/DIV
1.2
0.2
3493-3 G14
VIN = 12V
VOUT = 3.3V
IOUT = 0.5A
L = 10µH
COUT = 10µF
1.4
3493-3 G19
VIN = 12V
VOUT = 3.3V
IOUT = 50mA
L = 10µH
COUT = 10µF
1µs/DIV
3493-3 G20
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FB (Pin 1): The LT3493-3 regulates its feedback pin to
780mV. Connect the feedback resistor divider tap to this
pin. Set the output voltage according to VOUT = 0.78V •
(1 + R1/R2). A good value for R2 is 10k.
GND (Pin 2): Tie the GND pin to a local ground plane below
the LT3493-3 and the circuit components. Return the
feedback divider to this pin.
BOOST (Pin 3): The BOOST pin is used to provide a drive
voltage, higher than the input voltage, to the internal
bipolar NPN power switch.
SW (Pin 4): The SW pin is the output of the internal power
switch. Connect this pin to the inductor, catch diode and
boost capacitor.
VIN (Pin 5): The VIN pin supplies current to the LT3493-3’s
internal regulator and to the internal power switch. This
pin must be locally bypassed.
SHDN (Pin 6): The SHDN pin is used to put the LT34933 in shutdown mode. Tie to ground to shut down the
LT3493-3. Tie to 2.3V or more for normal operation. If the
shutdown feature is not used, tie this pin to the VIN pin.
SHDN also provides a soft-start function; see the Applications Information section.
Exposed Pad (Pin 7): The Exposed Pad must be soldered
to the PCB and electrically connected to ground. Use a
large ground plane and thermal vias to optimize thermal
performance.
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BLOCK DIAGRA
5
VIN
VIN
C2
INT REG
AND
UVLO
ON OFF
SLOPE
COMP
R3
6
Σ
BOOST
R
Q
S
Q
D2
3
SHDN
C3
DRIVER
C4
Q1
SW
OSC
L1
VOUT
4
D1
FREQUENCY
FOLDBACK
VC
C1
gm
780mV
2
GND
1
R2
FB
R1
3493-3 BD
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LT3493-3
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OPERATIO (Refer to Block Diagram)
The LT3493-3 is a constant frequency, current mode stepdown regulator. A 750kHz oscillator enables an RS flipflop, turning on the internal 1.75A power switch Q1. An
amplifier and comparator monitor the current flowing
between the VIN and SW pins, turning the switch off when
this current reaches a level determined by the voltage at
VC. An error amplifier measures the output voltage through
an external resistor divider tied to the FB pin and servos the
VC node. If the error amplifier’s output increases, more
current is delivered to the output; if it decreases, less
current is delivered. An active clamp (not shown) on the VC
node provides current limit. The VC node is also clamped
to the voltage on the SHDN pin; soft-start is implemented
by generating a voltage ramp at the SHDN pin using an
external resistor and capacitor.
An internal regulator provides power to the control circuitry. This regulator includes an undervoltage lockout to
prevent switching when VIN is less than 6.8V. The SHDN
pin is used to place the LT3493-3 in shutdown, disconnecting the output and reducing the input current to less
than 2µA.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate the
internal bipolar NPN power switch for efficient operation.
The oscillator reduces the LT3493-3’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the output current during startup and overload.
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LT3493-3
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APPLICATIO S I FOR ATIO
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1%
resistors according to:
⎛ V
⎞
R1 = R2 ⎜ OUT – 1⎟
⎝ 0.78 V ⎠
R2 should be 20k or less to avoid bias current errors.
Reference designators refer to the Block Diagram.
An optional phase lead capacitor of 22pF between VOUT
and FB reduces light-load output ripple.
Input Voltage Range
The input voltage range for LT3493-3 applications depends on the output voltage and on the absolute maximum
ratings of the VIN and BOOST pins.
The minimum input voltage is determined by either the
LT3493-3’s minimum operating voltage of 6.8V, or by its
maximum duty cycle. The duty cycle is the fraction of time
that the internal switch is on and is determined by the input
and output voltages:
DC =
VOUT + VD
VIN – VSW + VD
where VD is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.4V at maximum load). This leads to a minimum input
voltage of:
V
+V
VIN(MIN) = OUT D – VD + VSW
DCMAX
with DCMAX = 0.91 (0.88 over temperature).
The maximum input voltage is determined by the absolute
maximum ratings of the VIN and BOOST pins. For constant-frequency operation, the maximum input voltage is
determined by the minimum duty cycle, DCMIN = 0.10. If
the duty cycle requirement is less than DCMIN, the part will
enter pulse-skipping mode. The onset of pulse-skipping
occurs at:
VIN(PS) =
VOUT + VD
– VD + VSW
DCMIN
In pulse-skipping mode, the part skips pulses to control
the inductor current and regulate the output voltage,
possibly producing a spectrum of frequencies below
750kHz.
Note that this is a restriction on the operating input voltage
to remain in constant-frequency operation; the circuit will
tolerate transient inputs up to the absolute maximum
ratings of the VIN and BOOST pins when the output is in
regulation. The input voltage should be limited to VIN(PS)
during overload conditions (short-circuit or start-up).
VSW
20V/DIV
IL
0.5A/DIV
VOUT
200mV/DIV
AC COUPLED
COUT = 10µF
VOUT = 3V
VIN = 30V
ILOAD = 0.75A
L = 10µH
2µs/DIV
3493-3 F01
Figure 1
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Minimum On Time
The part will still regulate the output at input voltages that
exceed VIN(PS) (up to 40V), however, the output voltage
ripple increases as the input voltage is increased. Figure 1
illustrates switching waveforms in continuous mode for a
3V output application near VIN(PS) = 33V.
As the input voltage is increased, the part is required to
switch for shorter periods of time. Delays associated with
turning off the power switch dictate the minimum on time
of the part. The minimum on time for the LT3493-3 is
130ns. Figure 2 illustrates the switching waveforms when
the input voltage is increased to VIN = 35V.
Now the required on time has decreased below the minimum on time of 130ns. Instead of the switch pulse width
becoming narrower to accommodate the lower duty cycle
requirement, the switch pulse width remains fixed at
130ns. In Figure 2 the inductor current ramps up to a value
exceeding the load current and the output ripple increases
to ~200mV. The part then remains off until the output
voltage dips below 100% of the programmed value before
it begins switching again.
Provided that the output remains in regulation and that the
inductor does not saturate, operation above VIN(PS) is safe
and will not damage the part. Figure 3 illustrates the
switching waveforms when the input voltage is increased
to its absolute maximum rating of 40V.
As the input voltage increases, the inductor current ramps
up quicker, the number of skipped pulses increases and
the output voltage ripple increases. For operation above
VIN(MAX) the only component requirement is that the
components be adequately rated for operation at the
intended voltage levels.
The part is robust enough to survive prolonged operation
under these conditions as long as the peak inductor
current does not exceed 2.2A. Inductor current saturation
may further limit performance in this operating regime.
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is:
L = 1.6 (VOUT + VD)
where VD is the voltage drop of the catch diode (~0.4V) and
L is in µH. With this value there will be no subharmonic
oscillation for applications with 50% or greater duty cycle.
The inductor’s RMS current rating must be greater than
your maximum load current and its saturation current
should be about 30% higher. For robust operation in fault
conditions, the saturation current should be above 2.2A.
To keep efficiency high, the series resistance (DCR) should
be less than 0.1Ω. Table 1 lists several vendors and types
that are suitable.
Of course, such a simple design guide will not always
result in the optimum inductor for your application. A
VSW
20V/DIV
VSW
20V/DIV
IL
0.5A/DIV
IL
0.5A/DIV
VOUT
200mV/DIV
AC COUPLED
VOUT
200mV/DIV
AC COUPLED
COUT = 10µF
VOUT = 3V
VIN = 35V
ILOAD = 0.75A
L = 10µH
2µs/DIV
Figure 2
3493-3 F02
COUT = 10µF
VOUT = 3V
VIN = 40V
ILOAD = 0.75A
L = 10µH
2µs/DIV
3493-3 F03
Figure 3
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LT3493-3
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APPLICATIO S I FOR ATIO
Table 1. Inductor Vendors
Vendor
URL
Part Series
Inductance Range (µH)
Size (mm)
Sumida
www.sumida.com
CDRH4D28
CDRH5D28
CDRH8D28
1.2 to 4.7
2.5 to 10
2.5 to 33
4.5 × 4.5
5.5 × 5.5
8.3 × 8.3
Toko
www.toko.com
A916CY
D585LC
2 to 12
1.1 to 39
6.3 × 6.2
8.1 × 8.0
Würth Elektronik
www.we-online.com
WE-TPC(M)
WE-PD2(M)
WE-PD(S)
1 to 10
2.2 to 22
1 to 27
4.8 × 4.8
5.2 × 5.8
7.3 × 7.3
larger value provides a higher maximum load current and
reduces output voltage ripple at the expense of slower
transient response. If your load is lower than 1.2A, then
you can decrease the value of the inductor and operate
with higher ripple current. This allows you to use a
physically smaller inductor, or one with a lower DCR
resulting in higher efficiency. There are several graphs in
the Typical Performance Characteristics section of this
data sheet that show the maximum load current as a
function of input voltage and inductor value for several
popular output voltages. Low inductance may result in
discontinuous mode operation, which is okay, but further
reduces maximum load current. For details of the maximum output current and discontinuous mode operation,
see Linear Technology Application Note 44.
Catch Diode
Depending on load current, a 1A to 2A Schottky diode is
recommended for the catch diode, D1. The diode must
have a reverse voltage rating equal to or greater than the
maximum input voltage. The ON Semiconductor MBRM140
is a good choice; it is rated for 1A continuous forward
current and a maximum reverse voltage of 40V.
Input Capacitor
Bypass the input of the LT3493-3 circuit with a 1µF or
higher value ceramic capacitor of X7R or X5R type. Y5V
types have poor performance over temperature and applied voltage and should not be used. A 1µF ceramic is
adequate to bypass the LT3493-3 and will easily handle
the ripple current. However, if the input power source has
high impedance, or there is significant inductance due to
long wires or cables, additional bulk capacitance may be
necessary. This can be provided with a low performance
electrolytic capacitor.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage ripple
at the LT3493-3 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 1µF capacitor is capable of this task, but only if it is
placed close to the LT3493-3 and the catch diode; see the
PCB Layout section. A second precaution regarding the
ceramic input capacitor concerns the maximum input
voltage rating of the LT3493-3. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (underdamped) tank circuit. If the LT3493-3 circuit
is plugged into a live supply, the input voltage can ring to
twice its nominal value, possibly exceeding the LT34933’s voltage rating. This situation is easily avoided; see the
Hot Plugging Safely section.
Output Capacitor
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by
the LT3493-3 to produce the DC output. In this role it
determines the output ripple so low impedance at the
switching frequency is important. The second function is
to store energy in order to satisfy transient loads and
stabilize the LT3493-3’s control loop.
Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A
good value is:
COUT = 65/VOUT
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Table 2. Capacitor Vendors
Vendor
Phone
URL
Part Series
Comments
Panasonic
(714) 373-7366
www.panasonic.com
Ceramic,
Polymer,
Tantalum
EEF Series
Kemet
(864) 963-6300
www.kemet.com
Ceramic,
Tantalum
Sanyo
(408) 749-9714
www.sanyovideo.com
Ceramic,
Polymer,
Tantalum
Murata
(404) 436-1300
AVX
Taiyo Yuden
(864) 963-6300
www.murata.com
Ceramic
www.avxcorp.com
Ceramic,
Tantalum
www.taiyo-yuden.com
Ceramic
where COUT is in µF. Use X5R or X7R types and keep in
mind that a ceramic capacitor biased with VOUT will have
less than its nominal capacitance. This choice will provide
low output ripple and good transient response. Transient
performance can be improved with a high value capacitor,
but a phase lead capacitor across the feedback resistor R1
may be required to get the full benefit (see the Compensation section).
For small size, the output capacitor can be chosen according to:
COUT = 25/VOUT
where COUT is in µF. However, using an output capacitor
this small results in an increased loop crossover frequency and increased sensitivity to noise. A 22pF capacitor connected between VOUT and the FB pin is required to
filter noise at the FB pin and ensure stability.
High performance electrolytic capacitors can be used for
the output capacitor. Low ESR is important, so choose
one that is intended for use in switching regulators. The
ESR should be specified by the supplier and should be
0.1Ω or less. Such a capacitor will be larger than a
ceramic capacitor and will have a larger capacitance, because the capacitor must be large to achieve low ESR.
Table 2 lists several capacitor vendors.
Figure 4 shows the transient response of the LT3493-3
with several output capacitor choices. The output is 3.3V.
The load current is stepped from 250mA to 1A and back to
250mA, and the oscilloscope traces show the output
T494, T495
POSCAP
TPS Series
voltage. The upper photo shows the recommended value.
The second photo shows the improved response (less
voltage drop) resulting from a larger output capacitor
and a phase lead capacitor. The last photo shows the
response to a high performance electrolytic capacitor.
Transient performance is improved due to the large output
capacitance.
BOOST Pin Considerations
Capacitor C3 and diode D2 are used to generate a boost
voltage that is higher than the input voltage. In most cases
a 0.1µF capacitor and fast switching diode (such as the
1N4148 or 1N914) will work well. Figure 5 shows two
ways to arrange the boost circuit. The BOOST pin must be
at least 2.3V above the SW pin for best efficiency. For
outputs of 3.3V and above, the standard circuit (Figure 5a)
is best. For outputs between 3V and 3.3V, use a 0.22µF
capacitor. For outputs between 2.5V and 3V, use a 0.47µF
capacitor and a small Schottky diode (such as the BAT-54).
For lower output voltages the boost diode can be tied to the
input (Figure 5b). The circuit in Figure 5a is more efficient
because the BOOST pin current comes from a lower
voltage source. You must also be sure that the maximum
voltage rating of the BOOST pin is not exceeded.
The minimum operating voltage of an LT3493-3 application is limited by the undervoltage lockout (6.8V) and by
the maximum duty cycle as outlined above. For proper
start-up, the minimum input voltage is also limited by the
boost circuit. If the input voltage is ramped slowly, or the
LT3493-3 is turned on with its SHDN pin when the output
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ILOAD
2A/DIV
VOUT
IL
0.5A/DIV
32.4k
FB
10µF
10k
VOUT
0.1V/DIV
AC COUPLED
40µs/DIV
3493-3 F04a
40µs/DIV
3493-3 F04b
40µs/DIV
3493-3 F04c
ILOAD
2A/DIV
VOUT
32.4k
IL
0.5A/DIV
3.3nF
10µF
×2
FB
10k
VOUT
0.1V/DIV
AC COUPLED
ILOAD
2A/DIV
VOUT
32.4k
IL
0.5A/DIV
+
FB
100µF
10k
SANYO
4TPB100M
VOUT
0.1V/DIV
AC COUPLED
Figure 4. Transient Load Response of the LT3493-3 with Different Output Capacitors
as the Load Current is Stepped from 250mA to 1A. VIN = 12V, VOUT = 3.3V, L = 10µH
D2
D2
C3
BOOST
LT3493-3
VIN
VIN
C3
BOOST
LT3493-3
SW
VOUT
VIN
VIN
SW
VOUT
GND
GND
3493-3 F05b
3493-3 F05a
VBOOST – VSW ≅ VOUT
MAX VBOOST ≅ VIN + VOUT
VBOOST – VSW ≅ VIN
MAX VBOOST ≅ 2VIN
(5b)
(5a)
Figure 5. Two Circuits for Generating the Boost Voltage
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is already in regulation, then the boost capacitor may not
be fully charged. Because the boost capacitor is charged
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on the
input and output voltages, and on the arrangement of the
boost circuit. The minimum load generally goes to zero
once the circuit has started. In many cases the discharged
output capacitor will present a load to the switcher which
will allow it to start. For 3.3V applications, the undervoltage lockout is high enough (6.8V) that the boost capacitor
always gets charged. For 5V applications with output
loads less than 20mA, the minimum input voltage required
to charge the boost capacitor is 6.8V. Note this higher
input voltage requirements is only in worst-case situation
where VIN is being ramped very slowly. For lower start-up
voltage, the boost diode can be tied to VIN; however this
restricts the input range to one-half of the absolute maximum rating of the BOOST pin.
At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
400mV above VOUT. At higher load currents, the inductor
current is continuous and the duty cycle is limited by the
maximum duty cycle of the LT3493-3, requiring a higher
input voltage to maintain regulation.
Soft-Start
The SHDN pin can be used to soft-start the LT3493-3,
reducing the maximum input current during start-up. The
SHDN pin is driven through an external RC filter to create
a voltage ramp at this pin. Figure 6 shows the start-up
waveforms with and without the soft-start circuit. By
VSW
10V/DIV
RUN
SHDN
GND
IL
0.5A/DIV
VOUT
2V/DIV
VIN = 12V
VOUT = 3.3V
L = 10µH
COUT = 10µF
20µs/DIV
3493-3 F07a
VIN = 12V
VOUT = 3.3V
L = 10µH
COUT = 10µF
20µs/DIV
3493-3 F07b
VSW
10V/DIV
RUN
15k
SHDN
0.1µF
GND
IL
0.5A/DIV
VOUT
2V/DIV
Figure 6. To Soft-Start the LT3493-3, Add a Resistor and Capacitor to the SHDN pin. VIN = 12V, VOUT = 3.3V, COUT = 10µF, RLOAD = 5Ω
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choosing a large RC time constant, the peak start up
current can be reduced to the current that is required to
regulate the output, with no overshoot. Choose the value
of the resistor so that it can supply 20µA when the SHDN
pin reaches 2.3V.
Shorted and Reversed Input Protection
If the inductor is chosen so that it won’t saturate excessively, an LT3493-3 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT3493-3 is absent. This may occur in battery charging
applications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT3493-3’s
output. If the VIN pin is allowed to float and the SHDN pin
is held high (either by a logic signal or because it is tied to
VIN), then the LT3493-3’s internal circuitry will pull its
quiescent current through its SW pin. This is fine if your
system can tolerate a few mA in this state. If you ground
the SHDN pin, the SW pin current will drop to essentially
zero. However, if the VIN pin is grounded while the output
is held high, then parasitic diodes inside the LT3493-3 can
pull large currents from the output through the SW pin and
the VIN pin. Figure 7 shows a circuit that will run only when
the input voltage is present and that protects against a
shorted or reversed input.
D4
VIN
VIN
BOOST
LT3493-3
SHDN
GND
VOUT
SW
FB
BACKUP
3493-3 F08
Figure 7. Diode D4 Prevents a Shorted Input from Discharging
a Backup Battery Tied to the Output; It Also Protects the Circuit
from a Reversed Input. The LT3493-3 Runs Only When the Input
is Present
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT3493-3 circuits. However, these
capacitors can cause problems if the LT3493-3 is plugged
into a live supply (see Linear Technology Application Note
88 for a complete discussion). The low loss ceramic
capacitor combined with stray inductance in series with
the power source forms an underdamped tank circuit, and
the voltage at the VIN pin of the LT3493-3 can ring to twice
the nominal input voltage, possibly exceeding the LT34933’s rating and damaging the part. If the input supply is
poorly controlled or the user will be plugging the LT34933 into an energized supply, the input network should be
designed to prevent this overshoot.
Figure 8 shows the waveforms that result when an LT34933 circuit is connected to a 24V supply through six feet of
24-gauge twisted pair. The first plot is the response with
a 2.2µF ceramic capacitor at the input. The input voltage
rings as high as 35V and the input current peaks at 20A.
One method of damping the tank circuit is to add another
capacitor with a series resistor to the circuit. In Figure 8b
an aluminum electrolytic capacitor has been added. This
capacitor’s high equivalent series resistance damps the
circuit and eliminates the voltage overshoot. The extra
capacitor improves low frequency ripple filtering and can
slightly improve the efficiency of the circuit, though it is
likely to be the largest component in the circuit. An
alternative solution is shown in Figure 8c. A 1Ω resistor is
added in series with the input to eliminate the voltage
overshoot (it also reduces the peak input current). A 0.1µF
capacitor improves high frequency filtering. This solution
is smaller and less expensive than the electrolytic capacitor. For high input voltages its impact on efficiency is
minor, reducing efficiency less than one half percent for a
5V output at full load operating from 24V.
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CLOSING SWITCH
SIMULATES HOT PLUG
IIN
VIN
DANGER!
LT3493-3
+
VIN
20V/DIV
RINGING VIN MAY EXCEED
ABSOLUTE MAXIMUM
RATING OF THE LT3493-3
2.2µF
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
IIN
5A/DIV
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
20µs/DIV
(8a)
LT3493-3
+
10µF
35V
AI.EI.
+
VIN
20V/DIV
2.2µF
IIN
5A/DIV
(8b)
20µs/DIV
1Ω
LT3493-3
+
0.1µF
VIN
20V/DIV
2.2µF
IIN
5A/DIV
(8c)
3493-3 F09
20µs/DIV
Figure 8. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation When the LT3493-3 is Connected to a Live Supply
Figure 9 shows an equivalent circuit for the LT3493-3
control loop. The error amp is a transconductance amplifier with finite output impedance. The power section,
consisting of the modulator, power switch and inductor, is
modeled as a transconductance amplifier generating an
output current proportional to the voltage at the VC node.
Note that the output capacitor integrates this current, and
that the capacitor on the VC node (CC) integrates the error
amplifier output current, resulting in two poles in the loop.
RC provides a zero. With the recommended output capacitor, the loop crossover occurs above the RCCC zero. This
simple model works well as long as the value of the
CURRENT MODE
POWER STAGE
SW
gm =
+1.6A/V
LT3493-3
–
0.7V
OUT
R1
RC
60k
CC
100pF
GND
ERROR
AMPLIFIER
CPL
FB
gm =
300µA/V
VC
+
The LT3493-3 uses current mode control to regulate the
output. This simplifies loop compensation. In particular,
the LT3493-3 does not require the ESR of the output
capacitor for stability allowing the use of ceramic capacitors to achieve low output ripple and small circuit size.
inductor is not too high and the loop crossover frequency
is much lower than the switching frequency. With a larger
ceramic capacitor (very low ESR), crossover may be lower
and a phase lead capacitor (CPL) across the feedback
divider may improve the phase margin and transient
–
Frequency Compensation
ESR
780mV
+
C1
C1
1M
R2
3493-3 F10
Figure 9. Model for Loop Response
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response. Large electrolytic capacitors may have an ESR
large enough to create an additional zero, and the phase
lead may not be necessary.
If the output capacitor is different than the recommended
capacitor, stability should be checked across all operating
conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how
to test the stability using a transient load.
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 11 shows
the recommended component placement with trace,
ground plane and via locations. Note that large, switched
currents flow in the LT3493-3’s VIN and SW pins, the catch
diode (D1) and the input capacitor (C2). The loop formed
by these components should be as small as possible and
tied to system ground in only one place. These components, along with the inductor and output capacitor,
should be placed on the same side of the circuit board, and
their connections should be made on that layer. Place a
C2
local, unbroken ground plane below these components,
and tie this ground plane to system ground at one location,
ideally at the ground terminal of the output capacitor C1.
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB node small so that the ground pin and
ground traces will shield it from the SW and BOOST nodes.
Include vias near the exposed GND pad of the LT3493-3 to
help remove heat from the LT3493-3 to the ground plane.
High Temperature Considerations
The die temperature of the LT3493-3 must be lower than
the maximum rating of 125°C. This is generally not a
concern unless the ambient temperature is above 85°C.
For higher temperatures, care should be taken in the
layout of the circuit to ensure good heat sinking of the
LT3493-3. The maximum load current should be derated
as the ambient temperature approaches 125°C. The die
temperature is calculated by multiplying the LT3493-3
power dissipation by the thermal resistance from junction
to ambient. Power dissipation within the LT3493-3 can be
estimated by calculating the total power loss from an
efficiency measurement and subtracting the catch diode
D1
SYSTEM
GROUND
VOUT
C1
VIN
SHUTDOWN
3493-3 F11
: VIAS TO LOCAL GROUND PLANE
: OUTLINE OF LOCAL GROUND PLANE
Figure 10. A Good PCB Layout Ensures Proper, Low EMI Operation
3493-3f
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loss. The resulting temperature rise at full load is nearly
independent of input voltage. Thermal resistance depends on the layout of the circuit board, but 64°C/W is
typical for the (2mm × 3mm) DFN (DCB) package.
limited by the maximum rating of the BOOST pin. The 12V
circuit shows how to overcome this limitation using an
additional zener diode.
Outputs Greater Than 6V
Application notes AN19, AN35 and AN44 contain more
detailed descriptions and design information for Buck
regulators and other switching regulators. The LT1376
data sheet has a more extensive discussion of output
ripple, loop compensation and stability testing. Design
Note DN100 shows how to generate a bipolar output
supply using a Buck regulator.
For outputs greater than 6V, add a resistor of 1k to 2.5k
across the inductor to damp the discontinuous ringing of
the SW node, preventing unintended SW current. The 12V
Step-Down Converter circuit in the Typical Applications
section shows the location of this resistor. Also note that
for outputs above 6V, the input voltage range will be
Other Linear Technology Publications
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TYPICAL APPLICATIO S
0.78V Step-Down Converter
1N4148
VIN
6.8V TO 25V
VIN
BOOST
0.1µF 3.3µH
LT3493-3
ON OFF
VOUT
0.78V
1.2A
SW
SHDN
MBRM140
GND
FB
47µF
2.2µF
3493-3 TA02
1.8V Step-Down Converter
1N4148
VIN
6.8V TO 25V
VIN
BOOST
0.1µF
LT3493-3
ON OFF
5µH
SW
SHDN
MBRM140
GND
2.2µF
26.1k
FB
VOUT
1.8V
1.2A
22µF
20k
3493-3 TA03
3493-3f
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LT3493-3
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2.5V Step-Down Converter
BAT54
VIN
6.8V TO 28V
VIN
0.47µF 6.8µH
LT3493-3
ON OFF
VOUT
2.5V
1A, VIN > 7V
1.2A, VIN > 10V
BOOST
SW
SHDN
MBRM140
GND
22.1k
FB
22µF
10k
1µF
3493-3 TA04
3.3V Step-Down Converter
VIN
6.8V TO 36V
1N4148
VIN
0.1µF 8.2µH
LT3493-3
ON OFF
VOUT
3.3V
1A, VIN > 7V
1.2A, VIN > 12V
BOOST
SW
SHDN
MBRM140
GND
32.4k
FB
10µF
10k
1µF
3493-3 TA05
5V Step-Down Converter
VIN
6.8V TO 36V
1N4148
VIN
0.1µF 10µH
LT3493-3
ON OFF
SW
SHDN
MBRM140
GND
1µF
VOUT
5V
0.9A, VIN > 7V
1.1A, VIN > 14V
BOOST
59k
FB
10µF
11k
3493-3 TA06
3493-3f
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PACKAGE DESCRIPTION
DCB Package
6-Lead Plastic DFN (2mm × 3mm)
(Reference LTC DWG # 05-08-1715)
0.70 ±0.05
3.55 ±0.05
1.65 ±0.05
(2 SIDES)
2.15 ±0.05
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
1.35 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
R = 0.115
TYP
R = 0.05
TYP
2.00 ±0.10
(2 SIDES)
3.00 ±0.10
(2 SIDES)
0.40 ± 0.10
4
6
1.65 ± 0.10
(2 SIDES)
PIN 1 NOTCH
R0.20 OR 0.25
× 45° CHAMFER
PIN 1 BAR
TOP MARK
(SEE NOTE 6)
3
0.200 REF
0.75 ±0.05
1
(DCB6) DFN 0405
0.25 ± 0.05
0.50 BSC
1.35 ±0.10
(2 SIDES)
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (TBD)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
3493-3f
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TYPICAL APPLICATIO
12V Step-Down Converter
D1
6V
1N4148
1k*
0.25W
VIN
14.5V TO 36V
VIN
BOOST
0.1µF
LT3493-3
ON OFF
22µH
SW
SHDN
MBRM140
GND
71.5k
FB
VOUT
12V
1A
4.7µF
4.99k
1µF
*FOR CONTINUOUS OPERATION ABOVE 30V
USE TWO 2k, 0.25W RESISTORS IN PARALLEL.
D1: CMDZ5235B
3493-3 TA07
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1766
60V, 1.2A IOUT, 200kHz, High Efficiency Step-Down
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VIN: 5.5V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5mA, ISD = 25µA,
TSSOP16 and TSSOP16E Packages
LT1767
25V, 1.2A IOUT, 1.2MHz, High Efficiency Step-Down
DC/DC Converter
VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 1mA, ISD = 6µA,
MS8E Package
LT1933
500mA IOUT, 500kHz Step-Down Switching Regulator
in SOT-23
VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.6mA, ISD = 1µA,
ThinSOT Package
LT1936
36V, 1.4A IOUT, 500kHz, High Efficiency Step-Down
DC/DC Converter
VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.9mA, ISD = 1µA,
MS8E Package
LT1940
Dual 25V, 1.4A IOUT, 1.1MHz, High Efficiency Step-Down
DC/DC Converter
VIN: 3.6V to 25V, VOUT(MIN) = 1.2V, IQ = 3.5mA, ISD =
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