LT3508
Dual Monolithic 1.4A
Step-Down Switching
Regulator
DESCRIPTION
FEATURES
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Wide Input Voltage Range: 3.7V to 36V
Two 1.4A Output Switching Regulators with Internal
Power Switches
Adjustable 250kHz to 2.5MHz Switching Frequency
Synchronizable over the Full Frequency Range
Anti-Phase Switching Reduces Ripple
Uses Small Inductors and Ceramic Capacitors
Accurate Programmable Undervoltage Lockout
Independent Tracking, Soft-Start and Power Good
Circuits Ease Supply Sequencing
Output Adjustable Down to 800mV
Small 4mm × 4mm 24-Pin QFN or 16-Pin Thermally
Enhanced TSSOP Surface Mount Packages
APPLICATIONS
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The LT®3508 is a dual current mode PWM step-down
DC/DC converter with internal power switches capable of
generating two 1.4A outputs. The wide input voltage range
of 3.7V to 36V makes the LT3508 suitable for regulating
power from a wide variety of sources, including automotive batteries, 24V industrial supplies and unregulated wall
adapters. Both converters are synchronized to a single oscillator programmable up to 2.5MHz and run with opposite
phases, reducing input ripple current. Its high operating
frequency allows the use of small, low cost inductors and
ceramic capacitors, resulting in low, predictable output
ripple. Each regulator has independent tracking and softstart circuits and generates a power good signal when its
output is in regulation, easing power supply sequencing
and interfacing with microcontrollers and DSPs.
Cycle-by-cycle current limit, frequency foldback and thermal shutdown provide protection against shorted outputs,
and soft-start eliminates input current surge during startup. The low current ( VIN + 2.5V. See Minimum Operating Voltage in the
Applications Information section for details.
Note 5: Current limit is guaranteed by design and/or correlation to static
test. Slope compensation reduces current limit at higher duty cycles.
Note 6: Current flows into pin.
Note 7: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed the maximum operating junction temperature
range when overtemperature protection is active. Continuous operation
above the specified maximum operating junction temperature may impair
device reliability.
3508fd
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LT3508
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency, VOUT = 3.3V
Efficiency, VOUT = 5V
TA = 25°C
f = 700kHz
TA = 25°C
f = 700kHz
85
VIN = 24V
EFFICIENCY (%)
EFFICIENCY (%)
85
VIN = 32V
80
75
70
85
VIN = 12V
VIN = 24V
80
75
VIN = 32V
70
65
60
0
1.5
0.5
1
LOAD CURRENT (A)
TA = 25°C
f = 1MHz
80
VIN = 3.3V
75
VIN = 5V
70
VIN = 12V
65
60
0
0.5
1
LOAD CURRENT (A)
3508 G01
55
1.5
0
1.5
0.5
1
LOAD CURRENT (A)
3508 G02
3508 G03
Switch Current Limit
vs Temperature
Feedback Voltage
0.810
Switch Current Limit
vs Duty Cycle
3.0
3.0
2.5
2.5
TA = 25°C
TYPICAL
0.800
CURRENT LIMIT (A)
0.805
CURRENT LIMIT (A)
FEEDBACK VOLTAGE (V)
EFFICIENCY (%)
VIN = 12V
90
65
Efficiency, VOUT = 1.8V
90
95
2.0
1.5
1.0
2.0
MINIMUM
1.5
1.0
0.795
0.5
0.790
–50 –25
0
0.5
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
0
100
75
50
25
0
0
0.5
1.0
1.5
FREQUENCY (MHz)
2.0
2.5
3.0
RT = 33.2k
1.0
0.8
0.6
0.4
0.2
0
–50 –25
0
100
80
Switching Frequency Foldback
SWITCHING FREQUENCY (MHz)
SWITCHING FREQUENCY (MHz)
RT (kΩ)
1.2
TA = 25°C
125
40
60
DUTY CYCLE (%)
3508 G06
Switching Frequency
vs Temperature
Switching Frequency vs RT
150
20
3508 G05
3508 G04
175
0
25 50 75 100 125 150
TEMPERATURE (°C)
25 50 75 100 125 150
TEMPERATURE (°C)
TA = 25°C
2.5
RT = 7.50k
2.0
1.5
1.0
RT = 33.2k
0.5
RT = 169k
0
0
100 200 300 400 500 600 700 800
FEEDBACK VOLTAGE (mV)
3508 G07
3508 G08
3508 G09
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LT3508
TYPICAL PERFORMANCE CHARACTERISTICS
Quiescent Current
5.0
VC Voltages
TA = 25°C
4.5
VIN1
30
2.0
CLAMP VOLTAGE
OUTPUT CURRENT (μA)
4.0
VC VOLTAGE (V)
3.5
3.0
2.5
2.0
1.5
1.0
TO SWITCH
1.5
1.0
0.5
0
15 20 25 30
INPUT VOLTAGE (V)
35
0
–50 –25
40
0
3508 G10
15
SOURCING
10
0
25
50
75
100 125 150
TEMPERATURE (°C)
3508 G12
3508 G11
Switch Voltage Drop
350
20
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
Boost Pin Current
35
TA = 25°C
TA = 25°C
30
300
BOOST PIN CURRENT (mA)
10
250
200
150
100
25
20
15
10
5
50
0
0
1
0.5
SWITCH CURRENT (A)
0
1
0.5
SWITCH CURRENT (A)
0
1.5
1.5
3508 G14
3508 G13
SHDN Pin Current
Undervoltage Lockout
4.0
120
TA = –45°C
100
VIN1
3.5
3.0
TA = 125°C
INPUT VOLTAGE (V)
5
SWITCH VOLTAGE (mV)
0
SINKING
25
5
VIN2
0.5
SHDN PIN CURRENT (μA)
INPUT CURRENT (mA)
Error Amp Output Current
35
2.5
80
TA = 25°C
60
40
VIN2
2.5
2.0
1.5
1.0
20
0.5
0
0
5
10 15 20 25 30
SHDN PIN VOLTAGE (V)
35
40
3508 G15
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3508 G16
3508fd
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LT3508
PIN FUNCTIONS
BOOST1, BOOST2: The BOOST pins are used to provide
drive voltages, higher than the input voltage, to the internal
NPN power switches. Tie through a diode to a 2.8V or
higher supply, such as VOUT or VIN.
Exposed Pad: The exposed pad metal of the package provides both electrical contact to ground and good thermal
contact to the printed circuit board. The exposed pad must
be soldered to the circuit board for proper operation.
FB1, FB2: The LT3508 regulates each feedback pin to
0.800V. Connect the feedback resistor divider taps to
these pins.
GND: Tie the GND pins directly to the Exposed Pad and
ground plane.
PG1, PG2: The power good pins are the open-collector
outputs of an internal comparator. PG remains low until
the FB pin is within 10% of the final regulation voltage.
As well as indicating output regulation, the PG pins can
be used to sequence the two switching regulators. These
pins can be left unconnected. The PG outputs are valid
when VIN1 is greater than 3.7V and SHDN is high. The PG
comparators are disabled in shutdown.
RT/SYNC: The RT/SYNC pin is used to set the internal
oscillator frequency. Tie a 33.2k resistor from RT/SYNC
to GND for a 1MHz switching frequency. To synchronize
the part to an external frequency, drive the RT/SYNC pin
with a logic-level signal with positive and negative pulse
widths of at least 80ns.
SHDN: The shutdown pin is used to put the LT3508 in
shutdown mode. Pull the pin below 0.3V to shut down the
LT3508. The 2.63V threshold can function as an accurate
undervoltage lockout (UVLO), preventing the regulator
from operating until the input voltage has reached the
programmed level. Do not drive SHDN more than 6V
above VIN1.
SW1, SW2: The SW pins are the outputs of the internal
power switches. Connect these pins to the inductors, catch
diodes and boost capacitors.
TRACK/SS1, TRACK/SS2: The TRACK/SS pins are used
to soft-start the two channels, to allow one channel to
track the other output, or to allow both channels to track
another output. For tracking, tie a resistor divider to this
pin from the tracked output. For soft-start, tie a capacitor
to this pin. An internal 1.2μA soft-start current charges
the capacitor to create a voltage ramp at the pin. In the
TSSOP package, if these pins are unused, tie a 47pF or
greater capacitor from each pin to GND.
VC1, VC2: The VC pins are the outputs of the internal error
amps. The voltages on these pins control the peak switch
currents. These pins are normally used to compensate the
control loops, but can also be used to override the loops.
Pull these pins to ground with an open drain to shut down
each switching regulator separately.
VIN1: The VIN1 pin supplies current to the LT3508 internal
circuitry and to the internal power switch connected to
SW1 and must be locally bypassed. VIN1 must be greater
than 3.7V for channel 1 or channel 2 to operate. If VIN1
is greater than 3.7V, channel 2 can operate with VIN2 as
low as 3V.
VIN2: The VIN2 pin supplies current to the internal power
switch connected to SW2 and must be locally bypassed.
Connect this pin directly to VIN1 unless power for channel 2 is coming from a different source. VIN2 must be
greater than 3V and VIN1 must be greater than 3.7V for
channel 2 to operate.
3508fd
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LT3508
BLOCK DIAGRAM
SHDN
VIN1
INT REG
AND REF
RT/SYNC
CLK1
MASTER
OSC
CLK2
1.2μA
VIN
TRACK/SS
VIN
CIN
0.75V
+
Σ
SLOPE
R
S
C1
BOOST
D2
Q
SLAVE
OSC
CLK
C3
SW
L1
OUT
+
ERROR
AMP
0.625V
VC
TRACK/SS
C1
R1
R2
0.80V
CC
+
+
ILIMIT
CLAMP
75mV
PG
GND
+
–
CF
–
RC
FB
+
+
–
–
D1
3508 F01
Figure 1. Block Diagram of the LT3508 with Associated External Components (One of Two Switching Regulators Shown)
3508fd
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LT3508
OPERATION
The LT3508 is a dual constant frequency, current mode
regulator with internal power switches. Operation can be
best understood by referring to the Block Diagram. If the
SHDN pin is tied to ground, the LT3508 is shut down and
draws minimal current from the input source tied to the
VIN pins. If the SHDN pin exceeds 1V, the internal bias
circuits turn on, including the internal regulator, reference
and oscillator. The switching regulators will only begin to
operate when the SHDN pin exceeds 2.63V.
The switcher is a current mode regulator. Instead of directly
modulating the duty cycle of the power switch, the feedback
loop controls the peak current in the switch during each
cycle. Compared to voltage mode control, current mode
control improves loop dynamics and provides cycle-bycycle current limit. A pulse from the oscillator sets the
RS flip-flop and turns on the internal NPN power switch.
Current in the switch and the external inductor begins to
increase. When this current exceeds a level determined
by the voltage at VC, current comparator C1 resets the
flip-flop, turning off the switch. The current in the inductor
flows through the external Schottky diode and begins to
decrease. The cycle begins again at the next pulse from the
oscillator. In this way, the voltage on the VC pin controls
the current through the inductor to the output. The internal
error amplifier regulates the output current by continually
adjusting the VC pin voltage. The threshold for switching
on the VC pin is 0.8V, and an active clamp of 1.75V limits
the output current.
The switching frequency is set either by the resistance to
GND at the RT/SYNC pin or the frequency of the logic-level
signal driving the RT/SYNC pin. A detection circuit monitors
for the presence of a SYNC signal on the pin and switches
between the two modes. Unique circuitry generates the
appropriate slope compensation ramps and generates the
180° out-of-phase clocks for the two channels.
The switching regulator performs frequency foldback
during overload conditions. An amplifier senses when
VFB is less than 0.625V and begins decreasing the oscillator frequency down from full frequency to 12% of the
nominal frequency when VFB = 0V. The FB pin is less than
0.8V during start-up, short-circuit and overload conditions.
Frequency foldback helps limit switch current under these
conditions.
The switch driver operates either from VIN or from the
BOOST pin. An external capacitor and Schottky diode are
used to generate a voltage at the BOOST pin that is higher
than the input supply. This allows the driver to saturate the
internal bipolar NPN power switch for efficient operation.
The TRACK/SS pin serves as an alternative input to the
error amplifier. The amplifier will use the lowest voltage of
either the reference of 0.8V or the voltage on the TRACK/
SS pin as the positive input of error amplifier. Since the
TRACK/SS pin is driven by a constant current source, a
single capacitor on the pin will generate a linear ramp on
the output voltage. Tying the TRACK/SS pin to a resistor
divider from the output of one of the switching regulators
allows one output to track another.
The PG output is an open-collector transistor that is off
when the output is in regulation, allowing an external resistor to pull the PG pin high. Power good is valid when
the LT3508 is enabled (SHDN is high) and VIN1 is greater
than 3.7V.
3508fd
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LT3508
APPLICATIONS INFORMATION
Setting the Output Voltage
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
⎛V
⎞
R1= R2 ⎜ OUT – 1⎟
⎝ 0.8V ⎠
where VF is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.4V at maximum load).
Example: ISW = 1.5A and IBOOST = 50mA, VOUT = 3.3V,
βSW = 1.5A/50mA = 30, DCMAX = 1/(1+1/30) = 96%:
VIN(MIN) =
3.3V + 0.4V
– 0.4V + 0.4V = 3.8V
96%
R2 should be 20k or less to avoid bias current errors.
Reference designators refer to the Block Diagram.
Maximum Operating Voltage
Minimum Operating Voltage
The maximum operating voltage is determined by the
Absolute Maximum Ratings of the VIN and BOOST pins,
and by the minimum duty cycle:
The minimum operating voltage is determined either by
the LT3508’s undervoltage lockout or by its maximum duty
cycle. If VIN1 and VIN2 are tied together, the undervoltage
lockout is at 3.7V or below. If the two inputs are used
separately, then VIN1 has an undervoltage lockout of 3.7V
or below and VIN2 has an undervoltage lockout of 3V or
below. Because the internal supply runs off VIN1, channel 2 will not operate unless VIN1 > 3.7V. The duty cycle
is the fraction of time that the internal switch is on and is
determined by the input and output voltages:
VOUT + VF
DC =
VIN – VSW + VF
Unlike many fixed frequency regulators, the LT3508 can
extend its duty cycle by turning on for multiple cycles.
The LT3508 will not switch off at the end of each clock
cycle if there is sufficient voltage across the boost capacitor (C3 in Figure 1). Eventually, the voltage on the boost
capacitor falls and requires refreshing. Circuitry detects
this condition and forces the switch to turn off, allowing
the inductor current to charge up the boost capacitor. This
places a limitation on the maximum duty cycle as follows:
1
DCMAX =
1+
1
β SW
where βSW is equal to the SW pin current divided by the
BOOST pin current as shown in the Typical Performance
Characteristics section. This leads to a minimum input
voltage of:
VIN(MIN) =
VOUT + VF
– VF + VSW
DCMAX
DCMIN = tON(MIN) • f
where tON(MIN) is equal to 130ns (for TJ > 125°C tON(MIN)
is equal to 150ns) and f is the switching frequency.
Running at a lower switching frequency allows a lower
minimum duty cycle. The maximum input voltage before
pulse skipping occurs depends on the output voltage and
the minimum duty cycle:
VIN(PS) =
VOUT + VF
– VF + VSW
DCMIN
Example: f = 790kHz, VOUT = 3.3V, DCMIN = 130ns • 790kHz
= 0.103:
VIN(PS) =
3.3V + 0.4V
– 0.4V + 0.4V = 36V
0.103
The LT3508 will regulate the output current at input voltages
greater than VIN(PS). For example, an application with an
output voltage of 1.8V and switching frequency of 1.5MHz
has a VIN(PS) of 11.3V, as shown in Figure 2. Figure 3 shows
operation at 18V. Output ripple and peak inductor current
have significantly increased. Exceeding VIN(PS) is safe if
the output is in regulation, if the external components have
adequate ratings to handle the peak conditions and if the
peak inductor current does not exceed 3.2A. A saturating
inductor may further reduce performance. Do not exceed
VIN(PS) during start-up or overload conditions (for outputs
greater than 5V, use VOUT = 5V to calculate VIN(PS)). For
operation above 20V in pulse skipping mode, program
the switching frequency to 1.1MHz or less.
3508fd
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LT3508
APPLICATIONS INFORMATION
Table 1. Programming the Switching Frequency
VOUT
100mV/DIV
(AC)
SWITCHING FREQUENCY (MHz)
RT (kΩ)
2.5
7.50
2.2
9.76
2
11.5
IL
500mA/DIV
2μs/DIV
3508 F02
Figure 2. Operation Below VIN(PS). VIN = 10V, VOUT = 1.8V and
fSW = 1.5MHz
VOUT
100mV/DIV
(AC)
IL
500mA/DIV
2μs/DIV
3508 F03
Figure 3. Operation Above VIN(PS). VIN = 18V, VOUT = 1.8V
and fSW = 1.5MHz. Output Ripple and Peak Inductor Current
Increase
1.8
14
1.6
16.9
1.4
20.5
1.2
26.1
1
33.2
0.9
38.3
0.8
44.2
0.7
52.3
0.6
61.9
0.5
76.8
0.45
88.7
0.4
100
0.35
115
0.3
140
0.25
169
Inductor Selection and Maximum Output Current
Setting the Switching Frequency
A good first choice for the inductor value is:
The switching frequency is programmed either by driving
the RT/SYNC pin with a logic level SYNC signal or by tying
a resistor from the RT/SYNC pin to ground. A graph for
selecting the value of RT for a given operating frequency
is shown in the Typical Application section. Suggested
programming resistors for various switching frequencies
are shown in Table 1.
1.2µH
f
where VF is the voltage drop of the catch diode (~0.4V)
and f is in MHz. The inductor’s RMS current rating must be
greater than the maximum load current and its saturation
current should be at least 30% higher. For highest efficiency,
the series resistance (DCR) should be less than 0.1Ω.
Table 2 lists several vendors and types that are suitable.
Choosing a high switching frequency will allow the smallest
overall solution size. However, at high input voltages the
efficiency can drop significantly with increasing switching
frequency. The choice of switching frequency will also
impact the input voltage range, inductor and capacitor
selection, and compensation. See the related sections
for details.
L = (VOUT + VF ) •
Table 2. Inductor Vendors
VENDOR
URL
PART SERIES
TYPE
Coilcraft
www.coilcraft
MSS7341
Shielded
Murata
www.murata.com
LQH55D
Open
TDK
www.component.tdk.com
SLF7045
SLF10145
Shielded
Shielded
Toko
www.toko.com
DC62CB
D63CB
D75C
D75F
Shielded
Shielded
Shielded
Open
Sumida
www.sumida.com
CR54
CDRH74
CDRH6D38
CR75
Open
Shielded
Shielded
Open
3508fd
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LT3508
APPLICATIONS INFORMATION
The optimum inductor for a given application may differ
from the one indicated by this simple design guide. A larger
value inductor provides a higher maximum load current,
and reduces the output voltage ripple. If your load is lower
than the maximum load current, then you can relax the
value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor,
or one with a lower DCR resulting in higher efficiency.
Be aware that if the inductance differs from the simple
rule above, then the maximum load current will depend
on input voltage. In addition, low inductance may result
in discontinuous mode operation, which further reduces
maximum load current. For details of discontinuous mode
operation, see Application Note 44. Finally, for duty cycles
greater than 50% (VOUT/VIN > 0.5), a minimum inductance
is required to avoid sub-harmonic oscillations:
LMIN = (VOUT + VF ) •
0.8µH
f
where f is in MHz. The current in the inductor is a triangle
wave with an average value equal to the load current. The
peak switch current is equal to the output current plus
half the peak-to-peak inductor ripple current. The LT3508
limits its switch current in order to protect itself and the
system from overload faults. Therefore, the maximum
output current that the LT3508 will deliver depends on
the switch current limit, the inductor value, and the input
and output voltages.
When the switch is off, the potential across the inductor
is the output voltage plus the catch diode drop. This gives
the peak-to-peak ripple current in the inductor:
ΔIL =
(1– DC)(VOUT + VF )
L•f
where f is the switching frequency of the LT3508 and L
is the value of the inductor. The peak inductor and switch
current is:
ISW(PK) = IL(PK) = IOUT +
ΔIL
2
To maintain output regulation, this peak current must be
less than the LT3508’s switch current limit ILIM. ILIM is
at least 2A for at low duty cycles and decreases linearly
to 1.55A at DC = 90%. The maximum output current is a
function of the chosen inductor value:
IOUT(MAX) = ILIM –
ΔIL
ΔI
= 2A • (1– 0.25 • DC) – L
2
2
Choosing an inductor value so that the ripple current is
small will allow a maximum output current near the switch
current limit.
One approach to choosing the inductor is to start with the
simple rule given above, look at the available inductors, and
choose one to meet cost or space goals. Then use these
equations to check that the LT3508 will be able to deliver
the required output current. Note again that these equations
assume that the inductor current is continuous. Discontinuous operation occurs when IOUT is less than ∆IL/2.
Input Capacitor Selection
Bypass the VIN pins of the LT3508 circuit with a ceramic
capacitor of X7R or X5R type. For switching frequencies above 500kHz, use a 4.7μF capacitor or greater. For
switching frequencies below 500kHz, use a 10μF or higher
capacitor. If the VIN pins are tied together only a single
capacitor is necessary. If the VIN pins are separated, each
pin will need its own bypass. The following paragraphs
describe the input capacitor considerations in more detail.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at
the LT3508 input and to force this switching current into a
tight local loop, minimizing EMI. The input capacitor must
have low impedance at the switching frequency to do this
effectively, and it must have an adequate ripple current
rating. With two switchers operating at the same frequency
but with different phases and duty cycles, calculating the
input capacitor RMS current is not simple. However, a
conservative value is the RMS input current for the channel
that is delivering most power (VOUT times IOUT):
ICIN(RMS) = IOUT •
VOUT (VIN – VOUT )
VIN
<
IOUT
2
and is largest when VIN = 2VOUT (50% duty cycle). As
the second, lower power channel draws input current,
3508fd
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LT3508
APPLICATIONS INFORMATION
the input capacitor’s RMS current actually decreases as
the out-of-phase current cancels the current drawn by
the higher power channel. Considering that the maximum
load current from a single channel is ~1.4A, RMS ripple
current will always be less than 0.7A.
The high frequency of the LT3508 reduces the energy storage requirements of the input capacitor. The combination
of small size and low impedance (low equivalent series
resistance or ESR) of ceramic capacitors makes them the
preferred choice. The low ESR results in very low voltage
ripple. Ceramic capacitors can handle larger magnitudes
of ripple current than other capacitor types of the same
value. Use X5R and X7R types.
An alternative to a high value ceramic capacitor is a lower
value ceramic along with a larger electrolytic capacitor. The
electrolytic capacitor likely needs to be greater than 10μF
in order to meet the ESR and ripple current requirements.
The input capacitor is likely to see high surge currents
when the input source is applied. Tantalum capacitors
can fail due to an oversurge of current. Only use tantalum
capacitors with the appropriate surge current rating. The
manufacturer may also recommend operation below the
rated voltage of the capacitor.
A final caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plugging
the circuit into a live power source), this tank can ring,
doubling the input voltage and damaging the LT3508. The
solution is to either clamp the input voltage or dampen
the tank circuit by adding a lossy capacitor in parallel with
the ceramic capacitor. For details see Application Note 88.
LT3508’s control loop. Ceramic capacitors have very low
equivalent series resistance (ESR) and provide the best
ripple performance. A good value is:
COUT =
where COUT is in μF. Use X5R or X7R types. This choice
will provide low output ripple and good transient response.
Transient performance can be improved with a high value
capacitor if the compensation network is also adjusted to
maintain the loop bandwidth. A lower value of output capacitor can be used, but transient performance will suffer. With
an external compensation network, the loop gain can be
lowered to compensate for the lower capacitor value. Look
carefully at the capacitor’s data sheet to find out what the
actual capacitance is under operating conditions (applied
voltage and temperature). A physically larger capacitor, or
one with a higher voltage rating, may be required. High
performance electrolytic capacitors can be used for the
output capacitor. Low ESR is important, so choose one
that is intended for use in switching regulators. The ESR
should be specified by the supplier, and should be 0.05Ω
or less. Such a capacitor will be larger than a ceramic
capacitor and will have a larger capacitance, because the
capacitor must be large to achieve low ESR. Table 3 lists
several capacitor vendors.
Table 3. Capacitor Vendors
VENDOR
PART SERIES
COMMENTS
Panasonic
Ceramic
Polymer
Tantalum
EEF Series
Kemet
Ceramic
Tantalum
Sanyo
Ceramic
Polymer
Tantalum
Output Capacitor Selection
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by the
LT3508 to produce the DC output. In this role it determines
the output ripple, and low impedance at the switching
frequency is important. The second function is to store
energy in order to satisfy transient loads and stabilize the
50V 1MHz
•
VOUT
f
Murata
Ceramic
AVX
Ceramic
Tantalum
Taiyo Yuden
Ceramic
TDK
Ceramic
T494, T495
POSCAP
TPS Series
3508fd
12
LT3508
APPLICATIONS INFORMATION
Diode Selection
The catch diode (D1 from Figure 1) conducts current only
during switch off time. Average forward current in normal
operation can be calculated from:
ID(AVG) =
IOUT (VIN – VOUT )
VIN
The only reason to consider a diode with a larger current
rating than necessary for nominal operation is for the
worst-case condition of shorted output. The diode current will then increase to the typical peak switch current.
Peak reverse voltage is equal to the regulator input voltage.
Use a diode with a reverse voltage rating greater than the
input voltage. Table 4 lists several Schottky diodes and their
manufacturers. If operating at high ambient temperatures,
consider using a Schottky with low reverse leakage.
D2
VR
(V)
IAVE
(A)
MBR0520L
20
0.5
MBR0540
40
0.5
VF at 1A
(mV)
C3
BOOST
LT3508
VIN
Table 4. Schottky Diodes
PART NUMBER
and higher, the standard circuit (Figure 4a) is best. For
outputs between 2.8V and 3.3V, use a small Schottky
diode (such as the BAT-54). For lower output voltages,
the boost diode can be tied to the input (Figure 4b). The
circuit in Figure 4a is more efficient because the boost pin
current comes from a lower voltage source. Finally, the
anode of the boost diode can be tied to another source
(VAUX) that is at least 3V (Figure 4c). For example, if you
are generating a 3.3V output, and the 3.3V output is on
whenever the particular channel is on, the anode of the
BOOST diode can be connected to the 3.3V output. In any
case, be sure that the maximum voltage at the BOOST pin
is both less than 60V and the voltage difference between
the BOOST and SW pins is less than 30V.
VIN
VF at 2A
(mV)
VOUT
SW
GND
On Semiconductor
VBOOST – VSW ≅ VOUT
MAX VBOOST ≅ VIN + VOUT
(4a)
620
MBRM120E
20
1
530
MBRM140
40
1
550
D2
C3
BOOST
Diodes Inc.
LT3508
VIN
B0530W
30
0.5
B120
20
1
500
B130
30
1
500
B140HB
40
1
DFLS140
40
1.1
B240
40
2
VIN
VOUT
SW
GND
VBOOST – VSW ≅ VIN
MAX VBOOST ≅ 2VIN
510
(4b)
500
BOOST Pin Considerations
The capacitor and diode tied to the BOOST pin generate
a voltage that is higher than the input voltage. In most
cases, a 0.22μF capacitor and fast switching diode (such
as the CMDSH-3 or MMSD914LT1) will work well. For applications 1MHz or faster, a 0.1μF capacitor is sufficient.
Use a 0.47μF capacitor or greater for applications running
below 500kHz. Figure 4 shows three ways to arrange the
boost circuit. The BOOST pin must be more than 2.5V
above the SW pin for full efficiency. For outputs of 3.3V
D2
VAUX > 3V
BOOST
C3
LT3508
VIN
VIN
VOUT
SW
GND
VBOOST – VSW ≅ VAUX
MAX VBOOST ≅ VAUX + VIN
MINIMUM VALUE FOR VAUX = 3V
3508 F04
(4c)
Figure 4. Generating the Boost Voltage
3508fd
13
LT3508
APPLICATIONS INFORMATION
The minimum operating voltage of an LT3508 application is limited by the undervoltage lockout (≈3.7V) and
by the maximum duty cycle. The boost circuit also limits
the minimum input voltage for proper start-up. If the
input voltage ramps slowly, or the LT3508 turns on when
the output is already in regulation, the boost capacitor
may not be fully charged. Because the boost capacitor
charges with the energy stored in the inductor, the circuit
will rely on some minimum load current to get the boost
circuit running properly. This minimum load will depend
on input and output voltages, and on the arrangement of
the boost circuit. The minimum load current generally
goes to zero once the circuit has started. Figure 5 shows
a plot of minimum load to start and to run as a function
of input voltage. Even without an output load current, in
Minimum Input Voltage, VOUT = 3.3V
6.5
TA = 25°C
VOUT = 3.3V
INPUT VOLTAGE (V)
6.0
5.5
5.0
TO START
4.5
4.0
TO RUN
3.5
3.0
1
10
100
1000
LOAD CURRENT (mA)
10000
3508 F05a
Minimum Input Voltage, VOUT = 5V
9
TA = 25°C
VOUT = 5V
INPUT VOLTAGE (V)
8
7
TO START
6
TO RUN
5
4
1
100
10
1000
LOAD CURRENT (mA)
10000
3508 G05b
Figure 5. The Minimum Input Voltage Depends on Output
Voltage, Load Current and Boost Circuit
many cases the discharged output capacitor will present
a load to the switcher that will allow it to start. The plots
show the worst case, where VIN is ramping very slowly.
Frequency Compensation
The LT3508 uses current mode control to regulate the
output. This simplifies loop compensation. In particular, the
LT3508 does not require the ESR of the output capacitor
for stability, so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size.
Frequency compensation is provided by the components
tied to the VC pin, as shown in Figure 1. Generally a capacitor (CC) and a resistor (RC) in series to ground are used. In
addition, there may be a lower value capacitor in parallel.
This capacitor (CF) is not part of the loop compensation
but is used to filter noise at the switching frequency, and
is required only if a phase-lead capacitor is used or if the
output capacitor has high ESR.
Loop compensation determines the stability and transient
performance. Designing the compensation network is a bit
complicated and the best values depend on the application
and in particular the type of output capacitor. A practical
approach is to start with one of the circuits in this data
sheet that is similar to your application and tune the compensation network to optimize the performance. Stability
should then be checked across all operating conditions,
including load current, input voltage and temperature. The
LT1375 data sheet contains a more thorough discussion of
loop compensation and describes how to test the stability
using a transient load.
Figure 6 shows an equivalent circuit for the LT3508 control
loop. The error amplifier is a transconductance amplifier
with finite output impedance. The power section, consisting
of the modulator, power switch and inductor, is modeled
as a transconductance amplifier generating an output
current proportional to the voltage at the VC pin. Note that
the output capacitor integrates this current, and that the
capacitor on the VC pin (CC) integrates the error amplifier
output current, resulting in two poles in the loop. In most
cases a zero is required and comes from either the output
capacitor ESR or from a resistor RC in series with CC.
This simple model works well as long as the value of the
inductor is not too high and the loop crossover frequency
3508fd
14
LT3508
APPLICATIONS INFORMATION
is much lower than the switching frequency. A phase-lead
capacitor (CPL) across the feedback divider may improve
the transient response.
LT3508
CURRENT MODE
POWER STAGE
gm = 2.5S
VSW
ERROR
AMPLIFIER
OUTPUT
R1
CPL
ESR
–
+
0.8V
C1
+
VC
GND
R2
CF
POLYMER
OR
TANTALUM
R4 =
2.63V
VL – 2.63V
– 8µA
R3
CERAMIC
Example: switching should not start until the input is above
4.75V and is to stop if the input falls below 4V.
CC
VH = 4.75V, VL = 4.0V
3508 F06
Figure 6. Model for Loop Response
Shutdown and Undervoltage Lockout
Figure 7 shows how to add undervoltage lockout (UVLO)
to the LT3508. Typically, UVLO is used in situations where
the input supply is current limited, or has a relatively high
source resistance. A switching regulator draws constant
power from the source, so source current increases as
source voltage drops. This looks like a negative resistance
load to the source and can cause the source to current
limit or latch low under low source voltage conditions.
UVLO prevents the regulator from operating at source
voltages where the problems might occur.
An internal comparator will force the part into shutdown
below the minimum VIN1 of 3.7V. This feature can be used to
prevent excessive discharge of battery-operated systems.
LT3508
2.6V
–
VC
R3
SHDN
C1
VH – VL
7.5µA
C1
RC
VIN
R3 =
FB
gm =
300μS
2M
If an adjustable UVLO threshold is required, the SHDN
pin can be used. The threshold voltage of the SHDN pin
comparator is 2.63V. Current hysteresis is added above the
SHDN threshold. This can be used to set voltage hysteresis
of the UVLO using the following:
+
R4
TRACK/SS
8μA
7.5μA
R3 =
4.75V – 4V
= 100k
7.5µA
R4 =
2.63V
= 461k
4V – 2.63V
– 8µA
100k
Keep the connection from the resistor to the SHDN pin
short and make sure the interplane or surface capacitance
to switching nodes is minimized. If high resistor values are
used, the SHDN pin should be bypassed with a 1nF capacitor to prevent coupling problems from the switch node.
Soft-Start
The output of the LT3508 regulates to the lowest voltage
present at either the TRACK/SS pin or an internal 0.8V
reference. A capacitor from the TRACK/SS pin to ground
is charged by an internal 1.2μA current source resulting
in a linear output ramp from 0V to the regulated output
whose duration is given by:
tRAMP =
CSS • 0.8V
1.2µA
At power up, internal open-collector outputs discharge
both TRACK/SS pins. The pins clamp at 1.3V.
3508 F07
Figure 7. Undervoltage Lockout
3508fd
15
LT3508
APPLICATIONS INFORMATION
Output Tracking and Sequencing
Complex output tracking and sequencing between channels
can be implemented using the LT3508’s TRACK/SS and
PG pins. Figure 8 shows several configurations for output
tracking and sequencing of 5V and 3.3V applications.
Independent Start-Up
Independent soft-start for each channel is shown in Figure 8a. The output ramp time for each channel is set by the
soft-start capacitor as described in the soft-start section.
Ratiometric Start-Up
Coincident Start-Up
VOUT1
VOUT1
VOUT2
VOUT1
VOUT2
1V/DIV
VOUT2
1V/DIV
1V/DIV
20ms/DIV
20ms/DIV
TRACK/SS1 VOUT1
5V
20ms/DIV
5V
TRACK/SS1 VOUT1
0.1μF
LT3508
TRACK/SS2 VOUT2
LT3508
3.3V
5V
TRACK/SS1 VOUT1
0.1μF
0.22μF
LT3508
3.3V
TRACK/SS2 VOUT2
3.3V
TRACK/SS2 VOUT2
0.047μF
R1
28.7k
R2
10.0k
(8a)
(8b)
(8c)
Output Sequencing
Controlled Power Up and Down
VOUT1
VOUT1
VOUT2
VOUT2
1V/DIV
1V/DIV
EXTERNAL SOURCE
20ms/DIV
20ms/DIV
TRACK/SS1 VOUT1
0.1μF
5V
TRACK/SS1 VOUT1
EXTERNAL
SOURCE
LT3508
PG1
TRACK/SS2 VOUT2
3.3V
+
–
5V
LT3508
TRACK/SS2 VOUT2
3.3V
0.047μF
R1
28.7k
R2
10.0k
(8d)
(8e)
Figure 8
3508fd
16
LT3508
APPLICATIONS INFORMATION
If precise output ramp up and down is required, drive the
TRACK/SS pins as shown in Figure 8e.
Ratiometric tracking is achieved in Figure 8b by connecting
both the TRACK/SS pins together. In this configuration the
TRACK/SS pin source current is doubled (2.4μA) which
must be taken into account when calculating the output
rise time.
Multiple Inputs
For applications requiring large inductors due to high VIN
to VOUT ratios, a 2-stage step down approach may reduce
inductor size by allowing an increase in frequency. A dual
step-down application (Figure 9) steps down the input
voltage (VIN1) to the highest output voltage, then uses that
voltage to power the second output (VIN2). VOUT1 must
be able to provide enough current for its output plus the
input current at VIN2 when VOUT2 is at its maximum load.
Do not tie TRACK/SS1 and TRACK/SS2 together if using
multiple inputs. If VIN2 is below 3V, TRACK/SS2 pulls low
and would hold TRACK/SS1 low as well if the two pins
are tied together, which would prevent channel 1 from
operating.
By connecting a feedback network from VOUT1 to the
TRACK/SS2 pin with the same ratio that set the VOUT2
voltage, absolute tracking shown in Figure 8c is implemented. A small VOUT2 voltage offset will be present due
to the TRACK/SS2 1.2μA source current. This offset can
be corrected for by slightly reducing the value of R2. Use
a resistor divider such that when VOUT1 is in regulation,
TRACK/SS2 is pulled up to 1V or greater. If TRACK/SS is
below 1V, the output may regulate FB to a voltage lower
than the 800mV reference voltage.
For applications with multiple input voltages, the LT3508
can accommodate input voltages as low as 3V on VIN2.
This can be useful in applications regulating outputs from
a PCI Express bus, where the 12V input is power limited
and the 3.3V input has power available to drive other
outputs. In this case, tie the 12V input to VIN1 and the
3.3V input to VIN2. See the Typical Application section for
an example circuit.
Shorted and Reverse Input Protection
Figure 8d illustrates output sequencing. When VOUT1 is
within 10% of its regulated voltage, PG1 releases the
TRACK/SS2 soft-start pin allowing VOUT2 to soft-start.
In this case PG1 will be pulled up to 1.3V by the TRACK/
SS pin.
VIN
5.7V TO 36V
C1
4.7μF
ON OFF
D1
If the inductor is chosen so that it won’t saturate excessively, an LT3508 step-down regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
OUT1
VIN2
VIN1
BOOST1
OUT1
5V
0.9A
BOOST2
C2
0.1μF
L1 6.8μH
C3
0.1μF
SW1
D3
R1
56.2k
C4
10μF
R5
39k
C6
100pF
C8
1nF
L2
3.3μH
OUT2
1.8V
1A
SW2
D4
LT3508
FB1
R3
10.7k
D2
SHDN
R2
18.7k
FB2
VC1
VC2
TRACK/SS1
TRACK/SS2
PG1
PG2
GND
R6
47k
R4
15.0k
R7
100k
RT/SYNC
R8
33.2k
C9
3.3nF
C5
47μF
C7
330pF
fSW = 1MHz
3508 F09
POWER
GOOD
Figure 9. 1MHz, Wide Input Range 5V and 1.8V Outputs
3508fd
17
LT3508
APPLICATIONS INFORMATION
LT3508 is absent. This may occur in battery charging
applications or in battery back-up systems where a battery
or some other supply is diode OR-ed with the LT3508’s
output. If the VIN pin is allowed to float and the SHDN pin
is held high (either by a logic signal or because it is tied
to VIN), then the LT3508’s internal circuitry will pull its
quiescent current through its SW pin. This is fine if your
system can tolerate a few mA in this state. If you ground
the SHDN pin, the SW pin current will drop to essentially
zero. However, if the VIN pin is grounded while the output
is held high, then parasitic diodes inside the LT3508 can
pull large currents from the output through the SW pin
and the VIN pin. Figure 10 shows a circuit that will run
only when the input voltage is present and that protects
against a shorted or reversed input.
PARASITIC DIODE
D4
VIN
VIN
SW
VOUT
LT3508
PCB Layout
For proper operation and minimum EMI, care must be taken
during printed circuit board layout. Figure 11 shows the
recommended PCB layout with trace and via locations. Note
that large, switched currents flow in the LT3508’s VIN and
SW pins, the catch diode (D1) and the input capacitor (CIN).
The loop formed by these components should be as small
as possible. These components, along with the inductor
and output capacitor, should be placed on the same side
of the circuit board, and their connections should be made
on that layer. Place a local, unbroken ground plane below
these components. The SW and BOOST nodes should be
as small as possible. Finally, keep the FB and VC nodes
small so that the ground traces will shield them from the
SW and BOOST nodes. The exposed pad on the bottom of
the package must be soldered to ground so that the pad
acts as a heat sink. To keep thermal resistance low, extend
the ground plane as much as possible, and add thermal
vias under and near the LT3508 to additional ground planes
within the circuit board and on the bottom side.
3508 F10
Figure 10. Diode D4 Prevents a Shorted Input from Discharging
a Backup Battery Tied to the Output
(11a) Example Layout for FE16 Package
(11b) Example Layout for QFN Package
Figure 11. A Good PCB Layout Ensures Proper Low EMI Operation
3508fd
18
LT3508
APPLICATIONS INFORMATION
High Temperature Considerations
The die temperature of the LT3508 must be lower than the
maximum rating of 125°C (150°C for the H-grade). This is
generally not a concern unless the ambient temperature is
above 85°C. For higher temperatures, care should be taken
in the layout of the circuit to ensure good heat sinking of
the LT3508. The maximum load current should be derated
as the ambient temperature approaches 125°C (150°C
for the H-grade). The die temperature is calculated by
multiplying the LT3508 power dissipation by the thermal
resistance from junction to ambient. Power dissipation
within the LT3508 can be estimated by calculating the total
power loss from an efficiency measurement and subtracting the catch diode loss. Thermal resistance depends on
the layout of the circuit board, but values from 30°C/W to
60°C/W are typical. Die temperature rise was measured
on a 4-layer, 6.5cm × 7.5cm circuit board in still air at a
load current of 1.4A (fSW = 700kHz). For a 12V input to
3.3V output the die temperature elevation above ambient
was 13°C; for 24VIN to 3.3VOUT the rise was 18°C; for
12VIN to 5VOUT the rise was 14°C and for 24VIN to 5VOUT
the rise was 19°C.
Outputs Greater Than 6V
For outputs greater than 6V, add a resistor of 1k to 2.5k
across the inductor to damp the discontinuous ringing of
the SW node, preventing unintended SW current. The 12V
output circuit in the Typical Applications section shows
the location of this resistor.
Other Linear Technology Publications
Application Notes 19, 35 and 44 contain more detailed
descriptions and design information for step-down regulators and other switching regulators. The LT1376 data
sheet has a more extensive discussion of output ripple,
loop compensation and stability testing. Design Note 318
shows how to generate a dual polarity output supply using
a step-down regulator.
3508fd
19
LT3508
TYPICAL APPLICATIONS
1MHz, 3.3V and 1.8V Outputs with Sequencing
VIN
3.9V TO 16V
C1
4.7μF
D1
OUT2
ON OFF
VIN1 VIN2 SHDN
BOOST1
C2
0.1μF
L1 3.3μH
OUT1
1.8V
1.4A
C3
0.1μF
SW1
D4
LT3508
FB1
R5
47k
R3
15.0k
C4
47μF
C6
330pF
L2 4.7μH
R2
35.7k
FB2
VC1
VC2
TRACK/SS1
TRACK/SS2
PG1
PG2
GND
R6
39k
R4
11.5k
R8
33.2k
C5
10μF
R7
100k
RT/SYNC
C8
1nF
C7
150pF
fSW = 1MHz
C1 TO C5: X5R OR X7R
D1, D2: MMSD4148
D3: DIODES INC. B140
D4: DIODES INC. B240A
OUT2
3.3V
1.4A
SW2
D3
R1
18.7k
D2
BOOST2
3508 TA02
POWER
GOOD
3.3V and 5V Dual Output Step-Down Converter with Output Sequencing
VIN
5.7V TO 36V
C1
4.7μF
D1
ON OFF
VIN1 VIN2 SHDN
BOOST1
OUT1
3.3V
1.4A
C2
0.22μF
L1 6.8μH
C3
0.22μF
SW1
D3
R1
35.7k
C4
22μF
R5
51k
C6
150pF
C8
1nF
C1 TO C5: X5R OR X7R
D1, D2: MMSD4148
D3: DIODES INC. B140
D4: DIODES INC. B240A
OUT2
5V
1.4A
L2 10μH
SW2
D4
LT3508
FB1
R3
11.5k
D2
BOOST2
FB2
VC1
VC2
TRACK/SS1
TRACK/SS2
PG1
PG2
GND
R2
56.2k
R6
43k
fSW = 700kHz
C5
10μF
R7
100k
RT/SYNC
R8
52.3k
R4
10.7k
C7
100pF
3508 TA03
POWER
GOOD
3508fd
20
LT3508
TYPICAL APPLICATIONS
1MHz, Wide Input Range 5V and 1.8V Outputs
VIN
5.7V TO 36V
C1
4.7μF
ON OFF
D1
OUT1
VIN2
VIN1
BOOST1
OUT1
5V
0.9A
BOOST2
C2
0.1μF
L1 6.8μH
C3
0.1μF
SW1
D3
R1
56.2k
C4
10μF
R5
39k
C6
100pF
C8
1nF
C1 TO C5: X5R OR X7R
D1, D2: MMSD4148
D3: DIODES INC. B240A
D4: DIODES INC. B120
L2
3.3μH
OUT2
1.8V
1A
SW2
D4
LT3508
FB1
R3
10.7k
D2
SHDN
R2
18.7k
FB2
VC1
VC2
TRACK/SS1
TRACK/SS2
PG1
PG2
GND
R6
47k
R8
33.2k
fSW = 1MHz
C5
47μF
R7
100k
RT/SYNC
C9
3.3nF
R4
15.0k
C7
330pF
3508 TA04
POWER
GOOD
3508fd
21
LT3508
TYPICAL APPLICATIONS
1MHz, 5V and 12V Outputs
VIN
14V TO 36V
C1
4.7μF
D1
OUT2
OUT1
12V
1.4A*
ON OFF
VIN1 VIN2 SHDN
BOOST1
D2
C3
0.1μF
C2
0.1μF
L1 15μH
SW1
R2 1k
D3
D4
LT3508
FB1
C4
4.7μF
R5
43k
C6
100pF
C8
1nF
L2 6.8μH
SW2
R1 154k
R4
11.0k
OUT2
5V
1.4A*
BOOST2
FB2
VC1
VC2
TRACK/SS1
TRACK/SS2
PG1
PG2
GND
R3
56.2k
R6
39k
R8
100k
RT/SYNC
R9
33.2k
C5
10μF
R7
10.7k
C7
100pF
fSW = 1MHz
C1 TO C5: X5R OR X7R
D1, D2: MMSD4148
D3: DIODES INC. B240A
D4: DIODES INC. B140
R2: USE 0.25W RESISTOR. FOR CONTINUOUS OPERATION
ABOVE 30V, USE TWO 2k, 0.25W RESISTORS IN PARALLEL
3508 TA06
POWER
GOOD
*DERATE OUTPUT CURRENT AT HIGHER AMBIENT TEMPERATURES
AND INPUT VOLTAGES TO MAINTAIN JUNCTION TEMPERATURE
BELOW THE ABSOLUTE MAXIMUM
3508fd
22
LT3508
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
FE Package
16-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663 Rev I)
Exposed Pad Variation BA
4.90 – 5.10*
(.193 – .201)
2.74
(.108)
2.74
(.108)
16 1514 13 12 1110
6.60 t0.10
9
2.74
(.108)
4.50 t0.10
2.74 6.40
(.108) (.252)
BSC
SEE NOTE 4
0.45 t0.05
1.05 t0.10
0.65 BSC
1 2 3 4 5 6 7 8
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.09 – 0.20
(.0035 – .0079)
0.50 – 0.75
(.020 – .030)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
3. DRAWING NOT TO SCALE
0.25
REF
1.10
(.0433)
MAX
0s – 8s
0.65
(.0256)
BSC
0.195 – 0.30
(.0077 – .0118)
TYP
0.05 – 0.15
(.002 – .006)
FE16 (BA) TSSOP REV I 0211
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
3508fd
23
LT3508
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
UF Package
24-Lead Plastic QFN (4mm × 4mm)
(Reference LTC DWG # 05-08-1697)
0.70 ±0.05
4.50 ± 0.05
2.45 ± 0.05
3.10 ± 0.05 (4 SIDES)
PACKAGE OUTLINE
0.25 ±0.05
0.50 BSC
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
4.00 ± 0.10
(4 SIDES)
BOTTOM VIEW—EXPOSED PAD
0.75 ± 0.05
R = 0.115
TYP
PIN 1 NOTCH
R = 0.20 TYP OR
0.35 × 45° CHAMFER
23 24
0.40 ± 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
2.45 ± 0.10
(4-SIDES)
(UF24) QFN 0105
0.200 REF
0.00 – 0.05
0.25 ± 0.05
0.50 BSC
NOTE:
1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGGD-X)—TO BE APPROVED
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE, IF PRESENT
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3508fd
24
LT3508
REVISION HISTORY
(Revision history begins at Rev D)
REV
DATE
DESCRIPTION
PAGE NUMBER
D
8/11
Revised Typical Application drawing TA01a
Changed RT/SYNC pin name to RT/SYNC
1
1, 2, 6, 7, 8,
10, 17, 20, 21,
22, 26
Updated Pin Configuration drawings
2
Revised Exposed Pad, VC1, VC2 and VIN1 pin descriptions in Pin Functions section
6
Made minor text edits in Operation and Applications Information sections
Updated equations in Input Capacitor and Shutdown and Undervoltage Lockout sections and revised Figure 4 of
Applications Information
Moved last paragraph of Multiple Inputs section to the Soft-Start section in Applications Information
8, 10
11, 13, 15
17
3508fd
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
25
LT3508
TYPICAL APPLICATION
5V, 1.8V Output from PCI Express
VIN
12V
C1
4.7μF
R9
40.2k
BOOST1
BOOST2
C3
0.1μF
L1 6.8μH
OUT1
5V
0.9A
C4
0.1μF
SW1
D3
R1
52.3k
C6
10μF
R5
43k
D4
LT3508
C8
0.047μF
C9
100pF
OUT2
1.8V
1.4A
R2
18.7k
FB2
VC1
VC2
TRACK/SS1
TRACK/SS2
PG1
PG2
GND
R6
47k
R8
33.2k
fSW = 1MHz
R4
15.0k
C5
47μF
R7
100k
RT/SYNC
C10
0.047μF
C1 TO C6: X5R OR X7R
D1, D2: MMSD4148
D3: DIODES INC. B140
D4: DIODES INC. B120
L2
3.3μH
SW2
FB1
R3
10k
D2
SHDN
D1
R10
14.7k
VIN2
VIN1
VIN2
3.3V
C2
4.7μF
C7
330pF
3508 TA05
POWER
GOOD
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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25V, 2.75A (IOUT), 1.25MHz, High Efficiency Step-Down
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LT1766
60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down
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VIN: 5.5V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5mA, TSSOP16/TSSOP16E
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VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 1mA, MS8, MS8E Packages
LT1940/LT1940L
Dual Monolithic 1.4A, 1.1MHz Step-Down Switching
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Dual 600mA, 1.5MHz, Synchronous Step-Down
Regulator
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LT3493
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2mm × 3mm DFN
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LT3501/LT3510
Dual 3A/2A, 1.5MHz High Efficiency Step-Down
Switching Regulators
VIN: 3.6V to 25V, VOUT(MIN) = 0.8V, IQ = 3.7mA, ISD < 10μA,
TSSOP20E Package
LT3506/LT3506A
Dual Monolithic 1.6A, 1.1MHz Step-Down Switching
Regulators
VIN: 3.6V to 25V, VOUT(MIN) = 0.8V, IQ = 3.8mA, 16-Lead DFN and 16-Lead
TSSOPE Packages
LTC3701
Two Phase, Dual, 500kHz, Constant Frequency, Current
Mode, High Efficiency Step-Down DC/DC Controller
VIN: 2.5V to 10V, VOUT(MIN) = 0.8V, IQ = 460μA, SSOP-16 Package
LTC3736
Dual Two Phase, No RSENSE™, Synchronous Controller
with Output Tracking
VIN: 2.75V to 9.8V, VOUT(MIN) = 0.6V, IQ = 300μA, 4mm × 4mm QFN or
SSOP-24 Packages
LTC3737
Dual Two Phase, No RSENSE DC/DC Controller with
Output Tracking
VIN: 2.75V to 9.8V, VOUT(MIN) = 0.6V, IQ = 220μA, 4mm × 4mm QFN or
SSOP-24 Packages
3508fd
26 Linear Technology Corporation
LT 0811 REV D • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
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