LT3509
Dual 36V, 700mA
Step-Down Regulator
Features
Description
Two 700mA Switching Regulators with Internal
Power Switches
n Wide 3.6V to 36V Operating Range
n Overvoltage Lockout Protects Circuit Through 60V
Supply Transients
n Short-Circuit Robust
n Low Dropout Voltage: 95% Maximum Duty Cycle
n Adjustable 300kHz to 2.2MHz Switching Frequency
Synchronizable Over the Full Range
n Uses Small Inductors and Ceramic Capacitors
n Integrated Boost Diodes
n Internal Compensation
n Thermally Enhanced 14-Lead (4mm × 3mm)
DFN and 16 Lead MSOP Packages
The LT®3509 is a dual, current mode, step-down switching
regulator, with internal power switches each capable of
providing 700mA output current. This regulator provides a
compact and robust solution for multi-rail systems in harsh
environments. It incorporates several protection features
including overvoltage lockout and cycle-by-cycle current
limit. Thermal shutdown provides additional protection.
The loop compensation components and the boost diodes
are integrated on-chip. Switching frequency is set by a
single external resistor. External synchronization is also
possible. The high maximum switching frequency allows
the use of small inductors and ceramic capacitors for low
ripple. Constant frequency operation above the AM band
avoids interference with radio reception, making the LT3509
well suited for automotive applications. Each regulator
has an independent shutdown and soft-start control pin.
When both converters are powered down, the common
circuitry enters a low current shutdown state.
n
Applications
n
n
n
n
n
Automotive Electronics
Industrial Controls
Wall Transformer Regulation
Networking Devices
CPU, DSP, or FPGA Power
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners.
Typical Application
3.3V and 5V Dual Output Step-Down Converter
2.2µF
VIN
BD
BOOST1
BOOST2
80
0.1µF 10µH
SW1
5V
700mA
SW2
LT3509
31.6k
53.6k
MBRM140
DA1
DA2
FB1
FB2
MBRM140
22µF
10.2k
1nF
SYNC
1nF
RT
GND
VOUT = 3.3V
75
70
65
60
RUN/SS1 RUN/SS2
10µF
VOUT = 5V
85
0.1µF
6.8µH
3.3V
700mA
Efficiency
90
EFFICIENCY (%)
6.5V TO 36V
(TRANSIENT TO 60V)
60.4k
fSW = 700kHz
55
10.2k
3509 TA01a
50
0.0
VIN = 12V
fSW = 700kHz
0.1
0.2
0.3 0.4 0.5
LOAD CURRENT (A)
0.6
0.7
3509 TA01b
3509fd
For more information www.linear.com/LT3509
1
LT3509
Absolute Maximum Ratings
(Note 1)
VIN Pin (Note 2).........................................................60V
BD Pin........................................................................20V
BOOST Pins...............................................................60V
BOOST Pins above SW..............................................30V
RUN/SS, FB, RT, SYNC pins.........................................6V
Operating Junction Temperature Range (Notes 3, 6)
LT3509E............................................. –40°C to 125°C
LT3509I.............................................. –40°C to 125°C
LT3509H............................................. –40°C to 150°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec.)
MSOP Package.................................................. 300°C
pin configuration
TOP VIEW
TOP VIEW
DA1
1
14 FB1
BOOST1
2
13 RUN/SS1
SW1
3
VIN
4
SW2
5
10 RT
BOOST2
6
9 RUN/SS2
DA2
7
8 FB2
DA1
BOOST1
SW1
VIN
VIN
SW2
BOOST2
DA2
12 BD
15
11 SYNC
1
2
3
4
5
6
7
8
17
16
15
14
13
12
11
10
9
FB1
RUN/SS1
AGND
BD
SYNC
RT
RUN/SS2
FB2
MSE PACKAGE
16-LEAD PLASTIC MSOP
DE14 PACKAGE
14-LEAD (4mm × 3mm) PLASTIC DFN
θJA = 43°C/W, θJC = 4.3°C/W
EXPOSED PAD (PIN 15) IS GND, MUST BE SOLDERED TO PCB
θJA = 43°C/W, θJC = 4.3°C/W
EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
order information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3509EDE#PBF
LT3509EDE#TRPBF
3509
14-Lead (4mm × 3mm) Plastic DFN
–40°C to 125°C
LT3509IDE#PBF
LT3509IDE#TRPBF
3509
14-Lead (4mm × 3mm) Plastic DFN
–40°C to 125°C
LT3509EMSE#PBF
LT3509EMSE#TRPBF
3509
16-Lead Plastic MSOP with Exposed Pad
–40°C to 125°C
LT3509IMSE#PBF
LT3509IMSE#TRPBF
3509
16-Lead Plastic MSOP with Exposed Pad
–40°C to 125°C
LT3509HMSE#PBF
LT3509HMSE#TRPBF
3509
16-Lead Plastic MSOP with Exposed Pad
–40°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
2
3509fd
For more information www.linear.com/LT3509
LT3509
Electrical
Characteristics
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C, VIN = 12V. (Note 3)
PARAMETER
CONDITIONS
MIN
TYP
MAX
3.3
3.6
V
37
38.5
40
V
1.9
2.2
mA
9
15
µA
0.8
0.816
V
VIN Undervoltage Lockout
VIN Overvoltage Lockout
Input Quiescent Current
Not Switching VFB > 0.8V
Input Shutdown Current
V(RUN/SS[1,2]) < 0.3V
Feedback Pin Voltage
Reference Voltage Line Regulation
l
0.784
3.6V < VIN < 36V
0.01
RUN/SS Shutdown Threshold
0.4
0.6
RUN/SS Voltage for Full IOUT
RUN/SS Pin Pull-up Current
Feedback Pin Bias Current (Note 4)
0.7
VFB = 0.8V
l
DA Comparator Current Threshold
Boost Pin Current
%/V
0.8
V
2
V
1
1.3
µA
90
500
nA
1.05
1.4
1.9
A
0.7
0.95
1.2
A
22
36
mA
0.01
1.0
µA
l
Switch Current Limit
UNITS
ISW = 0.9A
Switch Leakage Current
Switch Saturation Voltage
ISW = 0.9A (Note 5)
0.32
Minumum Boost Voltage above Switch
ISW = 0.9A
1.5
2.2
Boost Diode Forward Voltage
IBD= 20mA
0.7
0.9
V
Boost Diode Leakage
VR = 30V
0.1
5
µA
Switching Frequency
RT = 40.2kΩ
RT = 180kΩ
RT = 14.1kΩ
1.0
260
2.15
1.08
290
2.5
l
Sync Pin Input Threshold
0.92
237
2.0
V
1.0
Switch Minimum Off-Time
l
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device reliability
and lifetime.
Note 2. Absolute Maximum Voltage at the VIN pin is 60V for non-repetitive
1 second transients and 36V for continuous operation.
Note 3. The LT3509E is guaranteed to meet performance specifications from
0°C to 125°C junction temperature. Specifications over the –40°C to 125°C
operating junction temperature range are assured by design, characterization
and correlation with statistical process controls. The LT3509I is guaranteed
over the full –40°C to 125°C temperature range. The LT3509H is guaranteed
80
V
MHz
kHz
MHz
V
150
ns
over the full –40°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes. Operating lifetime is
derated at junction temperatures greater than 125°C.
Note 4. Current flows out of pin.
Note 5. Switch Saturation Voltage is guaranteed by design.
Note 6. This IC includes overtemperature protection that is intended to protect
the device during momentary overload conditions. Junction temperature will
exceed the maximum operating temperature when overtemperature protection
is active. Continuous operation above the specified maximum operating
junction temperature may impair device reliability.
3509fd
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3
LT3509
Typical Performance Characteristics
Efficiency vs Load Current
VOUT = 5V, fSW = 2.0MHz
95
95
TA = 25ºC
85
EFFICIENCY (%)
80
VIN = 24V
75
70
65
0
0.4
0.2
0.6
75
70
75
65
60
60
55
0
0.2
0.4
0.6
VIN = 12V
70
65
50
0.8
0
0.4
0.2
ILOAD(A)
ILOAD(A)
0.6
0.8
ILOAD(A)
3509 G02
3509 G01
Switch VCE(SAT) vs ISW
0.35
80
VIN = 12V
80
55
0.8
TA = 25ºC
85
EFFICIENCY (%)
VIN = 12V
85
EFFICIENCY (%)
90
TA = 25ºC
90
90
60
Efficiency vs Load Current
VOUT = 1.8V, fSW = 0.7MHz
Efficiency vs Load Current
VOUT = 3.3V, fSW = 2.0MHz
3509 G03
IBOOST vs ISW
25
TA = 25ºC
TA = 25ºC
0.3
20
IBOOST (mA)
VCE(SAT) (V)
0.25
0.2
0.15
15
10
0.1
5
0.05
0
0
0.2
0.4
0.6
ISW(A)
0.8
0
1.0
0
0.2
0.4
0.6
ISW(A)
3509 G04
1
3509 G05
Boost Diode Characteristics
1.2
0.8
2.2
TA = 25ºC
Frequency vs RT
TA = 25ºC
2.0
FREQUENCY (MHz)
1
Vf (V)
0.8
0.6
0.4
1.0
0.5
0.2
0
1.5
0
50
100
BOOST DIODE CURRENT (mA)
150
0
0
20
3509 G06
4
40
60
80 100 120 140 160 180
RT(kΩ)
3509 G07
3509fd
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LT3509
Typical Performance Characteristics
fSW vs Temperature
(Measured at 1MHz)
FB Pin Voltage vs Temperature
0.810
1.04
30
0.800
0.795
1.02
1.01
1.00
0.99
0
0.97
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
45
160
TA = 25ºC
140
TA = 85ºC
120
MINIMUM ON-TIME (ns)
MAXIMUM VIN (V)
30
25
20
15
10
10
0
0.2
0.4
ILOAD (A)
0.6
0.8
3509 G10
Minimum On-Time vs
Temperature, ILOAD = 0.3A
100
80
60
40
20
5
0
15
3509 G09
Max VIN for Constant Frequency
VOUT = 5V, fSW = 2MHz
35
20
0
25 50 75 100 125 150
TEMPERATURE (˚C)
3509 G08
40
TA = 85ºC
5
0.98
0.790
–50 –25
TA = 25ºC
25
MAXIMUM VIN (V)
NORMALIZED (fSW)
0.805
0
0.2
0.4
0.6
0
–50 –25
0.8
ILOAD (A)
0
25 50 75 100 125 150
TEMPERATURE (°C)
3509 G12
3509 G11
ILIM vs Temperature
ILIM vs Duty Cycle
1.5
1.8
TA = 25ºC
1.6
1.3
SWITCH
1.4
1.2
1.0
DA
ILIM (A)
ILIM (A)
FEEDBACK PIN VOLTAGE (V)
1.03
Max VIN for Constant Frequency
VOUT = 3.3V, fSW = 2MHz
0.9
1
0.8
0.6
0.4
0.7
0.2
0.5
–50 –25
0
25 50
75 100 125 150
TEMPERATURE (°C)
0
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
3509 G13
3509 G14
3509fd
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5
LT3509
Pin Functions
(DFN/MSOP)
DA1, DA2 (Pins 1, 7/Pins 1, 8): The DA pins are the anode
connections for the catch diodes. These are connected
internally to the exposed ground pad by current sensing
resistors.
BOOST1, BOOST2 (Pins 2, 6/Pins 2, 7): The BOOST pins
are used to dynamically boost the power transistor base
above VIN to minimize the voltage drop and power loss in
the switch. These should be tied to the associated switch
pins through the boost capacitors.
SW1, SW2 (Pins 3, 5/Pins 3, 6): The SW pins are the
internal power switch outputs. These should be connected
to the associated inductors, catch diode cathodes, and the
boost capacitors.
VIN (Pin 4/Pins 4, 5): The VIN pins supply power to the
internal power switches and control circuitry. In the MSE
package the VIN pins must be tied together. The input
capacitor should be placed as close as possible to the
supply pins.
FB1, FB2 (Pins 14, 8/Pins 16, 9): The FB pins are used
to set the regulated output voltage relative to the internal
reference. These pins should be connected to a resistor
divider from the regulated output such that the FB pin is
at 0.8V when the output is at the desired voltage.
RUN/SS1, RUN/SS2 (Pins 13, 9/Pins 15, 10): The RUN/SS
pins enable the associated regulator channel. If both pins
are pulled to ground, the device will shut-down to a low
power state. In the range 0.8V to 2V, the regulators are
enabled but the peak switch current and the DA pin maximum current are limited to provide a soft-start function.
Above 2V, the full output current is available. The inputs
incorporate a 1µA pull-up so that they will float high or
charge an external capacitor to provide a current limited
soft-start. The pins are pulled down by approximately 250µA
6
in the case of overvoltage or overtemperature conditions
in order to discharge the soft-start capacitors. The pins
can also be driven by a logic control signal of up to 5.0V.
In this case, it is necessary place a 10k to 50k resistor
in series along with a capacitor from the RUN/SS pin to
ground to ensure that there will be a soft-start for both
initial turn on and in the case of fault conditions. Do not
tie these pins to VIN.
RT (Pin 10/Pin 11): The RT pin is used to set the internal
oscillator frequency. A 40.2k resistor from RT to ground
results in a nominal frequency of 1MHz.
SYNC (Pin 11/Pin12): The SYNC pin allows the switching
frequency to be synchronized to a external clock. Choose
RT resistor to set a free-run frequency at least 12% less
than the external clock frequency for correct operation.
The SYNC pin should not be allowed to float; if not used,
it should be tied low through a resistance 10kΩ or less.
BD (Pin 12/Pin 13): The BD pin is common anode connection of the internal Schottky boost diodes. This provides
the power for charging the BOOST capacitors. It should
be locally bypassed for best performance.
Exposed Pad (Pin 15/Pin 17): GND. This is the reference
and supply ground for the regulator. The exposed pad
must be soldered to the PCB and electrically connected to
supply ground. Use a large ground plane and thermal vias
to optimize thermal performance. The current in the catch
diodes also flows through the GND pad to the DA pins.
AGND (Pin 14, MSOP Package Only): This is the connected
to the ground connection of the chip and may be used as a
separate return for the low current control side components.
It should not be used as the only ground connection or as
a connection return for load side components.
3509fd
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LT3509
Block Diagram
COMMON
CIRCUITRY
1 OF 2 REGULATOR CHANNELS SHOWN
VIN
BD
C1
OVERVOLTAGE
DETECT
BOOST
RUN/SS1
MAIN CURRENT
COMPARATOR
SHUTDOWN
AND
SOFT-START
CONTROL
C3
VREF AND
CORE
VOLTAGE
REGULATOR
SYNC
RT
RT
SWITCH
LOGIC
RUN/SS2
C2
NOTE: THE BD PIN
IS COMMON TO
BOTH CHANNELS
BOOST
DIODE
SWITCH
DRIVER
DA CURRENT
COMPARATOR
–
POWER
SWITCH
ERROR
AMPLIFIER
OSCILLATOR
L1
SW
VREF
0.8V
VOUT
D1
–17mV
DA
VC
SLOPE
C4
C5
R1
18mΩ
FB
R2
CLAMP
GND
3509 BD
Figure 1. Functional Block Diagram
3509fd
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7
LT3509
Operation
Overview
The LT3509 is a dual, constant frequency, current mode
switching regulator with internal power switches. The two
independent channels share a common voltage reference
and oscillator and operate in phase. The switching frequency
is set by a single resistor and can also be synchronized to
an external clock. Operation can be best understood by
referring to the Block Diagram (Figure 1).
Startup and Shutdown
When the RUN/SS[1,2] pins are pulled low (50%.
Current Limiting
Current mode control provides cycle-by-cycle current
limiting by means of a clamp on the maximum current that
can be provided by the switch. A comparator monitors the
current flowing through the catch diode via the DA pin.
This comparator delays switching if the diode current is
higher than 0.95A (typical). This current level is indicative
of a fault condition such as a shorted output with a high
input voltage. Switching will only resume once the diode
current has fallen below the 0.95A limit. This way the DA
comparator regulates the valley current of the inductor to
0.95A during a short circuit. This will ensure the part will
survive a short-circuit event.
Over and Undervoltage Shutdown
A basic undervoltage lockout prevents switching if VIN
is below 3.3V (typical). The overvoltage shutdown stops
the part from switching when VIN is greater than 38.5V
(typical). This protects the device and its load during
momentary overvoltage events. After the input voltage
falls below 38.5V, the part initiates a soft start sequence
and resumes switching.
BOOST Circuit
To ensure best efficiency and minimum dropout voltage
the output transistor base drive is boosted above VIN by
the external boost capacitors (C4). When the SW pin is
low the capacitors are charged via the BOOST diodes and
the supply on BD.
3509fd
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LT3509
APPLICATIONS INFORMATION
Shutdown and Soft Start
Setting The Output Voltage
When the RUN/SS pins are pulled to ground, the part will
shut down to its lowest current state of approximately
9µA. If driving a large capacitive load it may be desirable
to use the current limiting soft-start feature. Connecting
capacitors to ground from the RUN/SS pins will control
the delay until full current is available. The pull-up current
is 1µA and the full current threshold is 2V so the start-up
time is given by:
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the resistors
according to:
T = 2 • C • 106 s
For example a 0.005µF capacitor will give a time to full
current of 10ms. If both outputs can come up together
then the two inputs can be paralleled and tied to one capacitor. In this case use twice the capacitor value to obtain
the same start-up time. During the soft-start time both
the peak current threshold and the DA current threshold
will track so the part will skip pulses as required to limit
the maximum inductor current. Starting up into a large
capacitor is not much different to starting into a shortcircuit in this respect.
VSW
10V/DIV
IL
0.2A/DIV
VOUT
5V/DIV
TIME 1ms/DIV
Figure 2. Soft-Start
3509 F02
R1=R2 •
VOUT
–1
0.8
The designators correspond to Figure 1. R2 should be 20k
or less to avoid bias current errors.
Frequency Setting
The timing resistor, RT , for any desired frequency in the
range 264kHz to 2.2MHz can be calculated from the
following formula:
1.215
RT =
– 0.215 • 40.2
fSW
where fSW is in MHz and RT is in kΩ.
Table 1. Standard E96 Resistors for Common Frequencies
FREQUENCY
TIMING RESISTOR RT (kΩ)
264 kHz
178
300 kHz
154
400kHz
113
500kHz
88.7
1MHz
40.2
2MHz
15.8
2.2MHz
13.7
Note: The device is specified for operation down to 300kHz. The 264kHz
value is to allow external synchronization at 300kHz
3509fd
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9
LT3509
APPLICATIONS INFORMATION
External Synchronization
The external synchronization provides a trigger to the
internal oscillator. As such, it can only raise the frequency
above the free-run value. To allow for device and
component tolerances, the free run frequency should be
set to at least 12% lower than the lowest supplied external
synchronization reference. The oscillator and hence the
switching frequency can then pushed up from 12% above
the free-run frequency, set by the selected RT. For example,
if the minimum external clock is 300kHz, the RT should
be chosen for 264kHz.
The SYNC input has a threshold of 1.0V nominal so it is
compatible with most logic levels. The duty cycle is not
critical provided the high or low pulse width is at least
80ns. If not used, the SYNC input should be tied low with
10kΩ less to avoid noise pickup.
Design Procedure
Before starting detailed design a number of key design
parameters should be established as these may affect
design decisions and component choices along the way.
One of the main things to determine apart from the desired
output voltages is the input voltage range. Both the normal
operating range and the extreme conditions of surges
and/or dips or brown-outs need to be known. Then the
operating frequency should be considered and if there
are particular requirements to avoid interference. If there
are very specific frequencies that need to be avoided then
external synchronization may be needed. This could also be
desirable if multiple switchers are used as low frequency
beating between similar devices can be undesirable. For
efficient operation this converter requires a boost supply so
that the base of the output transistor can be pumped above
10
the input voltage during the switch on time. Depending
on the input and output voltages the boost supply can be
provided by the input voltage, one of the regulated outputs
or an independent supply such as an LDO.
Input Voltage Range
Firstly, the LT3509 imposes some hard limits due to the
undervoltage lock-out and the overvoltage protection. A
given application will also have a reduced, normal operating
range over which maximum efficiency and lowest ripple
are obtained. This usually requires that the device is
operating at a fixed frequency without skipping pulses.
There may also be zones above and below the normal
range where regulation is maintained but efficiency and
ripple may be compromised. At the low end, insufficient
input voltage will cause loss of regulation and increased
ripple—this is the dropout range. At the high end if the
duty cycle becomes too low this will cause pulse skipping
and excessive ripple. This is the pulse‑skip region. Both
situations also lead to higher noise at frequencies other than
the chosen switching frequency. Occasional excursions
into pulse-skip mode, during surges for example, may be
tolerable. Pulse skipping will also occur at light loads even
within the normal operating range but ripple is usually not
degraded because at light load the output capacitor can
hold the voltage steady between pulses.
For input voltages greater than 30V, there are restrictions
on the inductor value. See the Inductor Selection section
for details.
To ensure the regulator is operating in continuous mode
it is necessary to calculate the duty cycle for the required
output voltage over the full input voltage range. This must
then be compared with minimum and maximum practical
duty cycles.
3509fd
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LT3509
APPLICATIONS INFORMATION
In any step-down switcher the duty cycle when operating
in continuous, or fixed frequency, mode is dependent
on the step-down ratio. This is because for a constant
average load current the decay of the inductor current
when the switch is off must match the increase in inductor
current when the switch is on. The can be estimated by
the following formula:
DC =
where:
VOUT + VF
VIN − VSW + VF
The minimum on time increases with increasing temperature so the value for the maximum operating temperature
should be used. See the Minimum On-Time vs Temperature
graph in the Typical Performance Characteristics.
The maximum input voltage for this duty cycle is given by:
DC = Duty Cycle (Fraction of Cycle when Switch is On)
VOUT = Output Voltage
VIN = Input Voltage
VIN(MAX ) =
VOUT + VF
− VF + VSW
DCMIN
Above this voltage the only way the LT3509 can maintain
regulation is to skip cycles so the effective frequency will
reduce. This will cause an increase in ripple and the switching noise will shift to a lower frequency. This calculation
will in practice drive the maximum switching frequency
for a desired step-down ratio.
VF = Catch Diode Forward Voltage
VSW = Switch Voltage Drop
Note: This formula neglects switching and inductor losses
so in practice the duty cycle may be slightly higher.
It is clear from this equation that the duty cycle will approach
100% as the input voltage is reduced and become smaller
as the input voltage increases. There are practical limits to
the minimum and maximum duty cycles for continuous
operation due to the switch minimum off and on times.
These are independent of operating frequency so it is clear
that range of usable duty cycle is inverserly proportional to
frequency. Therefore at higher frequency the input voltage
range (for constant frequency operation) will narrow.
The minimum duty cycle is given by:
DCMIN = fSW • t ON(MIN)
VOUT
100mV/DIV
(AC COUPLED)
IL
0.5A/DIV
TIME 1µs/DIV
3509 F03
Figure 3. Continuous Mode
VOUT
100mV/DIV
(AC COUPLED)
IL
0.5A/DIV
where:
fSW = Switching Frequency
TIME 1µs/DIV
tON(MIN) = Switch Minimum On-Time
3509 F04
Figure 4. Pulse Skipping
3509fd
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11
LT3509
APPLICATIONS INFORMATION
Minimum Input Voltage and Boost Architecture
The minimum operating voltage is determined either by
the LT3509’s internal undervoltage lockout of ~3.6V or
by its maximum duty cycle. The maximum duty cycle for
fixed frequency operation is given by:
DCMAX = 1− tOFF(MIN) • fSW
It follows that:
VIN(MIN) =
VOUT + VF
− VF + VSW
DCMAX
If a reduction in switching frequency can be tolerated the
minimum input voltage can drop to just above output
voltage. Not only is the output transistor base pumped
above the input voltage by the boost capacitor, the
switch can remain on through multiple switching cycles
resulting in a high effective duty cycle. Thus, this is a
true low dropout regulator. As it is necessary to recharge
the boost capacitor from time to time, a minimum width
off-cycle will be forced occasionally to maintain the charge.
Depending on the operating frequency, the duty cycle can
reach 97% to 98%, although at this point the output pulses
will be at a sub-multiple of the programmed frequency.
One other consideration is that at very light loads or no
load the part will go into pulse skipping mode. The part
will then have trouble getting enough voltage on to the
boost capacitors to fully saturate the switch. This is most
problematic when the BD pin is supplied from the regulated
output. The net result is that a higher input voltage will be
required to start up the boost system. The typical minimum
input voltage over a range of loads is shown in Figure 5
for 3.3V and Figure 6 for 5V.
When operating at such high duty cycles the peak currents
in the boost diodes are greater and this will require a the
BD supply to be somewhat higher than would be required
at less extreme duty cycles. If operation at low input/output
ratios and low BD supply voltages is required it may be
desirable to augment the internal boost diodes with external
discrete diodes in parallel.
Boost Pin Considerations
The boost capacitor, in conjunction with the internal boost
diode, provides a bootstrapped supply for the power switch
that is above the input voltage. For operation at 1MHz and
above and at reasonable duty cycles a 0.1µF capacitor
will work well. For operation at lower frequencies and/or
higher duty cycles something larger may be needed. A
good rule of thumb is:
C BOOST =
where fSW is in MHz and CBOOST is in µF
7
5.5
5
TO START
VIN TO START (V)
VIN TO START (V)
TO RUN
3.5
3
6
TO RUN
5.5
5
4.5
2.5
2
0.001
TO START
6.5
4.5
4
1
10 • fSW
0.1
0.01
LOAD CURRENT (A)
1
4
0.001
3509 F05
Figure 5. Minimum VIN for 3.3V VOUT
12
0.1
0.01
LOAD CURRENT (A)
1
3509 F06
Figure 6. Minimum VIN for 5V VOUT
3509fd
For more information www.linear.com/LT3509
LT3509
APPLICATIONS INFORMATION
Boost Pin Considerations
Figure 7 through Figure 9 show several ways to arrange
the boost circuit. The BOOST pin must be more than 2V
above the SW pin for full efficiency. For outputs of 3.3V
and higher, the standard circuit Figure 7 is best. For lower
output voltages, the boost diode can be tied to the input
Figure 8. The circuit in Figure 7 is more efficient because
the boost pin current comes from a lower voltage source.
Finally, as shown in Figure 9, the BD pin can be tied to
another source that is at least 3V. For example, if you are
generating 3.3V and 1.8V, and the 3.3V is on whenever
the 1.8V is on, the 1.8V boost diode can be connected to
the 3.3V output.
VIN
BD
VIN
BOOST
LT3509
CBOOST
L1
VOUT
SW
CIN
D1
GND
COUT
DA
3509 F07
VBOOST – VSW ≅ VOUT
MAX VBOOST ≅ VIN + VOUT
VOUT ≥ 3V
Figure 7. BD Tied to Regulated Output
BD
BOOST
VIN
VIN
A good first choice for the inductor value is:
L = ( VOUT + VF ) •
VOUT
D1
DA
2.1MHz
fSW
where VF is the voltage drop of the catch diode (~0.5V)
and L is in µH.
The inductor’s RMS current rating must be greater than the
maximum load current and its saturation current should
be at least 30% higher. For highest efficiency, the series
resistance (DCR) should be less than 0.15Ω. Table 2 lists
several vendors and types that are suitable.
The current in the inductor is a triangle wave with an average
value equal to the load current. The peak switch current
is equal to the output current plus half the peak-to-peak
inductor ripple current. The LT3509 limits its switch current
in order to protect itself and the system from overcurrent
faults. Therefore, the maximum output current that the
LT3509 will deliver depends on the switch current limit,
the inductor value and the input and output voltages.
VBD
BD
VIN
VIN
BOOST
LT3509
L1
SW
GND
Inductor Selection and Maximum Output Current
CBOOST
LT3509
CIN
In any case, be sure that the maximum voltage at the
BOOST pin is less than 60V and the voltage difference
between the BOOST and SW pins is less than 30V.
L1
VOUT
SW
CIN
D1
GND
COUT
CBOOST
DA
COUT
3509 F09
3509 F08
VBOOST – VSW ≅ VIN
MAX VBOOST ≅ 2VIN
VBOOST – VSW ≅ VBD
MAX VBOOST ≅ VIN + VBD
VBD ≥ 3V
Figure 9. Separate Boost Supply
Figure 8. Supplied from VIN
3509fd
For more information www.linear.com/LT3509
13
LT3509
APPLICATIONS INFORMATION
When the switch is off, the potential across the inductor is
the output voltage plus the catch diode forward voltage. This
gives the peak-to-peak ripple current in the inductor:
∆IL =(1– DC)
where:
VOUT + VF
L • fSW
DC = Duty Cycle
fSW = Switching Frequency
L = Inductor Value
VF = Diode Forward Voltage
The peak inductor and switch current is:
ISWPK =ILPK =IOUT +
∆IL
2
To maintain output regulation, this peak current must be
less than the LT3509’s switch current limit ILIM. This is
dependent on duty cycle due to the slope compensation.
For ILIM is at least 1.4A at low duty cycles and decreases
linearly to 1.0A at DC = 0.8.
The theoretical minimum inductance can now be calculated as:
LMIN =
1– DCMIN VOUT + VF
•
ILIM – IOUT
f
where DCMIN is the minimum duty cycle called for by the
application i.e.:
DCMIN =
14
There is a limit to the actual minimum duty cycle imposed
by the minimum on-time of the switch. For a robust design
it is important that inductor that will not saturate when
the switch is at its minimum on-time, the input voltage
is at maximum and the output is short circuited. In this
case the full input voltage, less the drop in the switch, will
appear across the inductor. This doesn’t require an actual
short, just starting into a capacitive load will provide the
same conditions. The Diode current sensing scheme will
ensure that the switch will not turn-on if the inductor
current is above the DA current limit threshold, which has
a maximum of 1.1A. The peak current under short-circuit
conditions can then be calculated from:
IPEAK =
VIN • tON(MIN)
L
+ 1.1A
The inductor should have a saturation current greater than
this value. For safe operation with high input voltages this
can often mean using a physically larger inductor as higher
value inductors often have lower saturation currents for
a given core size. As a general rule the saturation current
should be at least 1.8A to be short-circuit proof. However, it’s
generally better to use an inductor larger than the minimum
value. For robust operation at input voltages greater than
30V, use an inductor with a value of 4.2µH or greater, and
a saturation current rating of 1.8A or higher. The minimum
inductor has large ripple currents which increase core
losses and require large output capacitors to keep output
voltage ripple low. Select an inductor greater than LMIN
that keeps the ripple current below 30% of ILIM.
VOUT(MAX ) + VF
VIN(MIN) – VSW + VF
3509fd
For more information www.linear.com/LT3509
LT3509
APPLICATIONS INFORMATION
Table 2. Recommended Inductors
MANUFACTURER/
PART NUMBER
VALUE
(µH)
ISAT
(A)
DCR
(W)
HEIGHT
(mm)
LPS4018-222ML
2.2
2.8
0.07
1.7
LPS5030-332ML
3.3
2.5
0.066
2.9
LPS5030-472ML
4.7
2.5
0.083
2.9
LPS6225-682ML
6.8
2.7
0.095
2.4
LPS6225-103ML
10
2.1
0.105
2.4
CDRH4D22/HP-2R2N
2.2
3.2
0.0035
2.4
CDRH4D22/HP-3R5N
3.5
2.5
0.052
2.4
CDRH4D22/HP-4R7N
4.7
2.2
0.066
2.4
CDRH5D28/HP-6R8N
6.8
3.1
0.049
3.0
CDRH5D28/HP-8R2N
8.2
2.7
0.071
3.0
CDRH5D28R/HP-100N
10
2.45
0.074
3.0
SD52-2R2-R
2.2
2.30
0.0385
2.0
SD52-3R5-R
3.5
1.82
0.0503
2.0
SD52-4R7-R
4.7
1.64
0.0568
2.0
SD6030-5R8-R
5.8
1.8
0.045
3.0
SD7030-8R0-R
8.0
1.85
0.058
3.0
SD7030-100-R
10.0
1.7
0.065
3.0
A997AS-2R2N
2.2
1.6
0.06
1.8
A997AS-3R3N
3.3
1.2
0.07
1.8
A997AS-4R7M
4.7
1.07
0.1
1.8
7447745022
2.2
3.5
0.036
2.0
7447745033
3.3
3.0
0.045
2.0
7447745047
4.7
2.4
0.057
2.0
7447745076
7.6
1.8
0.095
2.0
7447445100
10
1.6
0.12
2.0
Coilcraft
Sumida
Cooper
Toko
Würth
The prior analysis is valid for continuous mode operation
(IOUT > ∆ILIM / 2). For details of maximum output current
in discontinuous mode operation, see Linear Technology’s
Application Note 44. Finally, for duty cycles greater than
50% (VOUT/VIN > 0.5), a minimum inductance is required
to avoid subharmonic oscillations. This minimum inductance is
L MIN = ( VOUT + VF )•
1.4
fSW
where fSW is in MHz and LMIN is in µH.
If using external synchronization, calculate LMIN using the
RT frequency and not the SYNC frequency.
Frequency Compensation
The LT3509 uses current mode control to regulate the
output, which simplifies loop compensation and allows
the necessary filter components to be integrated. The fixed
internal compensation network has been chosen to give
stable operation over a wide range of operating conditions
but assumes a minimum load capacitance. The LT3509
does not depend on the ESR of the output capacitor for
stability so the designer is free to use ceramic capacitors
to achieve low output ripple and small PCB footprint.
Figure 10 shows an equivalent circuit for the LT3509 control
loop. The error amp is a transconductance amplifier with
finite output impedance. The power section, consisting of
the modulator, power switch and inductor is modeled as a
transconductance amplifier generating an output current
proportional to the voltage at the COMP-NODE. The gain
of the power stage (gmp) is 1.1S. Note that the output
capacitor integrates this current and that the internal
capacitor integrates the error amplifier output current,
resulting in two poles in the loop. In most cases, a zero is
required and comes either from the output capacitor ESR
3509fd
For more information www.linear.com/LT3509
15
LT3509
APPLICATIONS INFORMATION
or from RC. This model works well as long as the inductor
current ripple is not too low (∆IRIPPLE > 5% IOUT) and the
loop crossover frequency is less than fSW/5. An optional
phase lead capacitor (CPL) across the feedback divider
may improve the transient response.
LT3509
1.1S
VOUT
VIN
CPL
RC
260µS
R1
+
–
COMPNODE
75k
COUT
VREF = 0.8V
95pF
1.73M
R2
3509 F10
Figure 10. Small-Signal Equivalent Circuit
Output Capacitor Selection
The output capacitor filters the inductor current to generate
an output with low voltage ripple. It also stores energy in
order to satisfy transient loads and stabilize the LT3509’s
control loop. Because the LT3509 operates at a high
frequency, minimal output capacitance is necessary. In
addition, the control loop operates well with or without
the presence of output capacitor series resistance (ESR).
Ceramic capacitors, which achieve very low output ripple
and small circuit size, are therefore an option.
16
You can estimate output ripple with the following
equations.
For ceramic capacitors where low capacitance value is
more significant than ESR:
VRIPPLE = ∆IL / (8 • fSW • COUT )
For electrolytic capacitors where ESR is high relative to
capacitive reactance:
VRIPPLE = ∆IL • ESR
where ∆IL is the peak-to-peak ripple current in the inductor.
The RMS content of this ripple is very low so the RMS
current rating of the output capacitor is usually not of
concern. It can be estimated with the formula:
IC(RMS) = ∆IL / 12
Another constraint on the output capacitor is that it must
have greater energy storage than the inductor; if the stored
energy in the inductor transfers to the output, the resulting
voltage step should be small compared to the regulation
voltage. For a 5% overshoot, this requirement indicates:
COUT > 10 • L •(ILIM / VOUT )2
The low ESR and small size of ceramic capacitors make
them the preferred type for LT3509 applications. Not all
ceramic capacitors are the same, however. Many of the
higher value capacitors use poor dielectrics with high
temperature and voltage coefficients. In particular, Y5V
3509fd
For more information www.linear.com/LT3509
LT3509
APPLICATIONS INFORMATION
and Z5U types lose a large fraction of their capacitance
with applied voltage and at temperature extremes. Because
loop stability and transient response depend on the value
of COUT, this loss may be unacceptable. Use X7R and X5R
types.
the top feedback resistor. The small-signal model shown
in Figure 10 can be used to model this in a simulator or
to give insight to an empirical design. Figure 11 shows
some load step responses with differing output capacitors
and CPL combinations.
The value of the output capacitor greatly affects the
transient response to a load step. It has to supply extra
current demand or absorb excess current delivery until
the feedback loop can respond. The loop response is
dependent on the error amplifier transconductance, the
internal compensation capacitor and the feedback network. Higher output voltages necessarily require a larger
feedback divider ratio. This will also reduce the loop gain
and slow the response time. Fortunately this effect can be
mitigated by use of a feed-forward capacitor, CPL, across
Input Capacitor
The input capacitor needs to supply the pulses of charge
demanded during the on time of the switches. Little total
capacitance is required as a few hundred millivolts of ripple
at the VIN pin will not cause any problems to the device.
When operating at 2MHz and 12V, 2µF will work well. At
the lowest operating frequency and/or at low input voltages
a larger capacitor such as 4.7µF is preferred.
ILOAD
700mA
300mA
ILOAD
700mA
300mA
VOUT (AC)
50mV/DIV
VOUT (AC)
50mV/DIV
COUT = 10µF
CPL = 0
TIME 20µs/DIV
COUT = 10µF
CPL = 82pF
TIME 20µs/DIV
3509 F11
Figure 11. Transient Load Response with Different Combinations
of COUT and CPL Load Current Step from 300mA to 700mA
R1 = 10k, R2 = 32.4k, VIN = 12V, VOUT = 3.3V, fSW = 2.0MHz
3509fd
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17
LT3509
APPLICATIONS INFORMATION
Diode Selection
The catch diode (D1 from Figure 1) conducts current only
during switch off time. Average forward current in normal
operation can be calculated from:
ID( AVG) = IOUT ( VIN – VOUT ) / VIN
The only reason to consider a diode with a larger current
rating than necessary for nominal operation is for the
worst-case condition of shorted output. The diode current
will then increase to the typical peak switch current limit.
If transient input voltages exceed 40V, use a Schottky
diode with a reverse voltage rating of 45V or higher. If
the maximum transient input voltage is under 40V, use a
Schottky diode with a reverse voltage rating greater than
the maximum input voltage. Table 3 lists several Schottky
diodes and their manufacturers:
Table 3. Schottky Diodes
MANUFACTURER/
PART NUMBER
VR
(V)
IAVE
(A)
VF at 1A
(mV)
40
1
550
40
1
450
On Semiconductor
MBRM140
MicroSemi
UPS140
Diodes Inc.
DFLS140L
40
1
550
1N5819HW
40
1
450
providing the boost supply to the BD pin. In this case the
voltage drop of the other switch will increase and lower
the efficiency. This could eventually cause the part to reach
the thermal shutdown limit. One other important feature
of the part that needs to be considered is that there is a
parasitic diode in parallel with the power switch. In normal
operation this is reverse biased but it could conduct if the
load can be powered from an alternate source when the
LT3509 has no input. This may occur in battery charging
applications or in battery backup systems where a battery or some other supply is diode ORed with one of the
LT3509 regulated outputs. If the SW pin is at more than
about 4V the VIN pin can attain sufficient voltage for LT3509
control circuitry to power-up to the quiescent bias level
and up to 2mA could be drawn from the backup supply.
This can be minimized if some discrete FETs or open-drain
buffers are used to pull down the RUN/SS pins. Of course
the gates need to be driven from the standby or battery
backed supply. If there is the possibility of a short circuit
at the input or just other parallel circuits connected to VIN
it would be best to add a protection diode in series with
VIN. This will also protect against a reversed input polarity.
These concepts are illustrated in Figure 12.
VIN
D2
BD
VIN
CIN
BOOST
RUN/SS1
Short and Reverse Protection
VOUT
SW
D1
Provided the inductors are chosen to not go deep into
their saturation region at the maximum ILIMIT current the
LT3509 will tolerate a short circuit on one or both outputs.
The excess current in the inductor will be detected by the
DA comparator and the frequency will reduced until the
valley current is below the limit. This shouldn’t affect the
other channel unless the channel that is shorted is also
18
CBOOST
L1
LT3509
RUN/SS2
GND
COUT
DA
SLEEP
3509 F12
Figure 12. Reverse Bias Protection
3509fd
For more information www.linear.com/LT3509
LT3509
APPLICATIONS INFORMATION
Hot Plugging Considerations
The small size, reliability and low impedance of ceramic
capacitors make them attractive for the input capacitor.
Unfortunately they can be hazardous to semiconductor
devices if combined with an inductive supply loop and a
fast power transition such as through a mechanical switch
or connector. The low loss ceramic capacitor combined
with the just a small amount of wiring inductance forms
an underdamped resonant tank circuit and the voltage at
the VIN pin of the LT3509 can ring to twice the nominal
input voltage. See Linear Technology Application Note 88
for more details.
PCB Layout and Thermal Design
The PCB layout is critical to both the electrical and thermal
performance of the LT3509. Most important is the connection to the Exposed Pad which provides the main ground
connection and also a thermal path for cooling the chip.
This must be soldered to a topside copper plane which
is also tied to backside and/or internal plane(s) with an
array of thermal vias.
• The loop from the regulated outputs through the output
capacitor back to the ground plane. Excess impedance
here will result in excessive ripple at the output.
The area of the SW and BOOST nodes should as small as
possible. Also the feedback components should be placed
as close as possible to the FB pins so that the traces are
short and shielded from the SW and BOOST nodes by the
ground planes.
Figure 13 shows a detail view of a practical board layout
showing just the top layer. The complete board is somewhat
larger at 7.5cm × 7.5cm. The device has been evaluated
on this board in still air running at 700kHz switching frequency. One channel was set to 5V and the other to 3.3V
and both channels were fully loaded to 700mA. The device
temperature reached approximately 15°C above ambient
for input voltages below 12V. At 24V input it was slightly
higher at 17°C above ambient.
To obtain the best electrical performance particular
attention should be paid to keeping the following current
paths short:
• The loop from the VIN pin through the input capacitor
back to the ground pad and plane. This sees high di/dt
transitions as the power switches turn on and off. Excess impedance will degrade the minimum usable input
voltage and could cause crosstalk between channels.
• The loops from the switch pins to the catch diodes and
back to the DA pins. The fast changing currents and
voltage here combined with long PCB traces will cause
ringing on the switch pin and may result in unwelcome
EMI.
Figure 13. Sample PCB Layout (Top Layer Only)
3509fd
For more information www.linear.com/LT3509
19
LT3509
Typical Applications
1.8V and 3.3V Outputs, Synchronized to 300kHz to 600kHz
VIN
4.5V TO 36V
(TRANSIENT TO 60V)
2.2µF
BD
BOOST1
BOOST2
0.22µF 15µH
0.22µF
10µH
VOUT
1.8V
0.7A
VIN
SW1
LT3509
UPS140
12.4k
CLOCK
22µF
1.6V
UPS140
DA1
DA2
FB1
FB2
31.6k
RUN/SS1 RUN/SS2
10k
SYNC
22nF
VOUT
3.3V
0.7A
SW2
RT
10k
GND
22nF
22µF
178k
0.4V
3509 TA03
NOTE: RT CHOSEN FOR 264kHz
Automotive Accessory Application
5V Logic Supply and 8V for LCD Display with Display Power Controlled by Logic
VIN
9.4V TO 36V
2.2µF
VOUT
5V
0.7A
VIN
BD
BOOST1
BOOST2
0.22µF 10µH
0.22µF
6.8µH
SW1
LT3509
DFLS140L
52.3k
DFLS140L
DA1
DA2
FB1
FB2
RUN/SS1 RUN/SS2
SYNC RT
10k
10µF
DISPLAY POWER
CONTROL
0V = OFF
3.3V = ON
20
22nF
VOUT
8V
0.7A
SW2
GND
40.2k
90.9k
10k
0.1µF
10k
10µF
3509 TA04
fSW = 1MHz
3509fd
For more information www.linear.com/LT3509
LT3509
package description
Please refer to http://www.linear.com/product/LT3509#packaging for the most recent package drawings.
DE Package
14-Lead Plastic DFN (4mm × 3mm)
(Reference LTC DWG # 05-08-1708 Rev B)
0.70 ±0.05
3.30 ±0.05
3.60 ±0.05
2.20 ±0.05
1.70 ±0.05
PACKAGE
OUTLINE
0.25 ±0.05
0.50 BSC
3.00 REF
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
4.00 ±0.10
(2 SIDES)
R = 0.05
TYP
3.00 ±0.10
(2 SIDES)
R = 0.115
TYP
8
0.40 ±0.10
14
3.30 ±0.10
1.70 ±0.10
PIN 1 NOTCH
R = 0.20 OR
0.35 × 45°
CHAMFER
PIN 1
TOP MARK
(SEE NOTE 6)
0.200 REF
0.75 ±0.05
(DE14) DFN 0806 REV B
7
1
0.25 ±0.05
0.50 BSC
3.00 REF
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING PROPOSED TO BE MADE VARIATION OF VERSION (WGED-3) IN JEDEC
PACKAGE OUTLINE MO-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
3509fd
For more information www.linear.com/LT3509
21
LT3509
package description
Please refer to http://www.linear.com/product/LT3509#packaging for the most recent package drawings.
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev F)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
5.10
(.201)
MIN
2.845 ±0.102
(.112 ±.004)
0.889 ±0.127
(.035 ±.005)
8
1
1.651 ±0.102
(.065 ±.004)
1.651 ±0.102 3.20 – 3.45
(.065 ±.004) (.126 – .136)
0.305 ±0.038
(.0120 ±.0015)
TYP
16
0.50
(.0197)
BSC
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
9
NO MEASUREMENT PURPOSE
0.280 ±0.076
(.011 ±.003)
REF
16151413121110 9
DETAIL “A”
0° – 6° TYP
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
4.90 ±0.152
(.193 ±.006)
GAUGE PLANE
0.53 ±0.152
(.021 ±.006)
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
1234567 8
0.17 – 0.27
(.007 – .011)
TYP
0.50
NOTE:
(.0197)
1. DIMENSIONS IN MILLIMETER/(INCH)
BSC
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
22
0.86
(.034)
REF
0.1016 ±0.0508
(.004 ±.002)
MSOP (MSE16) 0213 REV F
3509fd
For more information www.linear.com/LT3509
LT3509
Revision History
REV
DATE
DESCRIPTION
C
4/10
Changed Pin Name to RT
D
01/16
(Revision history begins at Rev C)
PAGE NUMBER
1, 2, 6, 7, 20,
21, 24
Revised Absolute Maximum Ratings
2
Updated Notes and Change/Add Values in Electrical Characteristics
3
Revised Values in Typical Performance Characteristics
5
Revised Values in Pin Functions
6
Revised Values in Startup and Shutdown Section
8
Revised Values in Shutdown and Soft-Start, Frequency Setting Sections, and Table 1
9
Clarified Sync Pin Function Description
6
Clarified External Synchronization Applications Information
10
3509fd
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LT3509
23
LT3509
Typical Applications
2MHz, 5V and 3.3V Outputs
VIN
6.5V TO 16V
(TRANSIENT TO 60V)
2.2µF
BD
BOOST2
0.1µF
0.1µF
6.8µH
VOUT
5V
0.7A
VIN
BOOST1
SW1
MBRM140
52.3k
4.7µH
SW2
MBRM140
LT3509
DA1
DA2
FB1
FB2
VOUT
3.3V
0.7A
31.6k
RUN/SS1 RUN/SS2
GND
SYNC RT
10k
10µF
16.9k
22nF
22nF
10µF
10k
fSW = 2MHz
3509 TA02
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VIN: 3.3V to 60V, VOUT = 1.25V, IQ = 100µA, ISD < 1µA, 3mm × 3mm
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LT3480
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High
VIN: 3.6V to 38V, VOUT = 0.78V, IQ = 70µA, ISD < 1µA, 3mm × 3mm
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LT3481
34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, High
VIN: 3.6V to 34V, VOUT = 1.26V, IQ = 50µA, ISD < 1µA, 3mm × 3mm
Efficiency Step-Down DC/DC Converter with Burst Mode Operation DFN-10, MSOP-10E Package
LT3493
36V, 1.4A(IOUT), 750kHz High Efficiency Step-Down DC/DC
VIN: 36V to 36V, VOUT = 0.8V, IQ = 1.9mA, ISD < 1µA, 2mm × 3mm
Converter
DFN-6 Package
LT3500
36V, 40Vmax, 2A, 2.5MHz High Efficiency DC/DC Converter
VIN: 3.6V to 36V, VOUT = 0.8V, IQ = 2.5mA, ISD < 10µA, 3mm × 3mm
and LDO Controller
DFN-10
LT3501
25V, Dual 3A (IOUT), 1.5MHz High Efficiency Step-Down
VIN: 3.3V to 25V, VOUT = 0.8V, IQ = 3.7mA, ISD = 10µA, TSSOP-20E
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Package
LT3505
36V with Transient Protection to 40V, 1.4A (IOUT), 3MHz,
VIN: 3.6V to 34V, VOUT = 0.78V, IQ = 2mA, ISD < 2µA, 3mm × 3mm
High Efficiency Step-Down DC/DC Converter
DFN-8, MSOP-8E Package
LT3506/
25V, Dual 1.6A (IOUT), 575kHz,/1.1MHz High Efficiency
VIN: 3.6V to 25V, VOUT = 0.8V, IQ = 3.8mA, ISD = 30µA, 5mm × 4mm
LT3506A
Step-Down DC/DC Converter
DFN-16 TSSOP-16E Package
LT3507
36V 2.5MHz, Triple (2.4A + 1.5A + 1.5A (IOUT)) with LDO
VIN: 4V to 36V, VOUT = 0.8V, IQ = 7mA, ISD = 1µA, 5mm × 7mm
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QFN-38
LT3508
36V with Transient Protection to 40V, Dual 1.4A (IOUT), 3MHz,
VIN: 3.7V to 37V, VOUT = 0.8V, IQ = 4.6mA, ISD = 1µA, 4mm × 4mm
High Efficiency Step-Down DC/DC Converter
QFN-24, TSSOP-16E Package
LT3510
25V, Dual 2A (IOUT), 1.5MHz High Efficiency Step-Down
VIN: 3.3V to 25V, VOUT = 0.8V, IQ = 3.7mA, ISD = 10µA, TSSOP-20E
DC/DC Converter
Package
LT3684
34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz,
VIN: 3.6V to 34V, VOUT = 1.26V, IQ = 850µA, ISD < 1µA, 3mm × 3mm
High Efficiency Step-Down DC/DC Converter
DFN-10, MSOP-10E Package
LT3685
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz,
VIN: 3.6V to 38V, VOUT = 0.78V, IQ = 70µA, ISD < 1µA, 3mm × 3mm
High Efficiency Step-Down DC/DC Converter
DFN-10, MSOP-10E Package
24 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/LT3509
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/LT3509
3509fd
LT 0116 REV D • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 2007
工商网监
湘ICP备2023018690号
