LT3799
Offline Isolated Flyback
LED Controller with Active PFC
FEATURES
DESCRIPTION
Isolated PFC LED Driver with Minimum Number of
External Components
n TRIAC Dimmable
n V and V
IN
OUT Limited Only by External Components
n Active Power Factor Correction (Typical PFC > 0.97)
n Low Harmonic Content
n No Opto-Coupler Required
n Accurate Regulated LED Current (±5% Typical)
n Open LED and Shorted LED Protection
n Thermally Enhanced 16-Lead MSOP Package
The LT®3799 is an isolated flyback controller with power
factor correction specifically designed for driving LEDs.
The controller operates using critical conduction mode
allowing the use of a small transformer. Using a novel
current sensing scheme, the controller is able to deliver a
well regulated current to the secondary side without using
an opto-coupler. A strong gate driver is included to drive
an external high voltage MOSFET. Utilizing an onboard
multiplier, the LT3799 typically achieves power factors
of 0.97. The FAULT pin provides notification of open and
short LED conditions.
n
APPLICATIONS
Offline 4W to 100W+ LED Applications
High DC VIN LED Applications
n
n
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and
True Color PWM is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Patents pending.
The LT3799 uses a micropower hysteretic start-up to
efficiently operate at offline input voltages, with a third
winding to provide power to the part. An internal LDO
provides a well regulated supply for the part’s internal
circuitry and gate driver.
TYPICAL APPLICATION
TRIAC Dimmable 20W LED Driver
LED Current vs Input Voltage
1.20
100k
0.22µF
0.1µF
499k
200Ω
1.15
20Ω
100k
4.7pF
10µF
499k
4:1:1
1.05
2k
VIN
DCM
VIN_SENSE
FB
VREF
100k
32.4k
40.2k
100k
NTC 16.2k
FAULT
CTRL3
GATE
CTRL2
SENSE
CTRL1
INTVCC
GND
0.1µF
COMP–
1.00
0.95
0.90
560µF
×2
0.85
20W
LED
POWER
20Ω
0.80
90
100
110
120
130
VIN (VAC)
140
150
3799 TA01b
0.05Ω
4.7µF
FAULT CT COMP+
1A
100k
4.99k
LT3799
6.34k
1.10
ILED (A)
90V
TO 150V
AC
2.2nF
3799 TA01a
0.1µF
3799fb
For more information www.linear.com/LT3799
1
LT3799
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
VIN, FAULT..................................................................32V
GATE, INTVCC............................................................12V
CTRL1, CTRL2, CTRL3, VIN_SENSE, COMP–.................4V
FB, CT, VREF, COMP+,....................................................3V
SENSE.......................................................................0.4V
DCM........................................................................±3mA
Maximum Junction Temperature........................... 125°C
Operating Temperature Range (Note 2)
LT3799E............................................. –40°C to 125°C
LT3799I.............................................. –40°C to 125°C
Storage Temperature Range................... –65°C to 150°C
TOP VIEW
CTRL1
CTRL2
CTRL3
VREF
FAULT
CT
COMP+
COMP–
1
2
3
4
5
6
7
8
17
GND
16
15
14
13
12
11
10
9
VIN_SENSE
SENSE
GATE
INTVCC
NC
VIN
DCM
FB
MSE PACKAGE
16-LEAD PLASTIC MSOP
θJA = 50°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
(http://www.linear.com/product/LT3799#orderinfo)
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3799EMSE#PBF
LT3799EMSE#TRPBF
3799
16-Lead Plastic MSOPE
–40°C to 125°C
LT3799IMSE#PBF
LT3799IMSE#TRPBF
3799
16-Lead Plastic MSOPE
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 18V, INTVCC = 11V, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
VIN Turn-On Voltage
22.2
23
24.2
V
VIN Turn-Off Voltage
11.8
12.3
13.0
V
VIN Hysteresis
VTURNON – VTURNOFF
10.7
V
VIN Shunt Regulator Voltage
I = 1mA
25.0
V
VIN Shunt Regulator Current Limit
15
mA
VIN Quiescent Current
Before Turn-On
After Turn-On
55
65
70
75
µA
µA
INTVCC Quiescent Current
Before Turn-On
After Turn-On
12
1.5
16
1.2
20.0
2.6
µA
mA
VIN_SENSE Threshold
Turn-Off
30
65
90
mV
1.3
V
2
1.98
2.03
2.03
V
V
VIN_SENSE Linear Range
0
VREF Voltage
0µA Load
200µA Load
Error Amplifier Voltage Gain
∆VCOMP+/∆VCOMP–, CTRL1 = 1V, CTRL2 = 2V, CTRL3 = 2V
100
V/V
Error Amplifier Transconductance
∆I = 5µA
50
µmhos
2
l
l
1.97
1.95
3799fb
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LT3799
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 18V, unless otherwise noted.
PARAMETER
CONDITIONS
FB Pin Bias Current
(Note 3), FB = 1V
MIN
CTRL1/CTRL2/CTRL3 Pin Bias Current
CTRL/CTRL2/CTRL3 = 1V
Maximum SENSE Current Limit Threshold
96
TYP
MAX
UNITS
100
600
nA
±25
nA
106
mV
100
SENSE Input Bias Current
Current Out of Pin, SENSE = 0V
15
µA
Current Loop Voltage Gain
∆VCTRL /∆VSENSE, 1000pF Cap from COMP+ to COMP–
21
V/V
CT Pin Charge Current
10
µA
CT Pin Discharge Current
200
nA
mV
CT Pin Low Threshold
Falling Threshold
240
CT Pin High Threshold
Rising Threshold
1.25
V
100
mV
CT Pin Low Hysteresis
FB Pin High Threshold
1.22
1.25
1.29
V
DCM Current Turn-On Threshold
Current Out of Pin
45
µA
Maximum Oscillator Frequency
COMP+ = 1.2V, V
300
kHz
Minimum Oscillator Frequency
COMP+ = 0V, V
IN_SENSE = 1V
IN_SENSE
Back-Up Oscillator Frequency
25
kHz
20
kHz
Linear Regulator
INTVCC Regulation Voltage
9.8
10
10.4
V
750
1150
mV
Dropout (VIN – INTVCC)
INTVCC = –10mA; Below VIN Turn-Off Voltage
Current Limit
Below Undervoltage Threshold
12
25
mA
Current Limit
Above Undervoltage Threshold
80
120
mA
Gate Driver
tr GATE Driver Output Rise Time
CL = 3300pF, 10% to 90%
20
ns
tf GATE Driver Output Fall Time
CL = 3300pF, 90% to 10%
20
ns
GATE Output Low (VOL)
0.05
GATE Output High (VOH)
V
INTVCC
– 0.05
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3799E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
V
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LT3799I is guaranteed to meet performance specifications from –40°C to
125°C operating junction temperature.
Note 3: Current flows out of the FB pin.
3799fb
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3
LT3799
TYPICAL PERFORMANCE CHARACTERISTICS
VIN Start-Up Voltage
vs Temperature
Input Voltage Hysteresis
vs Temperature
VIN IQ vs Temperature
24.0
12.0
140
100
IQ (µA)
INPUT VOLTAGE (V)
HYSTERESIS VOLTAGE (V)
120
23.5
23.0
VIN = 24V
80
VIN = 12V
60
40
22.5
11.6
11.2
10.8
10.4
20
–25
50
25
0
75
TEMPERATURE (°C)
100
0
–50
125
3799 G01
2.100
2.075
2.075
2.050
2.050
2.025
2.025
VREF (V)
VREF (V)
VREF vs Temperature
2.100
2.000
NO LOAD
1.975
200µA LOAD
1.925
1.925
–25
50
25
0
75
TEMPERATURE (°C)
100
125
1.900
100
125
3799 G03
Current Limit vs Temperature
MAX ILIM
NO LOAD
200µA LOAD
80
60
40
MIN ILIM
20
14
16
18
20
3799 G05
22 24
VIN (V)
26
28
32
30
0
–50
–25
3799 G05
50
25
0
75
TEMPERATURE (°C)
100
125
3799 G06
Minimum Oscillator Frequency
vs Temperature
375
70
350
60
325
50
FREQUENCY (kHz)
FREQUENCY (kHz)
50
25
0
75
TEMPERATURE (°C)
100
300
275
250
4
–25
3799 G02
Maximum Oscillator Frequency
vs Temperature
225
–50
10.0
–50
120
1.975
1.950
125
100
VREF vs VIN
2.000
1.950
1.900
–50
50
25
0
75
TEMPERATURE (°C)
–25
CURRENT LIMIT (mA)
22.0
–50
40
30
20
–25
50
25
0
75
TEMPERATURE (°C)
100
125
10
–50
3799 G07
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3799 G08
3799fb
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LT3799
TYPICAL PERFORMANCE CHARACTERISTICS
CT Pin Discharge Current
vs Temperature
CT Pin Charge Current
vs Temperature
200
CT DISCHARGE CURRENT (nA)
8
6
4
2
–25
50
25
0
75
TEMPERATURE (°C)
100
180
170
150
–50
125
1.2
100
10.05
9.6
10.00
3799 G12
25.00
24.75
50
25
0
75
TEMPERATURE (°C)
100
125
3799 G15
100
125
9.95
3799 G13
1.2
25
1.0
20
0.8
15
220V APPLICATION
0.4
5
0.2
50
25
0
75
TEMPERATURE (°C)
125
100
3799 G16
PAGE 17 SCHEMATIC
0.6
10
–25
10 12 14 16 18 20 22 24 26 28 30 34
VIN (V)
LED Current vs TRIAC Angle
30
0
–50
PART OFF
3799 G14
Maximum Shunt Current
vs Temperature
SHUNT CURRENT (mA)
ISHUNT = 10mA
125
3799 G11
10.10
9.8
50
0
25
75
TEMPERATURE (°C)
100
PART ON
10.15
10mA LOAD
–25
50
25
0
75
TEMPERATURE (°C)
INTVCC vs VIN
NO LOAD
9.4
–50
–25
3799 G10
10.0
125
25.50
–25
0
–50
125
10.20
25.75
24.50
–50
100
10.4
VIN Shunt Voltage vs Temperature
25.25
50
25
0
75
TEMPERATURE (°C)
10.25
1.1
50
25
0
75
TEMPERATURE (°C)
0.1
10.6
10.2
1.3
INTVCC (V)
CT PIN VOLTAGE (V)
1.4
–25
0.2
INTVCC vs Temperature
1.5
VIN SHUNT VOLTAGE (V)
–25
3799 G09
CT Pin High Threshold
vs Temperature
1.0
–50
0.3
160
INTVCC (V)
0
–50
190
ILED (A)
CT CHARGE CURRENT (µA)
10
0.4
CT PIN VOLTAGE (V)
12
26.00
CT Pin Low Threshold
vs Temperature
0
120V APPLICATION
0
30
120
90
60
150
TRIAC ANGLE (DEGREES)
180
3799 G17
3799fb
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5
LT3799
TYPICAL PERFORMANCE CHARACTERISTICS
LED Current vs Input Voltage
1.20
PAGE 17 SCHEMATIC:
OPTIMIZED FOR 120V
1.15
PAGE 17 SCHEMATIC:
OPTIMIZED FOR 220V
1.05
1.05
1.05
ILED (A)
1.10
1.00
PAGE 17 SCHEMATIC:
UNIVERSAL
1.15
1.10
1.00
1.00
0.95
0.95
0.95
0.90
0.90
0.90
0.85
0.85
0.85
0.80
170 180 190 200 210 220 230 240 250 260 270
VIN (VAC)
0.80
100
110
120
130
VIN (VAC)
140
150
3799 G18
0.99
0.99
0.98
0.98
0.97
0.97
POWER FACTOR
1.00
0.96
0.95
0.94
0.93
0.92
0.90
PAGE 17 SCHEMATIC:
OPTIMIZED FOR 120V
90
100
110
120
130
VIN (VAC)
140
150
3799 G21
95
PAGE 17 SCHEMATIC:
OPTIMIZED FOR 220V
0.94
0.93
0.97
0.96
0.95
0.94
0.93
0.92
0.92
0.91
0.91
0.90
170 180 190 200 210 220 230 240 250 260 270
VIN (VAC)
0.90
95
Efficiency vs Input Voltage
100
PAGE 17 SCHEMATIC:
95 UNIVERSAL
PAGE 17 SCHEMATIC:
OPTIMIZED FOR 220V
90
85
80
75
90
85
80
75
85
80
75
70
70
70
65
65
65
60
170 180 190 200 210 220 230 240 250 260 270
VIN (VAC)
60
60
90
100
110
120
130
VIN (VAC)
140
150
3799 G24
90 110 130 150 170 190 210 230 250 270
VIN (VAC)
3799 G23
Efficiency vs Input Voltage
PAGE 17 SCHEMATIC:
OPTIMIZED FOR 120V
PAGE 17 SCHEMATIC:
UNIVERSAL
0.99
0.98
0.95
100
EFFICIENCY (%)
EFFICIENCY (%)
1.00
3799 G22
90
6
Power Factor vs Input Voltage
0.96
Efficiency vs Input Voltage
100
3799 G20
Power Factor vs Input Voltage
Power Factor vs Input Voltage
1.00
0.91
90 110 130 150 170 190 210 230 250 270
VIN (VAC)
3799 G19
POWER FACTOR
90
EFFICIENCY (%)
0.80
POWER FACTOR
LED Current vs Input Voltage
1.20
1.10
ILED (A)
ILED (A)
1.15
LED Current vs Input Voltage
1.20
3799 G25
90 110 130 150 170 190 210 230 250 270
VIN (VAC)
3799 G26
3799fb
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LT3799
PIN FUNCTIONS
CTRL1, CTRL2, CTRL3 (Pin 1, Pin 2, Pin 3): Current Output
Adjustment Pins. These pins control the output current.
The lowest value of the three CTRL inputs is compared to
the negative input of the operational amplifier. Due to the
unique nature of the LT3799 control loop, the maximum
current does not directly correspond to the VCTRL voltages.
VREF (Pin 4): Voltage Reference Output Pin, Typically 2V.
This pin drives a resistor divider for the CTRL pin, either
for analog dimming or for temperature limit/compensation
of LED load. Can supply up to 200µA.
FAULT (Pin 5): Fault Pin. An open-collector pull-down on
FAULT asserts if FB is greater than 1.25V with the CT pin
higher than 1.25V.
CT (Pin 6): Timer Fault Pin. A capacitor is connected
between this pin and ground to provide an internal timer
for fault operations. During start-up, this pin is pulled to
ground and then charged with a 10µA current. Faults related
to the FB pin will be ignored until the CT pin reaches 1.25V.
If a fault is detected, the controller will stop switching and
begin to discharge the CT capacitor with a 200nA pull-down
current. When the pin reaches 240mV, the controller will
start to switch again.
COMP+, COMP– (Pin 7, Pin 8): Compensation Pins for
Internal Error Amplifier. Connect a capacitor between these
two pins to compensate the internal feedback loop.
FB (Pin 9): Voltage Loop Feedback Pin. FB is used to
detect open LED conditions by sampling the third winding
voltage. An open LED condition is reported if the CT pin
and the FB pin are higher than 1.25V.
DCM (Pin 10): Discontinuous Conduction Mode Detection
Pin. Connect a capacitor and resistor in series with this
pin to the third winding.
VIN (Pin 11): Input Voltage. This pin supplies current to
the internal start-up circuitry and to the INTVCC LDO. This
pin must be locally bypassed with a capacitor. A 25V shunt
regulator is internally connected to this pin.
INTVCC (Pin 13): Regulated Supply for Internal Loads
and GATE Driver. Supplied from VIN and regulates to 10V
(typical). INTVCC must be bypassed with a 4.7µF capacitor
placed close to the pin.
GATE (Pin 14): N-Channel MOSFET Gate Driver Output.
Switches between INTVCC and GND. This pin is pulled to
GND during shutdown state.
SENSE (Pin 15): The Current Sense Input for the Control
Loop. Kelvin connect this pin to the positive terminal of
the switch current sense resistor, RSENSE, and the source
of the N-channel MOSFET. The negative terminal of the
current sense resistor should be connected to the GND
plane close to the IC.
VIN_SENSE (Pin 16): Line Voltage Sense Pin. The pin is used
for sensing the AC line voltage to perform power factor
correction. Connect the output of a resistor divider from
the line voltage to this pin. The voltage on this pin should
be between 1.25V to 1.5V at the maximum input voltage.
GND (Exposed Pad Pin 17): Ground. The exposed pad
of the package provides both electrical contact to ground
and good thermal contact to the printed circuit board.
The exposed pad must be soldered to the circuit board
for proper operation.
3799fb
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7
LT3799
BLOCK DIAGRAM
D2
R4
C1
R5
FB
C4
5
CT
10
DCM
16
VIN_SENSE
ONE
SHOT
3
INTVCC
CURRENT
COMPARATOR
600mV
A7
COMP–
CTRL2
L1B
N:1
C7
VOUT –
–
A1
+
CTRL3
C5
R8
S
S
R
DRIVER
Q
MASTER
LATCH
SW1
+– A5
+
+
13
R7
1M
2
+
A8
–
1.22V
COMP+
CTRL1
VOUT +
11
VIN
FAULT
C6
1
L1A
R2
A3
–
8
C2
D1
S&H
FAULT
DETECTION
+
A2
+ –
7
T1
R1
C3
L1C
R10
9
6
•
VIN
R3
GATE
SENSE
A4
MULTIPLIER
LOW OUTPUT
CURRENT
OSCILLATOR
M1
15
R6
GND
A6
14
17
4 VREF
3799 BD
OPERATION
The LT3799 is a current mode switching controller IC
designed specifically for generating an average current
output in an isolated flyback topology. The special problem
normally encountered in such circuits is that information
relating to the output voltage and current on the isolated
secondary side of the transformer must be communicated
to the primary side in order to maintain regulation. Historically, this has been done with an opto-isolator. The LT3799
uses a novel method of using the external MOSFETs peak
current information from the sense resistor to calculate
the output current of a flyback converter without the need
of an opto-coupler. In addition, it also detects open LED
conditions by examining the third winding voltage when
the main power switch is off.
8
Power factor has become an important specification for
lighting. A power factor of one is achieved if the current
drawn is proportional to the input voltage. The LT3799
modulates the peak current limit with a scaled version of
the input voltage. This technique provides power factors
of 0.97 or greater.
The Block Diagram shows an overall view of the system.
The external components are in a flyback topology configuration. The third winding senses the output voltage
and also supplies power to the part in steady-state operation. The VIN pin supplies power to an internal LDO that
generates 10V at the INTVCC pin. The novel control circuitry
consists of an error amplifier, a multiplier, a transmission
3799fb
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LT3799
OPERATION
gate, a current comparator, a low output current oscillator
and a master latch, which will be explained in the following sections. The part also features a sample-and-hold
to detect open LED conditions, along with a FAULT pin. A
comparator is used to detect discontinuous conduction
mode (DCM) with a cap connected to the third winding.
The part features a 1.9A gate driver.
The LT3799 employs a micropower hysteretic start-up
feature to allow the part to work at any combination of
input and output voltages. In the Block Diagram, R3 is used
to stand off the high voltage supply voltage. The internal
LDO starts to supply current to the INTVCC when VIN is
above 23V. The VIN and INTVCC capacitors are charged by
the current from R3. When VIN exceeds 23V and INTVCC is
in regulation at 10V, the part will began to charge the CT
pin with 10µA. Once the CT pin reaches 340mV, switching
begins. The VIN pin has 10.7V of hysteresis to allow for
plenty of flexibility with the input and output capacitor
values. The third winding provides power to VIN when its
voltage is higher than the VIN voltage. A voltage shunt is
provided for fault protection and can sink up to 15mA of
current when VIN is over 25V.
During a typical cycle, the gate driver turns the external
MOSFET on and a current flows through the primary
winding. This current increases at a rate proportional
to the input voltage and inversely proportional to the
magnetizing inductance of the transformer. The control
loop determines the maximum current and the current
comparator turns the switch off when the current level
is reached. When the switch turns off, the energy in the
core of the transformer flows out the secondary winding
through the output diode, D1. This current decreases at a
rate proportional to the output voltage. When the current
decreases to zero, the output diode turns off and voltage
across the secondary winding starts to oscillate from the
parasitic capacitance and the magnetizing inductance of
the transformer. Since all windings have the same voltage
across them, the third winding rings too. The capacitor
connected to the DCM pin, C1, trips the comparator, A2,
which serves as a dv/dt detector, when the ringing occurs.
This timing information is used to calculate the output
current (description to follow). The dv/dt detector waits
for the ringing waveform to reach its minimum value and
then the switch turns back on. This switching behavior is
similar to zero volt switching and minimizes the amount of
energy lost when the switch is turned back on, improving
efficiency as much as 5%. Since this part operates on the
edge of continuous conduction mode and discontinuous
conduction mode, this operating mode is called critical
conduction mode (or boundary conduction mode).
Primary-Side Current Control Loop
The CTRL1/CTRL2/CTRL3 pins control the output current
of the flyback controller. To simplify the loop, assume
the VIN_SENSE pin is held at a constant voltage above
1V, eliminating the multiplier from the control loop. The
error amplifier, A5, is configured as an integrator with
the external capacitor, C6. The COMP+ node voltage is
converted to a current into the multiplier with the V/I
converter, A6. Since A7’s output is constant, the output
of the multiplier is proportional to A6 and can be ignored.
The output of the multiplier controls the peak current with
its connection to the current comparator, A1. The output
of the multiplier is also connected to the transmission
gate, SW1. The transmission gate, SW1, turns on when
the secondary current flows to the output capacitor. This
is called the flyback period (when the output diode D1 is
on). The current through the 1M resistor gets integrated
by A5. The lowest CTRL input is equal to the negative input
of A5 in steady state.
A current output regulator normally uses a sense resistor
in series with the output current and uses a feedback loop
to control the peak current of the switching converter. In
this isolated case the output current information is not
available, so instead the LT3799 calculates it using the
information available on the primary side of the transformer. The output current may be calculated by taking the
average of the output diode current. As shown in Figure
1, the diode current is a triangle waveform with a base of
the flyback time and a height of the peak secondary winding current. In a flyback topology, the secondary winding
current is N times the primary winding current, where
N is the primary to secondary winding ratio. Instead of
taking the area of the triangle, think of it as a pulse width
modulation (PWM) waveform. During the flyback time,
the average current is half the peak secondary winding
3799fb
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9
LT3799
OPERATION
SECONDARY
DIODE CURRENT
voltage is 120Hz, so the control loop unity gain frequency
needs to be set less than approximately 120Hz. Without a
large amount of energy storage on the secondary side, the
output current is affected by the supply voltage changes,
but the DC component of the output current is accurate.
IPK(sec)
TRIAC Dimming Features
SWITCH
WAVEFORM
TFLYBACK
TPERIOD
3799 F01
Figure 1. Secondary Diode Current and Switch Waveforms
current and zero during the rest of the cycle. The equation
for expressing the output current is:
IOUT = 0.5 • IPK • N • D´
where D´ is equal to the percentage of the cycle represented
by the flyback time.
The LT3799 has access to both the primary winding current, the input to the current comparator, and when the
flyback time starts and ends. Now the output current can
be calculated by averaging a PWM waveform with the
height of the current limit and the duty cycle of the flyback
time over the entire cycle. In the feedback loop previously
described, the input to the integrator is such a waveform.
The integrator adjusts the peak current until the calculated
output current equals the control voltage. If the calculated
output current is low compared to the control pin, the error
amplifier increases the voltage on the COMP+ node, thus
increasing the current comparator input.
When the VIN_SENSE voltage is connected to a resistor
divider of the supply voltage, the current limit is proportional to the supply voltage if COMP+ is held constant.
The output of the error amplifier is multiplied with the
VIN_SENSE pin voltage. If the LT3799 is configured with a
fast control loop, slower changes from the VIN_SENSE pin
will not interfere with the current limit or the output current.
The COMP+ pin will adjust to the changes of the VIN_SENSE.
The only way for the multiplier to function properly is to
set the control loop to be an order of magnitude slower
than the fundamental frequency of the VIN_SENSE signal. In
the offline case, the fundamental frequency of the supply
10
The LT3799 incorporates some special features that aid in
the design of an offline LED current source when used with
a TRIAC dimmer. TRIAC dimmers are not ideal switches
when turned off and allow milliamps of current to flow
through them. This is an issue if used with a low quiescent
part such as the LT3799. Instead of turning the main power
MOSFET off when the TRIAC is off, this power device is
kept on and sinks the current to properly load the TRIAC.
When the TRIAC turns on, the VIN_SENSE pin detects this
and enables the loop, but the current comparator is always
enabled and turns the switch off if it is tripped.
Start-Up
The LT3799 uses a hysteretic start-up to operate from
high offline voltages. A resistor connected to the supply
voltage protects the part from high voltages. This resistor is connected to the VIN pin on the part and also to a
capacitor. When the resistor charges the part up to 23V and
INTVCC is in regulation at 10V, the part begins to charge the
CT pin to 340mV and then starts to switch. The resistor
does not provide power for the part in steady state, but
relies on the capacitor to start-up the part, then the third
winding begins to provide power to the VIN pin along with
the resistor. An internal voltage clamp is attached to the
VIN pin to prevent the resistor current from allowing VIN
to go above the absolute maximum voltage of the pin.
The internal clamp is set at 25V and is capable of 28mA
(typical) of current at room temperature. But, ideally, the
resistor connected between the input supply and the VIN
pin should be chosen so that less than 10mA is being
shunted by this internal clamp.
CT Pin and Faults
The CT pin is a timing pin for the fault circuitry. When the
input voltages are at the correct levels, the CT pin sources
10µA of current. When the CT pin reaches 340mV, the part
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LT3799
OPERATION
begins to switch. The output voltage information from the
FB pin is sampled but ignored until the CT pin reaches
1.25V. When this occurs, if the FB pin is above 1.25V, the
fault flag pulls low. The FAULT pin is meant to be used
with a large pull-up resistor to the INTVCC pin or another
supply. The CT pin begins to sink 200nA of current. When
the CT pin goes below 240mV, the part will re-enable itself,
begin to switch, and start to source 10µA of current to the
CT pin but not remove the fault condition. When the CT
pin reaches 1.25V and FB is below 1.25V, the FAULT pin
will no longer pull low and switching will continue. If not
below 1.25V, the process repeats itself.
where
Programming Output Current
The VREF pin supplies a 2V reference voltage to be used
with the control pins. To set an output current, a resistor
divider is used from the 2V reference to one of the control
pins. The following equation sets the output current with
a resistor divider:
The maximum output current depends on the supply
voltage and the output voltage in a flyback topology.
With the VIN_SENSE pin connected to 1V and a DC supply
voltage, the maximum output current is determined at
the minimum supply voltage, and the maximum output
voltage using the following equation:
IOUT(MAX) = 2 • (1−D) •
N
42 • R SENSE
D=
VOUT • N
VOUT • N+ VIN
The maximum control voltage to achieve this maximum
output current is (1-D) • 47.5%.
For control voltages below the maximum, the output current is equal to the following equation:
IOUT = CTRL •
N
42 • R SENSE
⎛
⎞
2N
R1= R2 ⎜
− 1⎟
⎝ 42 • IOUT • R SENSE
⎠
where R1 is the resistor connected to the VREF pin and the
CTRL pin and R2 is the resistor connected to the CTRL
pin and ground.
where
VOUT • N
D=
VOUT • N+ VIN
The maximum control voltage to achieve this maximum
output current is 2V • (1-D).
It is suggested to operate at 95% of these values to give
margin for the part’s tolerances.
When designing for power factor correction, the output
current waveform is going to have a half sine wave squared
shape and will no longer be able to provide the above
currents. By taking the integral of a sine wave squared
over half a cycle, the average output current is found to
be half the value of the peak output current. In this case,
the recommended maximum average output current is
as follows:
N
IOUT(MAX) = 2•(1−D) •
• 47.5%
42
•
R
SENSE
When used with an AC input voltage, the LT3799 senses
when the VIN_SENSE goes below 65mV and above 65mV
for detecting when the TRIAC is off. During this low input
voltage time, the output current regulation loop is off but the
part still switches. This helps with output current regulation
with a TRIAC but introduces a line regulation error. When
VIN_SENSE is low, very little power is being delivered to the
output and since the output current regulation loop is off,
this time period needs to be accounted for in setting the
output current. This time period slightly varies with line
voltage. Figure 2 shows the correction factor in selecting
the resistor divider resistors. When used with AC, the following equation should be used with the correction factor:
IOUT = CTRL •
N
42 • R SENSE • CF
⎛
⎞
2N
R1= R2 ⎜
− 1⎟
⎝ 42 • IOUT • R SENSE • CF
⎠
3799fb
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11
LT3799
OPERATION
primary winding of the transformer. The minimum voltage of the switch node during this discontinuous ring is
VIN – VOUT • N. The LT3799 turns the switch back on at
this time, during the discontinuous switch waveform, by
sensing when the slope of the switch waveform goes from
negative to positive using the dv/dt detector. This switching
technique may increase efficiency by 5%.
OUTPUT CURRENT CORRECTION FACTOR
1.16
1.14
1.12
1.10
1.08
1.06
Sense Resistor Selection
1.04
1.02
0
0.5
1
PEAK VIN_SENSE
1.5
3799 F02
Figure 2. Output Current Correction Factor
where CF is the output current correction factor on the
Y-axis in Figure 2.
Critical Conduction Mode Operation
Critical conduction mode is a variable frequency switching
scheme that always returns the secondary current to zero
with every cycle. The LT3799 relies on boundary mode
and discontinuous mode to calculate the critical current
because the sensing scheme assumes the secondary
current returns to zero with every cycle. The DCM pin
uses a fast current input comparator in combination with
a small capacitor to detect dv/dt on the third winding. To
eliminate false tripping due to leakage inductance ringing,
a blanking time of between 600ns and 2.25µs is applied
after the switch turns off, depending on the current limit.
The detector looks for 40µA of current through the DCM
pin due to falling voltage on the third winding when the
secondary diode turns off. This detection is important
since the output current is calculated using this comparator’s output. This is not the optimal time to turn the
switch on because the switch voltage is still close to VIN
+ VOUT • N and would waste all the energy stored in the
parasitic capacitance on the switch node. Discontinuous
ringing begins when the secondary current reaches zero
and the energy in the parasitic capacitance on the switch
node transfers to the input capacitor. This is a secondorder network composed of the parasitic capacitance on
the switch node and the magnetizing inductance of the
12
The resistor, RSENSE, between the source of the external
N-channel MOSFET and GND should be selected to provide
an adequate switch current to drive the application without
exceeding the current limit threshold .
For applications without power factor correction, select a
resistor according to:
RSENSE =
2(1−D)N
• 95%
IOUT • 42
where
D=
VOUT • N
VOUT • N+ VIN
For applications with power factor correction, select a
resistor according to:
RSENSE =
2(1−D)N
• 47.5%
IOUT • 42
where
D=
VOUT • N
VOUT • N+ VIN
Minimum Current Limit
The LT3799 features a minimum current limit of approximately 7% of the peak current limit. This is necessary when
operating in critical conduction mode since low current
limits would increase the operating frequency to a very
high frequency. The output voltage sensing circuitry needs
a minimum amount of flyback waveform time to sense the
output voltage on the third winding. The time needed is
350ns. The minimum current limit allows the use of smaller
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LT3799
OPERATION
transformers since the magnetizing primary inductance
does not need to be as high to allow proper time to sample
the output voltage information.
Errors Affecting Current Output Regulation
There are a few factors affecting the regulation of current in
a manufacturing environment along with some systematic
issues. The main manufacturing issues are the winding
turns ratio and the LT3799 control loop accuracy. The
winding turns ratio is well controlled by the transformer
manufacturer’s winding equipment, but most transformers
do not require a tight tolerance on the winding ratio. We
have worked with transformer manufacturers to specify
±1% error for the turns ratio. Just like any other LED driver,
the part is tested and trimmed to eliminate offsets in the
control loop and an error of ±3% is specified at 80% of
the maximum output current. The error grows larger as
the LED current is decreased from the maximum output
current. At half the maximum output current, the error
doubles to ±6%.
There are a number of systematic offsets that may be eliminated by adjusting the control voltage from the ideal voltage.
It is difficult to measure the flyback time with complete
accuracy. If this time is not accurate, the control voltage
needs to be adjusted from the ideal value to eliminate the
offset but this error still causes line regulation errors. If
the supply voltage is lowered, the time error becomes a
smaller portion of the switching cycle period so the offset
becomes smaller and vice versa. This error may be compensated for at the primary supply voltage, but this does
not solve the problem completely for other supply voltages.
Another systematic error is that the current comparator
cannot instantaneously turn off the main power device.
This delay time leads to primary current overshoot. This
overshoot is less of a problem when the output current is
close to its maximum, since the overshoot is only related
to the slope of the primary current and not the current
level. The overshoot is proportional to the supply voltage,
so again this affects the line regulation.
Universal Input
The LT3799 operates over the universal input range of
90VAC to 265VAC . Output current regulation error may
be minimized by using two application circuits for the
wide input range: one optimized for 120VAC and another
optimized for 220VAC . The first application pictured in
the Typical Applications section shows three options:
universal input, 120VAC , and 220VAC . The circuit varies by
three resistors. In the Typical Performance Characteristics
section, the LED Current vs VIN graphs show the output
current line regulation for all three circuits.
Selecting Winding Turns Ratio
Boundary mode operation gives a lot of freedom in selecting
the turns ratio of the transformer. We suggest to keep the
duty cycle low, lower NPS, at the maximum input voltage
since the duty cycle will increase when the AC waveform is
decreases to zero volts. A higher NPS increases the output
current while keeping the primary current limit constant.
Although this seems to be a good idea, it comes at the
expense of a higher RMS current for the secondary-side
diode which might not be desirable because of the primary
side MOSFET’s superior performance as a switch. A higher
NPS does reduce the voltage stress on the secondary-side
diode while increasing the voltage stress on the primaryside MOSFET. If switching frequency at full output load is
kept constant, the amount of energy delivered per cycle by
the transformer also stays constant regardless of the NPS.
Therefore, the size of the transformer remains the same at
practical NPS’s. Adjusting the turns ratio is a good way to
find an optimal MOSFET and diode for a given application.
Switch Voltage Clamp Requirement
Leakage inductance of an offline transformer is high due
to the extra isolation requirement. The leakage inductance
energy is not coupled to the secondary and goes into
the drain node of the MOSFET. This is problematic since
400V and higher rated MOSFETs cannot always handle
this energy by avalanching. Therefore the MOSFET needs
protection. A transient voltage suppressor (TVS) and
diode are recommended for all offline application and
connected, as shown in Figure 3. The TVS device needs
a reverse breakdown voltage greater than (VOUT + Vf)*N
where VOUT is the output voltage of the flyback converter,
Vf is the secondary diode forward voltage, and N is the
turns ratio.
3799fb
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13
LT3799
OPERATION
the LT3799. Table 1 shows the details of several of these
transformers.
VSUPPLY
Loop Compensation
GATE
3799 F03
Figure 3. Clamp
Transformer Design Considerations
Transformer specification and design is a critical part of
successfully applying the LT3799. In addition to the usual
list of caveats dealing with high frequency isolated power
supply transformer design, the following information
should be carefully considered. Since the current on the
secondary side of the transformer is inferred by the current
sampled on the primary, the transformer turns ratio must
be tightly controlled to ensure a consistent output current.
A tolerance of ±5% in turns ratio from transformer to
transformer could result in a variation of more than ±5% in
output regulation. Fortunately, most magnetic component
manufacturers are capable of guaranteeing a turns ratio
tolerance of 1% or better. Linear Technology has worked
with several leading magnetic component manufacturers
to produce predesigned flyback transformers for use with
The current output feedback loop is an integrator configuration with the compensation capacitor between the
negative input and output of the operational amplifier.
This is a one-pole system therefore a zero is not needed
in the compensation. For offline applications with PFC,
the crossover should be set an order of magnitude lower
than the line frequency of 120Hz or 100Hz. In a typical
application, the compensation capacitor is 0.1µF.
In non-PFC applications, the crossover frequency may be
increased to improve transient performance. The desired
crossover frequency needs to be set an order of magnitude
below the switching frequency for optimal performance.
MOSFET and Diode Selection
With a strong 1.9A gate driver, the LT3799 can effectively
drive most high voltage MOSFETs. A low Qg MOSFET is
recommended to maximize efficiency. In most applications,
the RDS(ON) should be chosen to limit the temperature rise
of the MOSFET. The drain of the MOSFET is stressed to
VOUT • NPS + VIN during the time the MOSFET is off and
the secondary diode is conducting current. But in most
applications, the leakage inductance voltage spike exceeds
this voltage. The voltage of this stress is determined
Table 1. Predesigned Transformers—Typical Specifications, Unless Otherwise Noted
TRANSFORMER
PART NUMBER
SIZE
(L × W × H)
JA4429
7508110210
LPRI
(µH)
NPSA
(NP:NS:NA)
21.1mm × 21.1mm × 17.3mm
400
15.75mm × 15mm × 18.5mm
2000
750813002
15.75mm × 15mm × 18.5mm
750811330
TARGET
APPLICATION
(VOUT /IOUT)
RPRI
(mΩ)
RSEC
(mΩ)
1:0.24:0.24
252
126
Coilcraft
22V/1A
6.67:1:1.67
5100
165
Würth Elektronik
10V/0.4A
2000
20:1.0:5.0
6100
25
Würth Elektronik
3.8V/1.1A
43.2mm × 39.6mm × 30.5mm
300
6:1.0:1.0
150
25
Würth Elektronik
18V/5A
750813144
16.5mm × 18mm × 18mm
600
4:1:0.71
2400
420
Würth Elektronik
28V/0.5A
750813134
16.5mm × 18mm × 18mm
600
8:1:1.28
1850
105
Würth Elektronik
14V/1A
750811291
31mm × 31mm × 25mm
400
1:1:0.24
550
1230
Würth Elektronik
85V/0.4A
750813390
43.18mm × 39.6mm ×
30.48mm
100
1:1:0.22
150
688
Würth Elektronik
90V/1A
750811290
31mm × 31mm × 25mm
460
1:1:0.17
600
560
Würth Elektronik
125V/0.32A
X-11181-002
23.5mm × 21.4mm × 9.5mm
500
72:16:10
1000
80
Premo
30V/0.5A
14
MANUFACTURER
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LT3799
OPERATION
by the switch voltage clamp. Always check the switch
waveform with an oscilloscope to make sure the leakage
inductance voltage spike is below the breakdown voltage
of the MOSFET. A transient voltage suppressor and diode
are slower than the leakage inductance voltage spike,
therefore causing a higher voltage than calculated.
The secondary diode stress may be as much as
VOUT + 2 • VIN /NPS due to the anode of the diode ringing
with the secondary leakage inductance. An RC snubber
in parallel with the diode eliminates this ringing, so that
the reverse voltage stress is limited to VOUT + VIN /NPS.
With a high NPS and output current greater than 3A, the
IRMS through the diode can become very high and a low
forward drop Schottky is recommended.
Discontinuous Mode Detection
The discontinuous mode detector uses AC-coupling to
detect the ringing on the third winding. A 10pF capacitor
with a 500Ω resistor in series is recommended in most
designs. Depending on the amount of leakage inductance
ringing, an additional current may be needed to prevent
false tripping from the leakage inductance ringing. A resistor from INTVCC to the DCM pin adds this current. Up to
an additional 100µA of current may be needed in some
cases. The DCM pin is roughly 0.7V, therefore the resistor
value is selected using the following equation:
R=
10V − 0.7V
I
where I is equal to the additional current into the DCM pin.
switch proportional to the line voltage by using an internal
multiplier. A power factor of >0.97 is easily attainable for
most applications by following the design equations in this
datasheet. With proper design, LT3799 applications meet
IEC 6100-3-2 Class C harmonic standards.
Protection from Open LED and Shorted LED Faults
The LT3799 detects output overvoltage conditions by looking at the voltage on the third winding. The third winding
voltage is proportional to the output voltage when the main
power switch is off and the secondary diode is conducting
current. Sensing the output voltage requires delivering
power to the output. Using the CT pin, the part turns off
switching when a overvoltage condition occurs and rechecks to see if the overvoltage condition has cleared, as
described in “CT Pin and Faults” in the Operation section.
This greatly reduces the output current delivered to the
output but a Zener is required to dissipate 2% of the set
output current during an open LED condition. The Zener
diode’s voltage needs to be 10% higher than the output
voltage set by the resistor divider connected to the FB pin.
Multiple Zener diodes in series may be needed for higher
output power applications to keep the Zener’s temperature
within the specification.
During a shorted LED condition, the LT3799 operates at
the minimum operating frequency. In normal operation,
the third winding provides power to the IC, but the third
winding voltage is zero during a shorted LED condition.
This causes the part’s VIN UVLO to shutdown switching.
The part starts switching again when VIN has reached its
turn-on voltage.
Power Factor Correction/Harmonic Content
The LT3799 attains high power factor and low harmonic
content by making the peak current of the main power
3799fb
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15
LT3799
TYPICAL APPLICATIONS
Universal TRIAC Dimmable 20W LED Driver
L2
800µH
L1
33mH
90V
TO 265V
AC
BR1
C3
0.22µF
C1
0.068µF
C2
0.1µF
R1
200Ω
R3
499k
R4
499k
R7
100k
R6
D2 20Ω
R8
100k
C4
C5 4.7pF
10µF
D3
R13
2k
VIN
DCM
VIN_SENSE
FB
R5
3.48k
R18
100k
4:1:1
R4
100k
Z1
R15
4.99k
LT3799
R16
20Ω
VREF
R16
32.4k
R9
40.2k
100k
NTC
CTRL3
GATE
CTRL2
SENSE
CTRL1
INTVCC
R10
15.4k
C9
4.7µF
GND
FAULT CT COMP+
FAULT
1A
D4
C10
560µF
×2
D1
M1
4.02k
20W
LED
POWER
Z2
RS
0.05Ω C8
2.2nF
COMP–
3799 TA02
C7, 0.1µF
BR1: DIODES, INC. HD06
D1:
CENTRAL SEMICONDUCTOR CMR1U-06M
D2, D3: DIODES INC. BAV20W
DR: CENTRAL SEMICONDUCTOR CMR1U-02M
Z1:
FAIRCHILD SMBJ170A
Z2:
CENTRAL SEMICONDUCTOR CMZ5937B
T1:
COILCRAFT JA4429-AL
M1: FAIRCHILD FDPF15N65
Component Values for Input Voltage Ranges
R5 (Ω)
R10 (Ω)
RS (Ω)
R1 (Ω)
C2 (µF)
C3 (µF)
Optimized for 110V
6.34k
16.2k
0.05
200
0.1
0.22
Optimized for 220V
3.48k
24.9k
0.075
1.00k
0.033
0.1
Universal
3.48k
15.4k
0.05
200
0.1
0.22
16
3799fb
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LT3799
TYPICAL APPLICATIONS
Universal Input TRIAC Dimmable 4W LED Driver
L1
3.3mH
R20, 10k
90V
TO 265V
AC
C1
33nF
BR1
L1
3.3mH
R21, 10k
L2, 3.3mH
C3
68nF
C2
22nF
R1
750Ω
R3
499k
R4
499k
R7
100k
R6
D2 20Ω
R8
100k
C4
C5 4.7pF
10µF
D3
R13
10k
VIN
DCM
VIN_SENSE
FB
R5
3.48k
R18
100k
20:5:1
LT3799
CTRL3
GATE
CTRL2
SENSE
CTRL1
INTVCC
R10
32.4k
FAULT
BR1: DIODES, INC. HD06
D1:
CENTRAL SEMICONDUCTOR CMR1U-06M
D2, D3: CENTRAL SEMICONDUCTOR CMMSHI-100
D4:
CENTRAL SEMICONDUCTOR CMSH2-40L
Z1:
FAIRCHILD SMBJ170A
Z2:
CENTRAL SEMICONDUCTOR CMZ59198
T1:
WÜRTH ELEKTRONIK WE-750813002
M1: FAIRCHILD FQU5N60
GND
FAULT CT COMP+
C6
0.1µF
COMP–
1A
D4
Z1
R15
4.99k
VREF
R9
40.2k
R4
100k
R16
20Ω
C9
4.7µF
C10
1500µF
D1
M1
RS
0.3Ω
4W
LED
POWER
Z2
C8
2.2nF
3799 TA03
C7, 0.1µF
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17
LT3799
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev D)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ± 0.102
(.112 ± .004)
5.23
(.206)
MIN
2.845 ± 0.102
(.112 ± .004)
0.889 ± 0.127
(.035 ± .005)
8
1
1.651 ± 0.102
(.065 ± .004)
1.651 ± 0.102 3.20 – 3.45
(.065 ± .004) (.126 – .136)
0.305 ± 0.038
(.0120 ± .0015)
TYP
16
0.50
(.0197)
BSC
4.039 ± 0.102
(.159 ± .004)
(NOTE 3)
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
9
NO MEASUREMENT PURPOSE
0.280 ± 0.076
(.011 ± .003)
REF
16151413121110 9
DETAIL “A”
0° – 6° TYP
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
GAUGE PLANE
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
1234567 8
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
18
0.86
(.034)
REF
0.1016 ± 0.0508
(.004 ± .002)
MSOP (MSE16) 0711 REV D
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For more information www.linear.com/LT3799
LT3799
REVISION HISTORY
REV
DATE
DESCRIPTION
A
01/12
Added “Maximum” to SENSE Current Limit Threshold parameter; Updated Conditions and Typ and Max values for
Dropout (VIN – INTVCC) parameter; Changed Current Limit Typ value to 15.
3
Corrected Y-axis lables for G21, G22 and G23.
6
B
02/16
PAGE NUMBER
Reordered Pin Functions in numerical order.
7
Removed section title”Setting Control Voltages for LED Over Temperature and Brownout Conditions”.
12
Amended Current Limit Below Undervoltage Threshold
3
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representaFor more information
tion that the interconnection
of its circuits aswww.linear.com/LT3799
described herein will not infringe on existing patent rights.
19
LT3799
TYPICAL APPLICATION
Universal Input TRIAC Dimmable 14W LED Driver
L2
750µH
L1
39mH
90V
TO 265V
AC
BR1
C1
47nF
C2
0.1µF
C3
0.22µF
R1
250Ω
R3
499k
R2
250Ω
R4
499k
R7
100k
R6
D2 20Ω
R8
100k
C4
C5 4.7pF
10µF
D3
4:1:0.71
R13
2k
VIN
DCM
VIN_SENSE
FB
R5
3.48k
LT3799
R16
10k
R9
40.2k
PHOTOCELL
R17
10k
CTRL3
GATE
CTRL2
SENSE
CTRL1
INTVCC
R10
23.2k
GND
FAULT CT COMP
FAULT
C6
0.1µF
BR1: DIODES, INC. HD06
D1:
CENTRAL SEMICONDUCTOR CMR1U-06M
D2, D3: DIODES INC. BAV20W
D4:
DIODES INC. DFLS1150
Z1:
FAIRCHILD SMBJ170A
Z2:
CENTRAL SEMICONDUCTOR CMZ5938B
T1:
WÜRTH ELEKTRONIK WE750813144
M1: ST MICRO STD12N65M5
+
COMP–
0.5A
D4
Z1
R15
5.90k
VREF
R18
100k
R4
100k
C10
390µF
×2
D1
R16
20Ω
C9
4.7µF
14W
LED
POWER
Z2
RS
0.10Ω C8
2.2nF
3799 TA04
C7, 0.1µF
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT3755/ LT3755-1/
LT3755-2
High Side 60V, 1MHz LED Controller with 3000:1
True Color PWM™ Dimming
VIN : 4.5V to 40V, VOUT(MAX) = 60V, Dimming: 3000:1 True Color PWM,
ISD < 1µA, 3mm × 3mm QFN-16 and MSOP-16E Packages
LT3756/ LT3756-1/
LT3756-2
High Side 100V, 1MHz LED Controller with 3000:1
True Color PWM Dimming
VIN : 6V to 100V, VOUT(MAX) = 100V, Dimming: 3000:1 True Color PWM,
ISD < 1µA, 3mm × 3mm QFN-16 and MSOP-16E Packages
LT3743
Synchronous Step-Down 20A LED Driver with
Three-State LED Current Control
VIN : 5.5V to 36V, Dimming: 10000:1 True Color PWM, ISD < 1µA,
5mm × 8mm QFN-52 Package
LT3518
2.3A, 2.5MHz High Current LED Driver with 3000:1
Dimming
VIN : 3V to 30V, Dimming: 3000:1 True Color PWM, ISD < 1µA,
4mm × 4mm QFN-16 Package
LT3517
1.3A, 2.5MHz High Current LED Driver with 3000:1
Dimming
VIN : 3V to 30V, Dimming: 3000:1 True Color PWM, ISD < 1µA,
4mm × 4mm QFN-16 Package
LT3741
High Power, Constant-Current, Constant-Voltage
Synchronous Step-Down Controller
VIN : 6V to 36V, Average Current Mode Control, ISD < 1µA,
4mm × 4mm QFN-20 and TSSOP-20E Packages
20 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/LT3799
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/LT3799
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LT 0216 REV B • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 2011
工商网监
湘ICP备2023018690号
