LT3971/LT3971-3.3/LT3971-5
38V, 1.2A, 2MHz
Step-Down Regulator with
2.8µA Quiescent Current
FEATURES
DESCRIPTION
Ultralow Quiescent Current:
2.8µA IQ Regulating 12VIN to 3.3VOUT
n Fixed Output Voltages: 3.3V, 5V,
2.1µA IQ Regulating 12VIN to 3.3VOUT
n Low Ripple Burst Mode® Operation:
Output Ripple < 15mVP-P
n Wide Input Voltage Range: 4.3V to 38V
n 1.2A Maximum Output Current
n Adjustable Switching Frequency: 200kHz to 2MHz
n Synchronizable Between 250kHz to 2MHz
n Fast Transient Response
n Accurate 1V Enable Pin Threshold
n Low Shutdown Current: I = 700nA
Q
n Power Good Flag
n Soft-Start Capability
n Internal Compensation
n Output Voltage: 1.19V to 30V
n Small Thermally Enhanced 10-Lead MSOP, 16-Lead
MSOP and (3mm × 3mm) DFN Packages
The LT®3971 is an adjustable frequency monolithic buck
switching regulator that accepts a wide input voltage range
up to 38V. Low quiescent current design consumes only
2.8µA of supply current while regulating with no load. Low
ripple Burst Mode operation maintains high efficiency at
low output currents while keeping the output ripple below
15mV in a typical application. An internally compensated
current mode topology is used for fast transient response
and good loop stability. A high efficiency 0.33Ω switch
is included on the die along with a boost Schottky diode
and the necessary oscillator, control and logic circuitry.
An accurate 1V threshold enable pin can be used to shut
down the LT3971, reducing the input supply current to
700nA. A capacitor on the SS pin provides a controlled
inrush current (soft-start). A power good flag signals
when VOUT reaches 91% of the programmed output voltage. The LT3971 is available in small 10-lead MSOP and
3mm × 3mm DFN packages with exposed pads for low
thermal resistance. A 16-lead MSOP is also offered which
has enhanced pin-to-pin fault tolerance.
n
APPLICATIONS
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners.
Automotive Battery Regulation
Power for Portable Products
n Industrial Supplies
n
n
TYPICAL APPLICATION
No Load Supply Current
3.3V Step Down Converter
3.0
OFF ON
EN
VIN
BOOST
0.47µF
PG
SS
4.7µF
4.7µH
SW
LT3971-3.3
RT
BD
49.9k
f = 800kHz
SYNC
GND
VOUT
22µF
3971 TA01
VOUT
3.3V
1.2A
INPUT CURRENT (µA)
VIN
4.5V TO 38V
OUTPUT IN REGULATION
2.5
LT3971-5
2.0
LT3971-3.3
1.5
1.0
5
10
15
20
25
30
INPUT VOLTAGE (V)
35
3971 TA01b
3971fd
1
LT3971/LT3971-3.3/LT3971-5
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VIN, EN Voltage..........................................................38V
BOOST Pin Voltage....................................................55V
BOOST Pin Above SW Pin..........................................30V
FB, VOUT, RT, SYNC, SS Voltage..................................6V
PG, BD Voltage..........................................................30V
Boost Diode Current.....................................................1A
Operating Junction Temperature Range (Note 2)
LT3971E.............................................. –40°C to 125°C
LT3971I............................................... –40°C to 125°C
Storage Temperature Range............... –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
(MSE Only)........................................................ 300°C
PIN CONFIGURATION
LT3971
LT3971
TOP VIEW
BD
1
BOOST
2
SW
3
VIN
4
EN
5
TOP VIEW
TOP VIEW
10 SYNC
BD
BOOST
SW
VIN
EN
9 PG
11
GND
LT3971
8 RT
7 SS
6 FB
DD PACKAGE
10-LEAD (3mm × 3mm) PLASTIC DFN
θJA = 45°C, θJC = 10°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
1
2
3
4
5
11
GND
10
9
8
7
6
SYNC
PG
RT
SS
FB
MSE PACKAGE
10-LEAD PLASTIC MSOP
θJA = 45°C, θJC = 10°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
LT3971-3.3, LT3971-5
BD
NC
BOOST
NC
SW
NC
VIN
EN
1
2
3
4
5
6
7
8
17
GND
16
15
14
13
12
11
10
9
GND
SYNC
PG
RT
SS
NC
FB
FB
MSE PACKAGE
16-LEAD PLASTIC MSOP
θJA = 40°C
EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
LT3971-3.3, LT3971-5
TOP VIEW
BD
1
BOOST
2
SW
3
VIN
4
EN
5
TOP VIEW
10 SYNC
11
GND
BD
BOOST
SW
VIN
EN
9 PG
8 RT
7 SS
6 VOUT
1
2
3
4
5
11
GND
10
9
8
7
6
SYNC
PG
RT
SS
VOUT
MSE PACKAGE
10-LEAD PLASTIC MSOP
DD PACKAGE
10-LEAD (3mm × 3mm) PLASTIC DFN
θJA = 45°C, θJC = 10°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
θJA = 45°C, θJC = 10°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3971EDD#PBF
LT3971EDD#TRPBF
LFJF
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3971IDD#PBF
LT3971IDD#TRPBF
LFJF
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3971EMSE#PBF
LT3971EMSE#TRPBF
LTFJG
10-Lead Plastic MSOP
–40°C to 125°C
LT3971IMSE#PBF
LT3971IMSE#TRPBF
LTFJG
10-Lead Plastic MSOP
–40°C to 125°C
LT3971EMSE16#PBF
LT3971EMSE16#TRPBF
3971
16-Lead Plastic MSOP
–40°C to 125°C
LT3971IMSE16#PBF
LT3971IMSE16#TRPBF
3971
16-Lead Plastic MSOP
–40°C to 125°C
LT3971EDD-3.3#PBF
LT3971EDD-3.3#TRPBF
LFRM
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3971IDD-3.3#PBF
LT3971IDD-3.3#TRPBF
LFRM
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3971EMSE-3.3#PBF
LT3971EMSE-3.3#TRPBF
LTFRN
10-Lead Plastic MSOP
–40°C to 125°C
LT3971IMSE-3.3#PBF
LT3971IMSE-3.3#TRPBF
LTFRN
10-Lead Plastic MSOP
–40°C to 125°C
3971fd
2
LT3971/LT3971-3.3/LT3971-5
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3971EDD-5#PBF
LT3971EDD-5#TRPBF
LFRP
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3971IDD-5#PBF
LT3971IDD-5#TRPBF
LFRP
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3971EMSE-5#PBF
LT3971EMSE-5#TRPBF
LTFRQ
10-Lead Plastic MSOP
–40°C to 125°C
LT3971IMSE-5#PBF
LT3971IMSE-5#TRPBF
LTFRQ
10-Lead Plastic MSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VEN = 12V, VBD = 3.3V unless otherwise noted. (Note 2)
PARAMETER
Minimum Input Voltage
Quiescent Current from VIN
LT3971 FB Pin Current
Internal Feedback Resistor Divider (LT3971-X)
Feedback Voltage
CONDITIONS
(Note 4)
VEN Low
VEN High, VSYNC Low
VEN High, VSYNC Low
VFB = 1.19V
MIN
l
l
l
l
LT3971-3.3 Output Voltage
l
LT3971-5 Output Voltage
l
FB Voltage Line Regulation
Switching Frequency
Minimum Switch On Time
Minimum Switch Off Time
Switch Current Limit
Switch VCESAT
Switch Leakage Current
Boost Schottky Forward Voltage
Boost Schottky Reverse Leakage
Minimum Boost Voltage (Note 3)
BOOST Pin Current
EN Voltage Threshold
EN Voltage Hysteresis
EN Pin Current
LT3971 PG Threshold Offset from VFB
LT3971 PG Hysteresis
LT3971-X PG Threshold Offset from VOUT
LT3971-X PG Hysteresis
4.3V < VIN < 38V (Note 4)
RT = 11k
RT = 35.7k
RT = 255k
1.175
1.165
3.25
3.224
4.93
4.89
1.6
0.8
160
1.8
ISW = 1A
ISH = 100mA
VREVERSE = 12V
VIN = 5V
ISW = 1A, VBOOST = 15V
EN Rising
TYP
4
0.7
1.7
l
l
0.95
VFB Rising
60
VOUT Rising
5.5
0.1
10
1.19
1.19
3.3
3.3
5
5
0.0002
2
1
200
80
110
2.4
330
0.02
770
0.02
1.4
20
1.01
30
0.2
100
20
9
1.3
MAX
4.3
1.2
2.7
4.5
12
1.205
1.215
3.35
3.376
5.07
5.11
0.01
2.4
1.2
240
150
3
1
1
1.8
28
1.07
20
140
12.5
UNITS
V
μA
μA
μA
nA
MΩ
V
V
V
V
V
V
%/V
MHz
MHz
kHz
ns
ns
A
mV
μA
mV
μA
V
mA
V
mV
nA
mV
mV
%
%
3971fd
3
LT3971/LT3971-3.3/LT3971-5
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VEN = 12V, VBD = 3.3V unless otherwise noted. (Note 2)
PG Leakage
PG Sink Current
SYNC Threshold
SYNC Pin Current
SS Source Current
VPG = 3V
VPG = 0.4V
VSS = 1V
0.6
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency, VOUT = 5V
Efficiency, VOUT = 3.3V
VIN = 12V
90
VIN = 36V
VIN = 24V
60
50
40
20
VIN = 12V
70
VIN = 36V
60
50
0
0.2
0.4
0.6
0.8
LOAD CURRENT (A)
1
20
1.2
Efficiency, VOUT = 3.3V
50
VIN = 24V
VIN = 36V
50
40
30
10
0
0.2
0.4
0.6
0.8
LOAD CURRENT (A)
1
0
0.01
1.2
0.1
1
10
100
LOAD CURRENT (mA)
1000
3971 G03
No Load Supply Current
4.0
DIODES, INC.
DFLS2100
VIN = 36V
40
30
INPUT CURRENT (µA)
VIN = 24V
60
3.5
INPUT CURRENT (µA)
60
VIN = 12V
70
No Load Supply Current
100
VIN = 12V
70
Efficiency, VOUT = 5V
3971 G02
90
80
1.6
20
30
3971 G01
EFFICIENCY (%)
VIN = 24V
40
VOUT = 5V
R1 = 1M
R2 = 309k
1.0
VOUT = 5V
90 R1 = 1M
R2 = 309k
80
EFFICIENCY (%)
70
30
100
80
EFFICIENCY (%)
EFFICIENCY (%)
80
µA
μA
V
nA
μA
TA = 25°C, unless otherwise noted.
100
90
1
temperature range. High junction temperatures degrade operating
lifetimes. Operating lifetime is derated at junction temperatures greater
than 125°C.
Note 3: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the switch.
Note 4: This is the minimum input voltage for operation with accurate FB
regulation. Minimum input voltage for output regulation depends on the
application circuit.
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3971E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization, and correlation with statistical process controls. The
LT3971I is guaranteed over the full –40°C to 125°C operating junction
100
0.02
570
0.8
0.1
1
300
0.6
l
10
20
LT3971
VOUT = 3.3V
3.0
2.5
LT3971-5
2.0
LT3971-3.3
1.5
10
0
0.01
0.1
1
10
100
LOAD CURRENT (mA)
1000
3971 G04
1
–55
–25
5
35
65
95
TEMPERATURE (°C)
125
155
3971 G05
1.0
5
10
15
20
25
30
INPUT VOLTAGE (V)
35
3971 G06
3971fd
4
LT3971/LT3971-3.3/LT3971-5
TYPICAL PERFORMANCE CHARACTERISTICS
LT3971-3.3 Output Voltage
LT3971 Feedback Voltage
LT3971-5 Output Voltage
3.345
5.06
1.200
3.330
5.04
1.195
1.190
1.185
1.180
OUTPUT VOLTAGE (V)
1.205
OUTPUT VOLTAGE (V)
FEEDBACK VOLTAGE (V)
TA = 25°C, unless otherwise noted.
3.315
3.300
3.285
3.270
1.175
–55
–25
5
35
65
95
TEMPERATURE (°C)
125
155
3.255
–55
–25
5
35
65
95
TEMPERATURE (°C)
125
LOAD CURRENT (A)
MINIMUM
1.0
0.20
TYPICAL
1.5
MINIMUM
1.0
10
15
20
25
30
INPUT VOLTAGE (V)
35
0
40
0.15
0.10
0.05
0
–0.05
–0.10
–0.15
–0.20
5
25
30
15
20
INPUT VOLTAGE (V)
10
35
3971 G08
Switching Frequency
REFERENCED FROM VOUT AT 0.5A LOAD
0
200
400
600
800 1000
LOAD CURRENT (mA)
1200
3971 G10
Switch Current Limit
Switch Current Limit
2.5
2.4
850
800
750
700
650
2.5
SWITCH CURRENT LIMIT (A)
SWITCH CURRENT LIMIT (A)
900
FREQUENCY (kHz)
–0.30
3.0
950
600
–55
40
3971 G09
1000
155
0.25
–0.25
5
125
Load Regulation
0.5
0.5
0
5
35
65
95
TEMPERATURE (°C)
0.30
VOUT = 5V
2.0
TYPICAL
–25
3971 G29
LOAD REGULATION (%)
VOUT = 3.3V
2.5
LOAD CURRENT (A)
4.94
–55
155
Maximum Load Current
2.5
1.5
4.98
3971 G28
Maximum Load Current
2.0
5.00
4.96
3971 G07
3.0
5.02
2.0
1.5
1.0
0.5
2.3
2.2
2.1
2.0
1.9
1.8
1.7
1.6
–25
5
35
65
95
TEMPERATURE (°C)
125
155
3971 G11
0
0
20
40
60
DUTY CYCLE (%)
80
100
3971 G12
DUTY CYCLE = 30%
1.5
–55 –25
5
35
65
95
TEMPERATURE (°C)
125
155
3971 G13
3971fd
5
LT3971/LT3971-3.3/LT3971-5
TYPICAL PERFORMANCE CHARACTERISTICS
Boost Pin Current
500
25
BOOST PIN CURRENT (mA)
30
300
200
100
800
20
15
10
5
0
250
500
750 1000 1250
SWITCH CURRENT (mA)
0
1500
0
250
500
750 1000 1250
SWITCH CURRENT (mA)
3971 G14
400
800
350
SWITCH ON/OFF TIME (ns)
SWITCHING FREQUENCY (kHz)
900
500
400
300
200
0
20
60
40
80
VOUT (% OF REGULATION VOLTAGE)
MIN TOFF 1A LOAD
250
200
MIN TOFF 0.5A LOAD
150
100
MIN TON
0
–55
100
–25
35
95
5
65
TEMPERATURE (°C)
125
4.0
3.8
TO RUN
3.6
3.4
3.0
6.0
0
200
600
400
800 1000
LOAD CURRENT (mA)
1200
3971 G19
1.2
1.5
1.0
0.5
0
0.25 0.5 0.75 1 1.25 1.5 1.75
SS PIN VOLTAGE (V)
2
EN Threshold
1.04
TO START
5.8
5.6
5.4
5.0
1
1.05
VOUT = 5V
TO RUN
5.2
3.2
0.6
0.4
0.8
FB PIN VOLTAGE (V)
3971 G18
THRESHOLD VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
4.6
TO START
0.2
2.0
0
155
6.2
4.4
0
3971 G16
Minimum Input Voltage
6.4
VOUT = 3.3V
4.2
200
3971 G17
Minimum Input Voltage
4.8
300
Soft-Start
3971 G30
5.0
400
2.5
300
50
100
0
500
0
1500
Minimum Switch On-Time/
Switch Off-Time
600
600
3971 G15
LT3971-X Frequency Foldback
700
700
100
SWITCH CURRENT LIMIT (A)
VCESAT (mV)
400
Frequency Foldback
900
SWITCHING FREQUENCY (kHz)
Switch VCESAT
600
0
TA = 25°C, unless otherwise noted.
1.03
1.02
1.01
RISING THRESHOLD
1.00
0.99
FALLING THRESHOLD
0.98
0.97
0.96
0
200
400
800 1000
600
LOAD CURRENT (mA)
1200
3971 G20
0.95
–55
–25
5
35
65
95
TEMPERATURE (°C)
125
155
3971 G21
3971fd
6
LT3971/LT3971-3.3/LT3971-5
TYPICAL PERFORMANCE CHARACTERISTICS
95
1.4
94
1.2
1.0
0.8
0.6
0.4
0.2
0
Transient Load Response,
Load Current Stepped from 25mA
(Burst Mode Operation) to 525mA
Power Good Threshold
1.6
THRESHOLD VOLTAGE (%)
BOOST DIODE VF (V)
Boost Diode Forward Voltage
TA = 25°C, unless otherwise noted.
93
VOUT
100mV/DIV
92
91
90
89
88
87
IL
500mA/DIV
86
0
250
500
750 1000 1250
BOOST DIODE CURRENT (mA)
1500
85
–55
–25
3971 G22
5
35
65
95
TEMPERATURE (°C)
125
3971 G23
Transient Load Response,
Load Current Stepped from
0.5A to 1A
Switching Waveforms; Full
Frequency Continuous Operation
Switching Waveforms;
Burst Mode Operation
VOUT
100mV/
DIV
VSW
5V/DIV
VSW
5V/DIV
IL
500mA/
DIV
IL
500mA/DIV
IL
500mA/DIV
VOUT
20mV/DIV
VOUT
20mV/DIV
3.3V Start-Up and Dropout
3971 G27
1µs/DIV
VIN = 12V, VOUT = 3.3V
ILOAD = 1A
COUT = 22µF
3971 G26
5µs/DIV
VIN = 12V, VOUT = 3.3V
ILOAD = 10mA
COUT = 22µF
3971 G25
10µs/DIV
VIN = 12V, VOUT = 3.3V
COUT = 47µF
3971 G24
10µs/DIV
VIN = 12V, VOUT = 3.3V
COUT = 47µF
155
5V Start-Up and Dropout
3.3V Start-Up and Dropout
VIN
VIN
1V/DIV
1V/DIV
1V/DIV
VIN
VOUT
VOUT
0.5s/DIV
800kHz
3kΩ LOAD
3971 G31
0.5s/DIV
800kHz
6.7Ω LOAD
VOUT
3971 G32
0.5s/DIV
3971 G33
800kHz
5kΩ LOAD
3971fd
7
LT3971/LT3971-3.3/LT3971-5
TYPICAL PERFORMANCE CHARACTERISTICS
5V Start-Up and Dropout
PIN FUNCTIONS
3971 G34
1.6
5
1.2
4
INPUT VOLTAGE (V)
FEEDBACK VOLTAGE (V)
1V/DIV
VOUT
800kHz
10Ω LOAD
Minimum Input Voltage to Switch
Feedback Regulation Voltage
VIN
0.5s/DIV
TA = 25°C, unless otherwise noted.
0.8
0.4
0
2
2.5
3
3.5
4
INPUT VOLTAGE (V)
4.5
5
3971 G35
3
2
1
–55
–25
95
5
35
65
TEMPERATURE (°C)
125
155
3971 G36
(DFN, MSE10/MSE16)
BD (Pin 1/Pin 1): This pin connects to the anode of the
boost diode. The BD pin is normally connected to the output.
BOOST (Pin 2/Pin 3): This pin is used to provide a drive
voltage, higher than the input voltage, to the internal bipolar
NPN power switch.
SW (Pin 3/Pin 5): The SW pin is the output of an internal
power switch. Connect this pin to the inductor, catch diode,
and boost capacitor.
VIN (Pin 4/Pin 7): The VIN pin supplies current to the
LT3971’s internal circuitry and to the internal power switch.
This pin must be locally bypassed.
EN (Pin 5/Pin 8): The part is in shutdown when this pin
is low and active when this pin is high. The hysteretic
threshold voltage is 1.005V going up and 0.975V going
down. The EN threshold is only accurate when VIN is above
4.3V. If VIN is lower than 4.2V, ground EN to place the part
in shutdown. Tie to VIN if shutdown feature is not used.
FB (Pin 6, LT3971 Only/Pins 9, 10): The LT3971 regulates
the FB pin to 1.19V. Connect the feedback resistor divider
tap to this pin. Also, connect a phase lead capacitor between
FB and VOUT. Typically this capacitor is 10pF.
VOUT (Pin 6, LT3971-3.3 and LT3971-5 Only): The
LT3971‑3.3 and LT3971-5 regulate the VOUT pin to 3.3V and
5V respectively. This pin connects to the internal 10MΩ
feedback divider that programs the fixed output voltage.
SS (Pin 7/Pin 12): A capacitor is tied between SS and
ground to slowly ramp up the peak current limit of the
LT3971 on start-up. The soft-start capacitor is only actively
discharged when EN is low. The SS pin is released when
the EN pin goes high. Float this pin to disable soft-start.
For applications with input voltages above 25V, add a 100k
resistor in series with the soft-start capacitor.
RT (Pin 8/Pin 13): A resistor is tied between RT and ground
to set the switching frequency.
PG (Pin 9/Pin 14): The PG pin is the open-drain output of
an internal comparator. PGOOD remains low until the FB
pin is within 9% of the final regulation voltage. PGOOD is
valid when the LT3971 is enabled and VIN is above 4.3V.
SYNC (Pin 10/Pin 15): This is the external clock synchronization input. Ground this pin for low ripple Burst Mode
operation at low output loads. Tie to a clock source for
synchronization, which will include pulse-skipping at low
output loads. When in pulse-skipping mode, quiescent
current increases to 1.5mA.
GND (Exposed Pad Pin 11/Pin 16, Exposed Pad Pin 17):
Ground. The exposed pad must be soldered to PCB.
NC (None/Pins 2, 4, 6, 11): No Connect. These pins
are not connected to internal circuitry. Float these pins
to achieve FMEA fault tolerance. (See Fault Tolerance of
MS16E Package section.)
3971fd
8
LT3971/LT3971-3.3/LT3971-5
BLOCK DIAGRAM
VIN
C1
INTERNAL 1.19V REF
1V
EN
RT
–
+
VIN
+
–
Σ
SHDN
BD
SWITCH
LATCH
SLOPE COMP
OSCILLATOR
200kHz TO 2MHz
RT
BOOST
R
S
C3
Q
L1
SYNC
PG
ERROR AMP
+
–
+
–
1.09V
D1
1µA
SS
C5
SHDN
GND
R3
C4
R1
VOUT
FB
R2
C2
VC CLAMP
VC
R2
VOUT
SW
Burst Mode
DETECT
R1
LT3971-3.3
LT3971-5
ONLY
3991 BD
LT3971
ONLY
C5
LT3971-3.3: R1 = 6.39M, R2 = 3.61M
LT3971-5: R1 = 7.62M, R2 = 2.38M
3971fd
9
LT3971/LT3971-3.3/LT3971-5
OPERATION
The LT3971 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT,
sets an RS flip-flop, turning on the internal power switch.
An amplifier and comparator monitor the current flowing
between the VIN and SW pins, turning the switch off when
this current reaches a level determined by the voltage at
VC (see Block Diagram). An error amplifier measures the
output voltage through an external resistor divider tied to
the FB pin and servos the VC node. If the error amplifier’s
output increases, more current is delivered to the output;
if it decreases, less current is delivered. An active clamp
on the VC node provides current limit. The VC node is
also clamped by the voltage on the SS pin; soft-start is
implemented by generating a voltage ramp at the SS pin
using an external capacitor.
If the EN pin is low, the LT3971 is shut down and draws
700nA from the input. When the EN pin exceeds 1V, the
switching regulator will become active.
The switch driver operates from either VIN or from the
BOOST pin. An external capacitor is used to generate a
voltage at the BOOST pin that is higher than the input
supply. This allows the driver to fully saturate the internal
bipolar NPN power switch for efficient operation.
To further optimize efficiency, the LT3971 automatically
switches to Burst Mode operation in light load situations.
Between bursts, all circuitry associated with controlling
the output switch is shut down, reducing the input supply
current to 1.7μA. In a typical application, 2.8μA will be
consumed from the supply when regulating with no load.
The oscillator reduces the LT3971’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the output current during startup and overload.
The LT3971 contains a power good comparator which
trips when the FB pin is at 91% of its regulated value. The
PG output is an open-drain transistor that is off when the
output is in regulation, allowing an external resistor to pull
the PG pin high. Power good is valid when the LT3971 is
enabled and VIN is above 4.3V.
APPLICATIONS INFORMATION
To enhance efficiency at light loads, the LT3971 operates
in low ripple Burst Mode, which keeps the output capacitor
charged to the desired output voltage while minimizing
the input quiescent current. In Burst Mode operation the
LT3971 delivers single pulses of current to the output capacitor followed by sleep periods where the output power
is supplied by the output capacitor. When in sleep mode
the LT3971 consumes 1.7μA, but when it turns on all the
circuitry to deliver a current pulse, the LT3971 consumes
1.5mA of input current in addition to the switch current.
Therefore, the total quiescent current will be greater than
1.7μA when regulating.
As the output load decreases, the frequency of single current pulses decreases (see Figure 1) and the percentage
of time the LT3971 is in sleep mode increases, resulting
in much higher light load efficiency. By maximizing the
time between pulses, the converter quiescent current
1000
SWITCHING FREQUENCY (kHz)
Achieving Ultralow Quiescent Current
VIN = 12V
VOUT = 3.3V
800
600
400
200
0
0
20
40
60
80
LOAD CURRENT (mA)
100
120
3971 F01
Figure 1. Switching Frequency in Burst Mode Operation
gets closer to the 1.7μA ideal. Therefore, to optimize the
quiescent current performance at light loads, the current
in the feedback resistor divider and the reverse current
in the catch diode must be minimized, as these appear
to the output as load currents. Use the largest possible
3971fd
10
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
feedback resistors and a low leakage Schottky catch diode
in applications utilizing the ultralow quiescent current
performance of the LT3971. The feedback resistors should
preferably be on the order of MΩ and the Schottky catch
diode should have less than 1µA of typical reverse leakage at room temperature. These two considerations are
reiterated in the FB Resistor Network and Catch Diode
Selection sections.
It is important to note that another way to decrease the
pulse frequency is to increase the magnitude of each
single current pulse. However, this increases the output
voltage ripple because each cycle delivers more power to
the output capacitor. The magnitude of the current pulses
was selected to ensure less than 15mV of output ripple in
a typical application. See Figure 2.
VSW
5V/DIV
programmed by the RT resistor, and will be operating in
standard PWM mode. The transition between PWM and low
ripple Burst Mode operation will exhibit slight frequency
jitter, but will not disturb the output voltage.
To ensure proper Burst Mode operation, the SYNC pin
must be grounded. When synchronized with an external
clock, the LT3971 will pulse skip at light loads. The quiescent current will significantly increase to 1.5mA in light
load situations when synchronized with an external clock.
Holding the SYNC pin high yields no advantages in terms
of output ripple or minimum load to full frequency, so is
not recommended.
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the resistor
values according to:
IL
500mA/DIV
V
R1= R2 OUT − 1
1.19V
Reference designators refer to the Block Diagram. 1%
resistors are recommended to maintain output voltage
accuracy.
VOUT
20mV/DIV
5µs/DIV
3971 F02
VIN = 12V
VOUT = 3.3V
ILOAD = 10mA
Figure 2. Burst Mode Operation
While in Burst Mode operation, the burst frequency and
the charge delivered with each pulse will not change with
output capacitance. Therefore, the output voltage ripple
will be inversely proportional to the output capacitance.
In a typical application with a 22μF output capacitor, the
output ripple is about 10mV, and with a 47μF output capacitor the output ripple is about 5mV. The output voltage
ripple can continue to be decreased by increasing the
output capacitance.
At higher output loads (above 92mA for the front page
application) the LT3971 will be running at the frequency
The total resistance of the FB resistor divider should be
selected to be as large as possible to enhance low current
performance. The resistor divider generates a small load
on the output, which should be minimized to optimize the
low supply current at light loads.
When using large FB resistors, a 10pF phase lead capacitor
should be connected from VOUT to FB.
The LT3971-3.3 and LT3971-5 contain an internal 10M FB
resistor divider as well as an internal phase lead capacitor.
Setting the Switching Frequency
The LT3971 uses a constant frequency PWM architecture
that can be programmed to switch from 200kHz to 2MHz
by using a resistor tied from the RT pin to ground. A table
showing the necessary RT value for a desired switching
frequency is in Table 1.
3971fd
11
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
Table 1. Switching Frequency vs RT Value
SWITCHING FREQUENCY (MHz)
RT VALUE (kΩ)
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
255
118
71.5
49.9
35.7
28.0
22.1
17.4
14.0
11.0
Operating Frequency Tradeoffs
Selection of the operating frequency is a tradeoff between
efficiency, component size, minimum dropout voltage, and
maximum input voltage. The advantage of high frequency
operation is that smaller inductor and capacitor values may
be used. The disadvantages are lower efficiency, lower
maximum input voltage, and higher dropout voltage. The
highest acceptable switching frequency (fSW(MAX)) for a
given application can be calculated as follows:
fSW(MAX) =
VOUT + VD
tON(MIN)(VIN − VSW + VD )
where VIN is the typical input voltage, VOUT is the output
voltage, VD is the catch diode drop (~0.5V), and VSW is
the internal switch drop (~0.5V at max load). This equation
shows that slower switching frequency is necessary to
safely accommodate high VIN/VOUT ratio. Also, as shown
in the Input Voltage Range section, lower frequency allows
a lower dropout voltage. The input voltage range depends
on the switching frequency because the LT3971 switch has
finite minimum on and off times. The minimum switch on
and off times are strong functions of temperature. Use
the typical minimum on and off curves to design for an
application’s maximum temperature, while adding about
30% for part-to-part variation. The minimum and maximum
duty cycles that can be achieved taking minimum on and
off times into account are:
DCMIN = fSW tON(MIN)
DCMAX = 1− fSW tOFF(MIN)
where fSW is the switching frequency, the tON(MIN) is the
minimum switch on-time, and the tOFF(MIN) is the minimum
switch off-time. These equations show that duty cycle
range increases when switching frequency is decreased.
A good choice of switching frequency should allow
adequate input voltage range (see Input Voltage Range
section) and keep the inductor and capacitor values small.
Input Voltage Range
The minimum input voltage is determined by either the
LT3971’s minimum operating voltage of 4.3V or by its
maximum duty cycle (see equation in Operating Frequency
Tradeoffs section). The minimum input voltage due to
duty cycle is:
VIN(MIN) =
VOUT + VD
− VD + VSW
1− fSW tOFF(MIN)
where VIN(MIN) is the minimum input voltage, VOUT is
the output voltage, VD is the catch diode drop (~0.5V),
VSW is the internal switch drop (~0.5V at max load), fSW
is the switching frequency (set by RT), and tOFF(MIN) is
the minimum switch off-time. Note that higher switching frequency will increase the minimum input voltage.
If a lower dropout voltage is desired, a lower switching
frequency should be used.
The maximum input voltage for LT3971 applications
depends on switching frequency, the Absolute Maximum
Ratings of the VIN and BOOST pins, and the operating
mode. For a given application where the switching frequency and the output voltage are already selected, the
maximum input voltage (VIN(OP-MAX)) that guarantees
optimum output voltage ripple for that application can be
found by applying the following equation:
VIN(OP-MAX) =
VOUT + VD
–V +V
fSW • tON(MIN) D SW
where tON(MIN) is the minimum switch on-time. Note that
a higher switching frequency will decrease the maximum
operating input voltage. Conversely, a lower switching
frequency will be necessary to achieve normal operation
at higher input voltages.
The circuit will tolerate inputs above the maximum operating input voltage and up to the Absolute Maximum
3971fd
12
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
Ratings of the VIN and BOOST pins, regardless of chosen
switching frequency. However, during such transients
where VIN is higher than VIN(OP-MAX), the LT3971 will enter
pulse-skipping operation where some switching pulses are
skipped to maintain output regulation. The output voltage
ripple and inductor current ripple will be higher than in
typical operation. Do not overload when VIN is greater
than VIN(OP-MAX).
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is:
L=
VOUT + VD
fSW
where fSW is the switching frequency in MHz, VOUT is the
output voltage, VD is the catch diode drop (~0.5V) and L
is the inductor value in μH.
The inductor’s RMS current rating must be greater than the
maximum load current and its saturation current should be
about 30% higher. For robust operation in fault conditions
(start-up or short-circuit) and high input voltage (>30V),
the saturation current should be above 3.8A. To keep the
efficiency high, the series resistance (DCR) should be less
than 0.1Ω, and the core material should be intended for
high frequency applications. Table 2 lists several vendors
and suitable types.
The inductor value must be sufficient to supply the desired
maximum output current (IOUT(MAX)), which is a function
of the switch current limit (ILIM) and the ripple current.
IOUT(MAX) =ILIM –
ΔIL
2
The LT3971 limits its peak switch current in order to
protect itself and the system from overload faults. The
LT3971’s switch current limit (ILIM) is at least 2.4A at low
duty cycles and decreases linearly to 1.75A at DC = 0.8.
Table 2. Inductor Vendors
VENDOR
URL
PART SERIES
TYPE
Murata
www.murata.com
LQH55D
Open
TDK
www.componenttdk.com SLF7045
SLF10145
Shielded
Shielded
Toko
www.toko.com
D62CB
D63CB
D73C
D75F
Shielded
Shielded
Shielded
Open
Coilcraft
www.coilcraft.com
MSS7341
MSS1038
Shielded
Shielded
Sumida
www.sumida.com
CR54
CDRH74
CDRH6D38
CR75
Open
Shielded
Shielded
Open
When the switch is off, the potential across the inductor
is the output voltage plus the catch diode drop. This gives
the peak-to-peak ripple current in the inductor:
ΔIL =
(1− DC) • (VOUT + VD )
L • fSW
Where fSW is the switching frequency of the LT3971, DC is
the duty cycle and L is the value of the inductor. Therefore,
the maximum output current that the LT3971 will deliver
depends on the switch current limit, the inductor value,
and the input and output voltages. The inductor value may
have to be increased if the inductor ripple current does
not allow sufficient maximum output current (IOUT(MAX))
given the switching frequency, and maximum input voltage
used in the desired application.
The optimum inductor for a given application may differ
from the one indicated by this simple design guide. A larger
value inductor provides a higher maximum load current
and reduces the output voltage ripple. If your load is lower
than the maximum load current, than you can relax the
value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor,
or one with a lower DCR resulting in higher efficiency. Be
aware that if the inductance differs from the simple rule
above, then the maximum load current will depend on
the input voltage. In addition, low inductance may result
in discontinuous mode operation, which further reduces
3971fd
13
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
maximum load current. For details of maximum output current and discontinuous operation, see Linear Technology’s
Application Note 44. Finally, for duty cycles greater than
50% (VOUT/VIN>0.5), a minimum inductance is required to
avoid sub-harmonic oscillations. See Application Note 19.
One approach to choosing the inductor is to start with
the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then
use the equations above to check that the LT3971 will be
able to deliver the required output current. Note again
that these equations assume that the inductor current is
continuous. Discontinuous operation occurs when IOUT
is less than ΔIL/2.
Input Capacitor
Bypass the input of the LT3971 circuit with a ceramic
capacitor of X7R or X5R type. Y5V types have poor
performance over temperature and applied voltage, and
should not be used. A 4.7μF to 10μF ceramic capacitor
is adequate to bypass the LT3971 and will easily handle
the ripple current. Note that larger input capacitance is
required when a lower switching frequency is used (due
to longer on-times). If the input power source has high
impedance, or there is significant inductance due to
long wires or cables, additional bulk capacitance may be
necessary. This can be provided with a low performance
electrolytic capacitor.
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT3971 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 4.7μF capacitor is capable of this task, but only if it is
placed close to the LT3971 (see the PCB Layout section).
A second precaution regarding the ceramic input capacitor
concerns the maximum input voltage rating of the LT3971.
A ceramic input capacitor combined with trace or cable
inductance forms a high quality (under damped) tank
circuit. If the LT3971 circuit is plugged into a live supply,
the input voltage can ring to twice its nominal value, possibly exceeding the LT3971’s voltage rating. This situation
is easily avoided (see the Hot Plugging Safely section).
Output Capacitor and Output Ripple
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by the
LT3971 to produce the DC output. In this role it determines
the output ripple, so low impedance (at the switching
frequency) is important. The second function is to store
energy in order to satisfy transient loads and stabilize the
LT3971’s control loop. Ceramic capacitors have very low
equivalent series resistance (ESR) and provide the best
ripple performance. A good starting value is:
COUT =
100
VOUT fSW
where fSW is in MHz, and COUT is the recommended output
capacitance in μF. Use X5R or X7R types. This choice will
provide low output ripple and good transient response.
Transient performance can be improved with a higher
value capacitor. Increasing the output capacitance will
also decrease the output voltage ripple. A lower value of
output capacitor can be used to save space and cost but
transient performance will suffer.
When choosing a capacitor, look carefully through the
data sheet to find out what the actual capacitance is under
operating conditions (applied voltage and temperature). A
physically larger capacitor or one with a higher voltage rating
may be required. Table 3 lists several capacitor vendors.
Table 3. Recommended Ceramic Capacitor Vendors
MANUFACTURER
WEBSITE
AVX
www.avxcorp.com
Murata
www.murata.com
Taiyo Yuden
www.t-yuden.com
Vishay Siliconix
www.vishay.com
TDK
www.tdk.com
Catch Diode Selection
The catch diode (D1 from Block Diagram) conducts current only during switch off time. Average forward current
in normal operation can be calculated from:
ID(AVG) =IOUT
VIN – VOUT
VIN
where IOUT is the output load current. The only reason to
consider a diode with a larger current rating than necessary
3971fd
14
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
for nominal operation is for the worst-case condition of
shorted output. The diode current will then increase to the
typical peak switch current. Peak reverse voltage is equal
to the regulator input voltage. Use a diode with a reverse
voltage rating greater than the input voltage.
Table 4. Schottky Diodes. The Reverse Current Values Listed Are
Estimates Based Off of Typical Curves for Reverse Current
vs Reverse Voltage at 25°C.
IAVE
(A)
MBR0520L
20
0.5
MBR0540
40
0.5
620
MBRM120E
20
1
530
MBRM140
40
1
550
B0530W
30
0.5
B0540W
40
0.5
620
1
B120
20
1
500
1.1
B130
30
1
500
1.1
B140
40
1
500
1.1
B150
50
1
700
B220
20
2
500
20
B230
30
2
500
0.6
B140HB
40
1
DFLS240L
40
2
DFLS140
40
1.1
510
1
DFLS160
60
1
500
2.5
DFLS2100
100
2
770
B240
40
2
PART NUMBER
VF at 1A
(mV)
VF at 2A
(mV)
IR at VR =
20V 25°C
(µA)
VR
(V)
On Semiconductor
30
0.4
595
0.5
20
Diodes Inc.
15
0.4
1
500
4
860
0.01
500
0.45
Central Semiconductor
CMSH1 - 40M
40
1
500
CMSH1 - 60M
60
1
700
CMSH1 - 40ML
40
1
400
CMSH2 - 40M
40
2
550
CMSH2 - 60M
60
2
700
CMSH2 - 40L
40
2
400
CMSH2 - 40
40
2
500
CMSH2 - 60M
60
2
700
An additional consideration is reverse leakage current.
When the catch diode is reversed biased, any leakage
current will appear as load current. When operating under
light load conditions, the low supply current consumed
by the LT3971 will be optimized by using a catch diode
with minimum reverse leakage current. Low leakage
Schottky diodes often have larger forward voltage drops
at a given current, so a trade-off can exist between low
load and high load efficiency. Often Schottky diodes with
larger reverse bias ratings will have less leakage at a given
output voltage than a diode with a smaller reverse bias
rating. Therefore, superior leakage performance can be
achieved at the expense of diode size. Table 4 lists several
Schottky diodes and their manufacturers.
Ceramic Capacitors
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT3971 due to their piezoelectric nature.
When in Burst Mode operation, the LT3971’s switching
frequency depends on the load current, and at very light
loads the LT3971 can excite the ceramic capacitor at audio
frequencies, generating audible noise. Since the LT3971
operates at a lower current limit during Burst Mode operation, the noise is typically very quiet to a casual ear. If
this is unacceptable, use a high performance tantalum or
electrolytic capacitor at the output.
A final precaution regarding ceramic capacitors concerns
the maximum input voltage rating of the LT3971. As previously mentioned, a ceramic input capacitor combined
with trace or cable inductance forms a high quality (under
damped) tank circuit. If the LT3971 circuit is plugged into a
live supply, the input voltage can ring to twice its nominal
value, possibly exceeding the LT3971’s rating. This situation
is easily avoided (see the Hot Plugging Safely section).
BOOST and BD Pin Considerations
Capacitor C3 and the internal boost Schottky diode (see
the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases
a 0.47μF capacitor will work well. Figure 3 shows three
ways to arrange the boost circuit. The BOOST pin must
be more than 2.3V above the SW pin for best efficiency.
3971fd
15
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
For outputs of 3V and above, the standard circuit (Figure 3a)
is best. For outputs between 2.8V and 3V, use a 1μF boost
capacitor. A 2.5V output presents a special case because it
is marginally adequate to support the boosted drive stage
while using the internal boost diode. For reliable BOOST pin
operation with 2.5V outputs use a good external Schottky
diode (such as the ON Semi MBR0540), and a 1μF boost
capacitor (Figure 3b). For output voltages below 2.5V,
the boost diode can be tied to the input (Figure 3c), or to
another external supply greater than 2.8V. However, the
circuit in Figure 3a is more efficient because the BOOST pin
current comes from a lower voltage source. You must also
be sure that the maximum voltage ratings of the BOOST
and BD pins are not exceeded.
VIN
BD
VIN
BOOST
LT3971
4.7µF
GND
The minimum operating voltage of an LT3971 application
is limited by the minimum input voltage (4.3V) and by
the maximum duty cycle as outlined in the Input Voltage
Range section. For proper start-up, the minimum input
voltage is also limited by the boost circuit. If the input
voltage is ramped slowly, the boost capacitor may not
be fully charged. Because the boost capacitor is charged
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The minimum load generally goes to zero once the
circuit has started. Figure 4 shows a plot of minimum load
to start and to run as a function of input voltage. In many
cases the discharged output capacitor will present a load
to the switcher, which will allow it to start. The plots show
the worst-case situation where VIN is ramping very slowly.
C3
SW
5.0
VOUT
4.8
INPUT VOLTAGE (V)
4.6
(3a) For VOUT > 2.8V
VIN
D2
BD
VIN
4.7µF
GND
C3
SW
TO START
4.2
4.0
TO RUN
3.8
3.6
3.4 VOUT = 3.3V
TA = 25°C
3.2 L = 4.7µH
f = 800kHz
3.0
10
BOOST
LT3971
4.4
VOUT
100
LOAD CURRENT (mA)
1000
100
LOAD CURRENT (mA)
1000
6.4
6.2
VIN
BD
VIN
BOOST
LT3971
4.7µF
INPUT VOLTAGE (V)
(3b) For 2.5V < VOUT < 2.8V
GND
C3
VOUT
SW
5.8
5.6
5.4
5.2
5.0
3971 FO3
(3c) For VOUT < 2.5V; VIN(MAX) = 27V
Figure 3. Three Circuits for Generating the Boost Voltage
TO START
6.0
TO RUN
VOUT = 5V
TA = 25°C
L = 4.7µH
f = 800kHz
10
3971 F04
Figure 4. The Minimum Input Voltage Depends on
Output Voltage, Load Current and Boost Circuit
3971fd
16
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
For lower start-up voltage, the boost diode can be tied to
VIN; however, this restricts the input range to one-half of
the absolute maximum rating of the BOOST pin.
At light loads, the inductor current becomes discontinuous and this reduces the minimum input voltage to approximately 400mV above VOUT. At higher load currents,
the inductor current is continuous and the duty cycle is
limited by the maximum duty cycle of the LT3971, requiring
a higher input voltage to maintain regulation.
Enable Pin
Be aware that when the input voltage is below 4.3V, the
input current may rise to several hundred μA. And the part
may be able to switch at cold or for VIN(EN) thresholds less
than 7V. Figure 6 shows the magnitude of the increased
input current in a typical application with different programmed VIN(EN).
When operating in Burst Mode for light load currents, the
current through the VIN(EN) resistor network can easily be
greater than the supply current consumed by the LT3971.
Therefore, the VIN(EN) resistors should be large to minimize
their effect on efficiency at low loads.
The LT3971 is in shutdown when the EN pin is low and
active when the pin is high. The rising threshold of the
EN comparator is 1.01V, with 30mV of hysteresis. The EN
pin can be tied to VIN if the shutdown feature is not used.
INPUT CURRENT (µA)
R3
VIN(EN) =
+1
R4
INPUT CURRENT (µA)
400
Adding a resistor divider from VIN to EN programs the
LT3971 to regulate the output only when VIN is above a
desired voltage (see Figure 5). Typically, this threshold,
VIN(EN), is used in situations where the input supply is current limited, or has a relatively high source resistance. A
switching regulator draws constant power from the source,
so source current increases as source voltage drops. This
looks like a negative resistance load to the source and can
cause the source to current limit or latch low under low
source voltage conditions. The VIN(EN) threshold prevents
the regulator from operating at source voltages where the
problems might occur. This threshold can be adjusted by
setting the values R3 and R4 such that they satisfy the
following equation:
where output regulation should not start until VIN is above
VIN(EN). Due to the comparator’s hysteresis, switching will
not stop until the input falls slightly below VIN(EN).
R3
EN
1V
+
–
300
200
100
0
SHDN
R4
3971 F05
Figure 5. Programmed Enable Threshold
0
1
2
3
VIN(EN) = 12V
R3 = 11M
R4 = 1M
4 5 6 7 8 9 10 11 12
INPUT VOLTAGE (V)
6V VIN(EN) Input Current
500
400
300
200
100
0
LT3971
VIN
12V VIN(EN) Input Current
500
0
1
VIN(EN) = 6V
R3 = 5M
R4 = 1M
2
3
4
INPUT VOLTAGE (V)
5
6
3971 F06
Figure 6. Input Current vs Input Voltage
for a Programmed VIN(EN) of 6V and 12V
3971fd
17
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
Soft-Start
The SS pin can be used to soft-start the LT3971 by throttling
the maximum input current during start-up. An internal 1μA
current source charges an external capacitor generating a
voltage ramp on the SS pin. The SS pin clamps the internal
VC node, which slowly ramps up the current limit. Maximum
current limit is reached when the SS pin is about 1.5V or
higher. By selecting a large enough capacitor, the output
can reach regulation without overshoot. For applications
with input voltages above 25V, a 100k resistor in series
with the soft-start capacitor is recommended. Figure 7
shows start-up waveforms for a typical application with
a 10nF capacitor on SS for a 3.3Ω load when the EN pin
is pulsed high for 13ms.
The external SS capacitor is only actively discharged when
EN is low. With EN low, the external SS cap is discharged
through approximately 150Ω. The EN pin needs to be low
long enough for the external cap to completely discharge
through the 150Ω pull-down prior to start-up.
VSS
1V/DIV
The LT3971 will not enter Burst Mode operation at low
output loads while synchronized to an external clock, but
instead will pulse skip to maintain regulation.
The LT3971 may be synchronized over a 250kHz to 2MHz
range. The RT resistor should be chosen to set the LT3971
switching frequency 20% below the lowest synchronization
input. For example, if the synchronization signal will be
250kHz and higher, the RT should be selected for 200kHz.
To assure reliable and safe operation the LT3971 will only
synchronize when the output voltage is near regulation as
indicated by the PG flag. It is therefore necessary to choose
a large enough inductor value to supply the required output
current at the frequency set by the RT resistor (see the
Inductor Selection section). The slope compensation is set
by the RT value, while the minimum slope compensation
required to avoid subharmonic oscillations is established
by the inductor size, input voltage, and output voltage.
Since the synchronization frequency will not change the
slopes of the inductor current waveform, if the inductor
is large enough to avoid subharmonic oscillations at the
frequency set by RT, than the slope compensation will be
sufficient for all synchronization frequencies.
Shorted and Reversed Input Protection
VOUT
2V/DIV
IL
0.5A/DIV
2ms/DIV
3971 F07
Figure 7. Soft-Start Waveforms for Front-Page Application
with 10nF Capacitor on SS. EN is Pulsed High for About
13ms with a 3.3Ω Load Resistor
Synchronization
To select low ripple Burst Mode operation, tie the SYNC
pin below 0.6V (this can be ground or a logic low output).
Synchronizing the LT3971 oscillator to an external frequency can be done by connecting a square wave (with
20% to 80% duty cycle) to the SYNC pin. The square
wave amplitude should have valleys that are below 0.6V
and peaks above 1.0V (up to 6V).
If the inductor is chosen so that it won’t saturate excessively, a LT3971 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT3971 is absent. This may occur in battery charging applications or in battery backup systems where a battery
or some other supply is diode ORed with the LT3971’s
output. If the VIN pin is allowed to float and the EN pin
is held high (either by a logic signal or because it is tied
to VIN), then the LT3971’s internal circuitry will pull its
quiescent current through its SW pin. This is fine if your
system can tolerate a few μA in this state. If you ground
the EN pin, the SW pin current will drop to essentially
zero. However, if the VIN pin is grounded while the output
is held high, regardless of EN, parasitic diodes inside the
LT3971 can pull current from the output through the SW
pin and the VIN pin. Figure 8 shows a circuit that will run
only when the input voltage is present and that protects
against a shorted or reversed input.
3971fd
18
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
D4
MBRS140
VIN
VIN
BOOST
EN
SW
L1
C2
VOUT
VOUT
LT3971
GND
BD
FB
+
BACKUP
RPG
C4
3971 F07
Figure 8. Diode D4 Prevents a Shorted Input from Discharging a
Backup Battery Tied to the Output. It Also Protects the Circuit from
a Reversed Input. The LT3971 Runs Only When the Input is Present
PCB Layout
For proper operation and minimum EMI, care must be taken
during printed circuit board layout. Figure 9 shows the
recommended component placement with trace, ground
plane and via locations. Note that large, switched currents
flow in the LT3971’s VIN and SW pins, the catch diode
(D1), and the input capacitor (C1). The loop formed by
these components should be as small as possible. These
components, along with the inductor and output capacitor,
should be placed on the same side of the circuit board,
and their connections should be made on that layer. Place
a local, unbroken ground plane below these components.
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB and RT nodes small so that the ground
traces will shield them from the SW and BOOST nodes.
The Exposed Pad on the bottom of the package must be
soldered to ground so that the pad acts as a heat sink. To
keep thermal resistance low, extend the ground plane as
much as possible, and add thermal vias under and near
the LT3971 to additional ground planes within the circuit
board and on the bottom side.
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT3971 circuits. However, these capacitors can cause problems if the LT3971 is plugged into
a live supply. The low loss ceramic capacitor, combined
with stray inductance in series with the power source,
forms an under damped tank circuit, and the voltage at
C5
D1
GND
RT
C3
R1
R2
C1
GND
3971 F09
VIAS TO LOCAL GROUND PLANE
VIAS TO VOUT
VIAS TO SYNC
VIAS TO RUN/SS
VIAS TO PG
VIAS TO VIN
OUTLINE OF LOCAL
GROUND PLANE
Figure 9. A Good PCB Layout Ensures Proper, Low EMI Operation
the VIN pin of the LT3971 can ring to twice the nominal
input voltage, possibly exceeding the LT3971’s rating and
damaging the part. If the input supply is poorly controlled
or the user will be plugging the LT3971 into an energized
supply, the input network should be designed to prevent
this overshoot. See Linear Technology Application Note
88 for a complete discussion.
High Temperature Considerations
For higher ambient temperatures, care should be taken in
the layout of the PCB to ensure good heat sinking of the
LT3971. The Exposed Pad on the bottom of the package
must be soldered to a ground plane. This ground should be
tied to large copper layers below with thermal vias; these
layers will spread heat dissipated by the LT3971. Placing
additional vias can reduce thermal resistance further. The
maximum load current should be derated as the ambient
temperature approaches the maximum junction rating.
Power dissipation within the LT3971 can be estimated by
calculating the total power loss from an efficiency measurement and subtracting the catch diode loss and inductor
3971fd
19
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
loss. The die temperature is calculated by multiplying the
LT3971 power dissipation by the thermal resistance from
junction to ambient.
There are four items which require consideration in terms
of the application circuit to achieve fault tolerance: VIN-EN
pin short, SYNC-GND pin short, SYNC-PG pin short, and
PG-RT pin short. If the EN pin is driven with a logic input,
then a series resistor is needed to protect the circuit generating the logic input in the event of an EN-VIN pin short.
If the SYNC pin is driven with a clock, a series resistor is
needed so that the clock source, which may be going to
other devices, is not pulled down in the event of a SYNCGND pin short. If the PG pull-up resistor is connected to a
voltage source higher than 6V, then the PG resistor needs
to be large enough such that the resistor divider formed
by a PG-RT pin short does not violate the RT pin absolute
maximum. Likewise, a SYNC resistor to GND is needed so
that the resistor divider formed by a PG-SYNC pin short
does not violate the SYNC pin absolute maximum. This
means that typical applications where EN is tied to VIN,
SYNC is grounded, and PG is floating or connected to a
pull-up resistor to an output less than 6V are already set
up for fault tolerance. Figure 10, shows how fault tolerance can be achieved when PG, EN, and SYNC features
are used in a high output voltage application.
Also keep in mind that the leakage current of the power
Schottky diode goes up exponentially with junction temperature. When the power switch is closed, the power
Schottky diode is in parallel with the power converter’s
output filter stage. As a result, an increase in a diode’s
leakage current results in an effective increase in the load,
and a corresponding increase in input power. Therefore,
the catch Schottky diode must be selected with care to
avoid excessive increase in light load supply current at
high temperatures.
Fault Tolerance of MS16E Package
The MS16E package is designed to tolerate single fault
conditions. Shorting two adjacent pins together or leaving
one single pin floating does not raise the output voltage
or cause damage to the LT3971 regulator. However, the
application circuit must meet a few requirements discussed
in this section in order to achieve fault tolerance.
Tables 5 and 6 show the effects that result from shorting
adjacent pins or from a floating pin, respectively.
VIN
15V TO 38V
OFF ON
100k
VIN
BOOST
EN
SW
10µF
0.47µF 10µH
LT3971
1nF
150k
SS
PG
RT
BD
49.9k
SYNC
GND
PGOOD
1M
FB
110k
CLOCK IN
10pF
10µF
VOUT
12V
1.2A
49.9k
fSW = 800kHz
3971 F10
Figure 10. Fault Tolerant Application with EN, SYNC and PG Functions in Use when Using the MS16E Package
3971fd
20
LT3971/LT3971-3.3/LT3971-5
APPLICATIONS INFORMATION
Table 5: Effects of Pin Shorts
PINS
EFFECT
VIN-EN
No effect. In most applications, EN is tied to VIN. If EN is driven with a logic signal, a series resistor is recommended to protect the circuit
generating the logic signal from the full VIN voltage.
SS-RT
VOUT may fall below regulation voltage. The switching frequency will be increased and the current limit will be reduced.
RT-PG
No effect if PG is floated.
VOUT will fall below regulation if PG is connected to the output with a resistor pull-up as long as the resister divider formed by the PG pin
pull-up and the RT resistor prevents the RT pin absolute maximum from being violated. (see discussion in Fault Tolerance section)
In both cases, the switching frequency will be significantly increased if the output goes below regulation, which may cause the LT3975 to
go into pulse-skipping mode if the minimum on-time is violated.
PG-SYNC
No effect if PG is floated.
No effect if PG is connected to the output with a resistor pull-up as long as there is a resistor to GND on the SYNC pin or the SYNC pin is
tied to GND. This is to ensure that the resistor divider formed by the PG pin pull-up and the SYNC pin resistor to GND prevents the SYNC
pin absolute maximum from being violated. (see discussion in Fault Tolerance section)
SYNC-GND No effect. If the SYNC pin is driven with a clock, a series resistor is recommended to prevent the clock source from getting shorted out.
Table 6: Effects of Floating Pins
PIN
EFFECT
SS
No effect; soft-start feature will not function.
BD
VOUT may fall below regulation voltage. With the BD pin disconnected, the boost capacitor cannot be charged and thus the power switch
cannot fully saturate, which increases power dissipation.
BOOST
VOUT may fall below regulation voltage. With the BOOST pin disconnected, the boost capacitor cannot be charged and thus the power switch
cannot fully saturate, which increases power dissipation.
SW
VOUT will fall below regulation voltage.
VIN
VOUT will fall below regulation voltage.
EN
VOUT may fall below regulation voltage. Part may work normally or be shutdown depending on how the application circuit couples to the
floating EN pin.
RT
VOUT may fall below regulation voltage.
PG
No effect.
SYNC
FB
GND
No effect. The LT3971 may be in Burst Mode operation or pulse-skipping mode depending on how the application circuit couples to the
floating SYNC pin.
No effect; there are two FB pins.
No effect; there are two GND connections. If exposed pad is floated, thermal performance will be degraded.
Other Linear Technology Publications
Application Notes 19, 35 and 44 contain more detailed
descriptions and design information for buck regulators
and other switching regulators. The LT1376 data sheet
has a more extensive discussion of output ripple, loop
compensation and stability testing. Design Note 318
shows how to generate a bipolar output supply using a
buck regulator.
3971fd
21
LT3971/LT3971-3.3/LT3971-5
TYPICAL APPLICATIONS
5V Step-Down Converter
VIN
7V TO 38V
VIN
EN
OFF ON
BOOST
0.47µF
PG
4.7µF
SS
LT3971
SW
4.7µH
RT
BD
49.9k
GND
SYNC
10pF
22µF
309k
f = 800kHz
VOUT
5V
1.2A
1M
FB
3971 TA02
3.3V Step Down Converter
No Load Supply Current
VIN
4.5V TO 38V
4.0
0.47µF
PG
SS
4.7µF
3.5
BOOST
LT3971
4.7µH
SW
INPUT CURRENT (µA)
VIN
EN
OFF ON
VOUT
3.3V
1.2A
RT
BD
49.9k
SYNC
GND
10pF
1.78M
FB
3.0
2.5
2.0
1.5
22µF
1M
1.0
10
0
3971 TA11
20
30
INPUT VOLTAGE (V)
40
3971 TA11b
5V Step-Down Converter
2.5V Step-Down Converter
VIN
7V TO 38V
VIN
4.3V TO 38V
VIN
EN
OFF ON
VIN
BOOST
0.47µF
PG
4.7µF
SS
LT3971-5
SW
EN
OFF ON
BOOST
1µF
PG
4.7µH
4.7µF
SS
RT
LT3971
SW
4.7µH
RT
BD
49.9k
SYNC
f = 800kHz
GND
BD
VOUT
22µF
3971 TA03
VOUT
5V
1.2A
118k
SYNC
f = 400kHz
GND
10pF
1M
FB
909k
47µF
VOUT
2.5V
1.2A
3971 TA04
3971fd
22
LT3971/LT3971-3.3/LT3971-5
TYPICAL APPLICATIONS
1.8V Step-Down Converter
12V Step-Down Converter
VIN
4.3V TO 27V
VIN
15V TO 38V
VIN
BD
EN
OFF ON
VIN
BOOST
0.47µF
PG
SS
4.7µF
BOOST
0.47µF
PG
4.7µH
SW
LT3971
EN
OFF ON
10µF
SS
RT
SW
LT3971
10µH
RT
10pF
118k
SYNC
GND
BD
VOUT
1.8V
1.2A
511k
FB
100µF
1M
f = 400kHz
49.9k
GND
SYNC
10pF
110k
f = 800kHz
3971 TA05
VOUT
12V
1.2A
1M
FB
10µF
3971 TA06
3.3V Step-Down Converter with Undervoltage Lockout, Soft-Start, and Power Good
VIN
6V TO 38V
5M
VIN
BOOST
EN
0.47µF
SW
4.7µF
SS
100k
4.7µH
150k
LT3971
RT
PG
BD
1M
PGOOD
10pF
1nF
49.9k
SYNC
GND
1M
FB
562k
f = 800kHz
VOUT
3.3V
1.2A
22µF
3971 TA07
5V, 2MHz Step-Down Converter with Soft-Start
VIN
9V TO 25V
VIN
EN
OFF ON
BOOST
0.47µF
PG
SS
2.2µF
LT3971
SW
2.2µH
RT
BD
1nF
11k
SYNC
f = 2MHz
GND
10pF
1M
FB
309k
22µF
VOUT
5V
1.2A
3971 TA08
3971fd
23
LT3971/LT3971-3.3/LT3971-5
TYPICAL APPLICATIONS
4V Step-Down Converter with a High Impedance Input Source
+
–
11M
24V
+
VIN
EN
CBULK
100µF
1M
BOOST
0.47µF
PG
SS
4.7µF
LT3971
SW
* AVERAGE OUTPUT POWER CANNOT
EXCEED THAT WHICH CAN BE PROVIDED
BY HIGH IMPEDANCE SOURCE.
NAMELY,
V2
•η
POUT(MAX) =
4R
4.7µH
RT
BD
1nF
10pF
49.9k
SYNC
GND
VOUT
4V
1.2A*
1M
FB
412k
f = 800kHz
100µF
3971 TA09a
Sourcing a Maximum Load Pulse
WHERE V IS VOLTAGE OF SOURCE, R IS
INTERNAL SOURCE IMPEDANCE, AND η IS
LT3971 EFFICIENCY. MAXIMUM OUTPUT
CURRENT OF 1.2A CAN BE SUPPLIED FOR A
SHORT TIME BASED ON THE ENERGY
WHICH CAN BE SOURCED BY THE BULK
INPUT CAPACITANCE.
Start-Up from High Impedance Input Source
VOUT
200mV/DIV
VIN
1V/DIV
VIN
5V/DIV
VOUT
2V/DIV
IL
1A/DIV
IL
500mA/DIV
3971 TA09c
2ms/DIV
3971 TA09b
500µs/DIV
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1699 Rev C)
R = 0.125
TYP
6
0.40 ± 0.10
10
0.70 ±0.05
3.55 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
3.00 ±0.10
(4 SIDES)
1.65 ± 0.10
(2 SIDES)
PIN 1 NOTCH
R = 0.20 OR
0.35 × 45°
CHAMFER
PIN 1
PACKAGE TOP MARK
OUTLINE (SEE NOTE 6)
0.25 ± 0.05
0.50
BSC
2.38 ±0.05
(2 SIDES)
0.200 REF
0.75 ±0.05
0.00 – 0.05
5
1
(DD) DFN REV C 0310
0.25 ± 0.05
0.50 BSC
2.38 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
3971fd
24
LT3971/LT3971-3.3/LT3971-5
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MSE Package
10-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1664 Rev H)
BOTTOM VIEW OF
EXPOSED PAD OPTION
1.88 ± 0.102
(.074 ± .004)
5.23
(.206)
MIN
1
0.889 ± 0.127
(.035 ± .005)
0.05 REF
10
0.305 ± 0.038
(.0120 ± .0015)
TYP
RECOMMENDED SOLDER PAD LAYOUT
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
10 9 8 7 6
DETAIL “A”
0° – 6° TYP
1 2 3 4 5
GAUGE PLANE
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
0.18
(.007)
0.497 ± 0.076
(.0196 ± .003)
REF
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
0.254
(.010)
0.29
REF
1.68
(.066)
1.68 ± 0.102 3.20 – 3.45
(.066 ± .004) (.126 – .136)
0.50
(.0197)
BSC
1.88
(.074)
SEATING
PLANE
0.86
(.034)
REF
1.10
(.043)
MAX
0.17 – 0.27
(.007 – .011)
TYP
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD
SHALL NOT EXCEED 0.254mm (.010") PER SIDE.
0.1016 ± 0.0508
(.004 ± .002)
MSOP (MSE) 0911 REV H
3971fd
25
LT3971/LT3971-3.3/LT3971-5
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev E)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
5.23
(.206)
MIN
2.845 ±0.102
(.112 ±.004)
0.889 ±0.127
(.035 ±.005)
8
1
1.651 ±0.102
(.065 ±.004)
1.651 ±0.102 3.20 – 3.45
(.065 ±.004) (.126 – .136)
0.305 ±0.038
(.0120 ±.0015)
TYP
16
0.50
(.0197)
BSC
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
9
NO MEASUREMENT PURPOSE
0.280 ±0.076
(.011 ±.003)
REF
16151413121110 9
DETAIL “A”
0° – 6° TYP
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
4.90 ±0.152
(.193 ±.006)
GAUGE PLANE
0.53 ±0.152
(.021 ±.006)
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
1234567 8
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.86
(.034)
REF
0.1016 ±0.0508
(.004 ±.002)
MSOP (MSE16) 0911 REV E
3971fd
26
LT3971/LT3971-3.3/LT3971-5
REVISION HISTORY
REV
DATE
DESCRIPTION
A
2/11
Added fixed voltage options LT3971-3.3 and LT3971-5 reflected throughout data sheet
B
8/11
Added fixed voltage options LT3971-3.3 and LT3971-5 in DFN package
C
10/11
Modified Note 4
D
7/12
PAGE NUMBER
1 through 24
2
4
Add Start-Up and Dropout, Feedback Regulation curves to the Typical Performance Characteristics
7, 8
Added MSOP-16E package option with enhanced pin-to-pin fault tolerance
1, 2
Clarified pin function for MSOP-16E package option
Clarified saturation current at 3.8A
Clarified enhanced pin-to-pin fault tolerance
8
13
20, 21
3971fd
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LT3971/LT3971-3.3/LT3971-5
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ISD
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