LT3988
Dual 60V Monolithic 1A
Step-Down Switching
Regulator
Description
Features
Wide Input Range:
Operation from 4.1V to 60V
Overvoltage Lockout Protects Circuit Through
80V Transients
n Two 1A Output Switching Regulators with Internal
Power Switches
n Short Circuit Robust
n Adjustable 250kHz to 2.5MHz Switching Frequency,
Synchronizable Over the Full Range
n Integrated Boost Diodes
n Integrated Loop Compensation
n Anti-Phase Switching Reduces Ripple
n Low Shutdown I ( 125°C tON(MIN) =
200ns) and tOFF(MIN) = 240ns. fMAX1 is the frequency at
which the minimum duty cycle is exceeded. The regulator
will skip ON pulses in order to reduce the overall duty cycle
3988f
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LT3988
Applications Information
at frequencies above fMAX1. It will continue to regulate but
with increased inductor current and increased output ripple.
fMAX2 is the frequency at which the maximum duty cycle
is exceeded. If there is sufficient charge on the BOOST
capacitor, the regulator will skip OFF periods to increase
the overall duty cycle at frequencies above fMAX2. Note
that the restriction on the operating input voltage refers
to steady-state limits to keep the output in regulation;
the circuit will tolerate input voltage transients up to the
absolute maximum rating.
Switching Frequency
Once the upper and lower bounds for the switching
frequency are found from the duty cycle requirements,
the frequency may be set within those bounds. Lower
frequencies result in lower switching losses, but require
larger inductors and capacitors. The user must decide
the best trade-off.
The switching frequency is set by a resistor connected from
the RT pin to ground, or by forcing a clock signal into the
SYNC pin. The LT3988 applies a voltage across this resistor
and uses the current to set the oscillator speed. The RT
resistor value for a given switching frequency is given by:
RT =
1.31 46.56
+
– 7.322
f
f2
250kHz ≤ f ≤ 2.5MHz
where f is in MHz and RT is in kΩ.
The frequency sync signal will support VIH logic levels from
1.5V to 5V CMOS or TTL. The duty cycle is not important,
but it needs a minimum on time of 100ns and a minimum
off time of 100ns. RT should be set to provide a frequency
within ±25% of the final sync frequency.
The slope recovery circuit sets the slope compensation
to the appropriate value for the synchronized frequency.
Choose the inductor value based on the lowest potential
switching frequency.
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is:
L=
VOUT + VF
0.6A • f
where VF is the voltage drop of the catch diode (~0.4V) and f
is in MHz. The inductor’s RMS current rating must be greater
than the maximum load current and its saturation current
Table 1. Inductors
MFG
URL
PART SERIES
INDUCTANCE RANGE (µH)
SIZE (mm) (L × W × H)
Coilcraft
http://www.coilcraft.com
XPL7030
XFL4020
XAL50XX
0.13 to 22
1 to 4.7
0.16 to 22
7×7×3
4 × 4 × 2.15
5.28 × 5.48 × 5.1
Cooper
http://www.cooperbussmann.com
DRA74
DR1040
0.33 to 1000
1.5 to 330
7.6 × 7.6 × 4.35
10.5 × 10.3 × 4
CWS
http://www.coilws.com
SP-0703
SP-0704
SB-1004
3 to 100
2.2 to 100
10 to 1500
7×7×3
7×7×4
10.1 × 10.1 × 4.5
Murata
http://www.murata.com
LQH55D
LQH6PP
LQH88P
0.12 to 10000
1 to 100
1 to 100
5 × 5.7 × 4.7
6 × 6 × 4.3
8 × 8 × 3.8
Sumida
http://www.sumida.com
CDMC6D28
CDEIR8D38F
0.2 to 4.7
4 to 22
7.25 × 6.7 × 3
8.5 × 8.3 × 4
Toko
http://www.toko.co.jp
DS84LCB
FDV0620
1 to 100
0.2 to 4.7
8.4 × 8.3 × 4
6.7 × 7.4 × 2
Vishay
http://www.vishay.com
IHLP-2020AB-11
IHLP-2020BZ-11
IHLP-2525CZ-11
0.1 to 4.7
0.1 to 10
1 to 22
5.49 × 5.18 × 1.2
5.49 × 5.18 × 2
6.86 × 6.47 × 3
Würth
http://www.we-online.de
WE-PD2-S
WE-PD-M
WE-PD2-XL
1 to 68
1 to 1000
10 to 820
4 × 4.5 × 3.2
7.3 × 7.3 × 4.5
9 × 10 × 5.4
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LT3988
Applications Information
should be at least 30% higher. For highest efficiency, the
series resistance (DCR) should be less than 0.1Ω. Table 1
lists several vendors and types that are suitable.
The current in the inductor is a triangle wave with an
average value equal to the load current. The peak switch
current is equal to the output current plus half the peak-topeak inductor ripple current. The LT3988 limits its switch
current in order to protect itself and the system from
overload faults. Therefore, the maximum output current
that the LT3988 will deliver depends on the switch current
limit, the inductor value and the input and output voltages.
When the switch is off, the potential across the inductor
is the output voltage plus the catch diode drop. This gives
the peak-to-peak ripple current in the inductor:
∆IL = (1– DC) •
VOUT + VF
L•f
where f is the switching frequency of the LT3988 and L is
the value of the inductor. In continuous mode, the peak
inductor and switch current is:
ISWPK =ILPK =
∆IL
+I
2 L
To maintain output regulation, this peak current must
be less than the LT3988’s switch current limit, ILIM. For
both switches, ILIM is at least 1.5A at low duty cycle and
decreases linearly to 1.1A at DC = 90%. (See chart in the
Typical Performance Characteristics section).
The minimum inductance can now be calculated as:
LMIN =
1– DCMIN VOUT + VF
•
2•f
ILIM – IOUT
However, it’s generally better to use an inductor larger
than the minimum value. The minimum inductor has large
ripple currents which increase core losses and require
large output capacitors to keep output voltage ripple low.
This analysis is valid for continuous mode operation (IOUT >
∆IL/2). For details of maximum output current in discontinuous mode operation, see Linear Technology’s Application
Note AN44. Finally, for duty cycles greater than 50% (VOUT/
VIN > 0.5), a minimum inductance is required to avoid
subharmonic oscillations. This minimum inductance is:
LMIN =
VOUT + VF
1.25A • f
with LMIN in μH and f in MHz.
For robust operation under fault conditions at input voltages of 40V or greater, use an inductor value of 47µH or
larger and a clock rate of 1MHz or lower.
Output Capacitor Selection
The output capacitor filters the inductor current to generate
an output with low voltage ripple. It also stores energy in
order to satisfy transient loads and stabilize the LT3988’s
control loop. Because the LT3988 operates at a high
frequency, minimal output capacitance is necessary. In
addition, the control loop operates well with or without
the presence of output capacitor series resistance (ESR).
Ceramic capacitors, which achieve very low output ripple
and small circuit size, are therefore an option. You can
estimate output ripple with the following equations:
VRIPPLE =
∆IL
8 • f • COUT for ceramic capacitors
and
V
= ∆IL • ESR for electrolytic capacitors
RIPPLE
(tantalum and aluminum)
where ΔIL is the peak-to-peak ripple current in the inductor.
The RMS content of this ripple is very low so the RMS
current rating of the output capacitor is usually not of
concern. It can be estimated with the formula:
IC(RMS) =
∆IL
12
Another constraint on the output capacitor is that it must
have greater energy storage than the inductor; if the stored
energy in the inductor transfers to the output, the resulting
voltage step should be small compared to the regulation
voltage. For a 5% overshoot, this requirement indicates:
I
COUT > 10 • L • LIM
VOUT
2
3988f
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LT3988
Applications Information
The low ESR and small size of ceramic capacitors make
them the preferred type for LT3988 applications. Not all
ceramic capacitors are the same, however. Many of the
higher value capacitors use poor dielectrics with high
temperature and voltage coefficients. In particular, Y5V
and Z5U types lose a large fraction of their capacitance
with applied voltage and at temperature extremes. Because
loop stability and transient response depend on the value
of COUT, this loss may be unacceptable. Use X7R and X5R
types.
Electrolytic capacitors are also an option. The ESRs of
most aluminum electrolytic capacitors are too large to
deliver low output ripple. Tantalum, as well as newer,
lower-ESR organic electrolytic capacitors intended for
power supply use are suitable. Choose a capacitor with a
low enough ESR for the required output ripple. Because
the volume of the capacitor determines its ESR, both the
size and the value will be larger than a ceramic capacitor
that would give similar ripple performance. One benefit
is that the larger capacitance may give better transient
response for large changes in load current. Table 2 lists
several capacitor vendors.
Table 2. Low ESR Surface Mount Capacitors
MFG
TYPE
SERIES
AVX
Ceramic
Tantalum
TPS
Ceramic
X7R, 1812 MLCC
Kemet
Johansen
Tantalum
Tantalum Organic
Aluminum Organic
T491,T494,T498
T520,T521,T528
A700
Panasonic
Aluminum Organic
SP CAP
Sanyo
Tantalum
Aluminum Organic
POSCAP
Taiyo-Yuden
Ceramic
TDK
Ceramic
Diode Selection
The catch diode (D1 from Figure 1) conducts the inductor
current during the switch off time. Use a Schottky diode
rated for 1A to 2A average current. Peak reverse voltage
across the diode is equal to the regulator input voltage.
Use a diode with a reverse voltage rating greater than the
input voltage. The OVLO function of the LT3988 turns off
the switch when VIN > 64V (typ) allowing use of Schottky
12
diodes with a 70V rating for input voltages up to 80V. Table 3
lists several Schottky diodes and their manufacturers.
Table 3. Schottky Diodes
PART NUMBER
VR
(V)
IAVG
(A)
VF AT 1A
(mV)
40
1
490
VF AT 2A
(mV)
On Semiconductor
NSR10F40NXT5G
MBRA160T3
60
1
510
MBRS190T3
90
1
750
MBRS260T3G
60
2
40
1
430
Diodes Inc
B140
500
B160
60
1
700
B170
70
1
790
B180
80
1
790
B260
60
2
700
B280
80
2
790
DFLS140L
40
1
550
DFLS160L
60
1
500
DFLS260
60
2
620
Boost Pin Considerations
The external capacitor and the internal diode tied to the
BOOST pin generate a voltage that is higher than the input
voltage. In most cases, a small ceramic capacitor will work
well. The capacitor value is a function of the switching
frequency, peak current, duty cycle and boost voltage.
Figure 2 shows three ways to arrange the boost circuit. The
BOOST pin must be more than 2.3V above the SW pin for
full efficiency. For outputs of 3.3V and higher, the standard
circuit (Figure 2a) is best. For lower output voltages, the BD
pin can be tied to the input (Figure 2b). The circuit in Figure
2a is more efficient because the BOOST pin current comes
from a lower voltage source. Finally, as shown in Figure
2c, the BD pin can be tied to another source that is at least
3V. For example, if you are generating 3.3V and 1.8V and
the 3.3V is on whenever the 1.8V is on, the BD pin can be
connected to the 3.3V output. (see Output Voltage Tracking).
Be sure that the maximum voltage at the BOOST pin is less
than 80V and the voltage difference between the BOOST
and SW pins is less than 30V. The minimum operating
voltage of an LT3988 application is limited by the internal
4V undervoltage lockout and by the maximum duty cycle.
3988f
LT3988
Applications Information
VIN3 > 3V
BD
VIN
BOOST
VIN
BD
C3
VOUT
SW
VIN
BOOST
VIN
GND
BD
C3
VOUT
SW
VIN
VIN
GND
VBOOST – VSW ≅ VOUT
MAX VBOOST ≅ VIN + VOUT
VOUT
SW
GND
VBOOST – VSW ≅ VIN
MAX VBOOST ≅ 2VIN
(2a)
BOOST
VBOOST – VSW ≅ VIN3
MAX VBOOST ≅ VIN3 + VIN
MIN VALUE FOR VIN3 = 3V
(2b)
3988 F02
(2c)
Figure 2. Generating the Boost Voltage
The boost circuit also limits the minimum input voltage for
proper start-up. If the input voltage ramps slowly, or the
LT3988 turns on when the output is already in regulation,
the boost capacitor may not be fully charged. Because
the boost capacitor charges with the energy stored in the
inductor, the circuit will rely on some minimum load current
to get the boost circuit running properly. This minimum
load will depend on input and output voltages, and on the
arrangement of the boost circuit. The minimum load current generally goes to zero once the circuit has started.
Figure 4 shows a plot of input voltage to start and to run
as a function of load current. Even without an output load
current, in many cases the discharged output capacitor
will present a load to the switcher that will allow it to start.
Converter with Backup Output Regulator
There is another situation to consider: systems where the
output will be held high when the input to the LT3988 is
absent. If the VIN pin is grounded while the output is held
high, then diodes inside the LT3988 can pull large currents
from the output through the SW and VIN pins. A Schottky
diode in series with the input to the LT3988, as shown in
Figure 3, will protect the LT3988 and the system from a
shorted or reversed input.
LT3988
D4
SW
VIN
GND
The boost current is generally small but can become significant at high duty cycles. The required boost current is:
3988 F03
Figure 3. Diode D4 Prevents a Shorted Input from
Discharging a Backup Battery Tied to the Output
V I
IBOOST = OUT OUT
VIN 40
Minimum Input Voltage, VOUT = 3.3V
5.5
Minimum Input Voltage, VOUT = 5V
7.0
TA = 25°C
TA = 25°C
6.6
INPUT VOLTAGE (V)
5.0
INPUT VOLTAGE (V)
VOUT
TO START
4.5
4.0
TO START
6.2
5.8
TO RUN
TO RUN
3.5
0
200
400
600
800
LOAD CURRENT (mA)
1000
3988 F04a
5.4
0
200
400
600
800
LOAD CURRENT (mA)
1000
3988 F04b
Figure 4. The Minimum Input Voltage Depends on Output Voltage, Load Current, and Boost Circuit
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LT3988
Applications Information
Input Capacitor Selection
Bypass the input of the LT3988 circuit with a 4.7μF or higher
ceramic capacitor of X7R or X5R type. A lower value or
a less expensive Y5V type will work if there is additional
bypassing provided by bulk electrolytic capacitors, or if the
input source impedance is low. The following paragraphs
describe the input capacitor considerations in more detail.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at
the LT3988 input and to force this switching current into a
tight local loop, minimizing EMI. The input capacitor must
have low impedance at the switching frequency to do this
effectively and it must have an adequate ripple current rating. With two switchers operating at the same frequency
but with different phases and duty cycles, calculating the
input capacitor RMS current is not simple; however, a
conservative value is the RMS input current for the phase
delivering the most power (VOUT • IOUT):
IIN(RMS) = IOUT •
VOUT ( VIN – VOUT )
VIN
I
< OUT
2
and is largest when VIN = 2VOUT (50% duty cycle). As
the second, lower power channel draws input current,
the input capacitor’s RMS current actually decreases as
the out-of-phase current cancels the current drawn by the
higher power channel. Considering that the maximum load
current from a single phase (if SW1 and SW2 are both at
maximum current) is ~1A, RMS ripple current will always
be less than 0.5A.
The high frequency of the LT3988 reduces the energy
storage requirements of the input capacitor, so that the
capacitance required is often less than 10μF. The combination of small size and low impedance (low equivalent
series resistance or ESR) of ceramic capacitors makes
them the preferred choice. The low ESR results in very
low voltage ripple. Ceramic capacitors can handle larger
magnitudes of ripple current than other capacitor types
of the same value.
An alternative to a high value ceramic capacitor is a lower
value along with a larger electrolytic capacitor, for example
a 1μF ceramic capacitor in parallel with a low ESR tantalum
capacitor. For the electrolytic capacitor, a value larger than
10μF will be required to meet the ESR and ripple current
requirements. Because the input capacitor is likely to see
high surge currents when the input source is applied, tantalum capacitors should be surge rated. The manufacturer
may also recommend operation below the rated voltage
of the capacitor. Be sure to place the 1μF ceramic as close
as possible to the VIN and GND pins on the IC for optimal
noise immunity.
A final caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plugging
the circuit into a live power source), this tank can ring,
doubling the input voltage and damaging the LT3988. The
solution is to either clamp the input voltage or dampen the
tank circuit by adding a lossy capacitor in parallel with the
ceramic capacitor. For details, see Application Note 88.
Frequency Compensation
The LT3988 uses current mode control to regulate the
output. This simplifies loop compensation. In particular, the
LT3988 does not depend on the ESR of the output capacitor
for stability, so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size. The LT3988
is internally compensated with the RC network tied to the
VC node. The internal compensation network is optimized
to provide stability over the full frequency range. Figure 5
shows an equivalent circuit for the LT3988 control loop.
The error amplifier is a transconductance amplifier with
LT3988
CURRENT MODE
POWER STAGE
OUT
gm = 2A/V
FB
VC
RC
300k
CC
40pF
CPL
R1
RESR
gm = 40µA/V
7M
ERROR
AMPLIFIER
0.75V
COUT
R2
3988 F05
Figure 5. Model For Loop Response
3988f
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LT3988
Applications Information
finite output impedance. The power section, consisting of
the modulator, power switch and inductor, is modeled as a
transconductance amplifier generating an output current
proportional to the voltage at the VC node.
Note that the output capacitor integrates this current, and
that the capacitor on the VC node (CC) integrates the error amplifier output current, resulting in two poles in the
loop. RC provides a zero. With the recommended output
capacitor, the loop crossover occurs above the RCCC zero.
This simple model works well as long as the value of the
inductor is not too high and the loop crossover frequency
is much lower than the switching frequency. With a larger
ceramic capacitor (very low ESR), crossover may be lower
and a phase lead capacitor (CPL) across the feedback
divider may improve the phase margin and transient
response. Large electrolytic capacitors may have an ESR
large enough to create an additional zero, and the phase
lead may not be necessary. If the output capacitor is different than the recommended capacitor, stability should
be checked across all operating conditions, including
load current, input voltage and temperature. The LT1375
data sheet contains a more thorough discussion of loop
compensation and describes how to test the stability using a transient load.
Shutdown
The EN/UVLO pin is used for two purposes, to place the
LT3988 in a low current shutdown mode, and to override
the internal undervoltage lockout thresholds with a user
programmable threshold. When the EN/UVLO pin is pulled
to under 0.5V (typ), the LT3988 is in shutdown mode and
draws less than 1µA from the input supply. When the
EN/UVLO pin is driven above 0.5V (typ) and less than 1.2V
(typ), the internal regulator is activated and the oscillators
are operating, but the switching operation of both channels remains inhibited. When the EV/UVLO pin is driven
above 1.2V (typ), the undervoltage lockout asserted by the
EN/UVLO function is released, allowing switching operation of both channels. Internal undervoltage detectors will
still prevent switching operation on channel 1 until VIN1 is
greater than 3.9V (typ) and on channel 2 until VIN2 is greater
than 2.6V (typ). The EN/UVLO undervoltage lockout has
120mV (typ) of hysteresis. The EN/UVLO pin is rated up
to 80V and can be connected directly to the input voltage.
The EN/UVLO pin may be driven by a voltage divider from
VIN1, allowing an externally programmable undervoltage
lockout to be set above the internal 3.9V threshold. The
undervoltage threshold and hysteresis are given by:
R1
V
VUVTH = 1.2 1+ ;R1= R2 UVTH – 1
R2
1.2
R1
V
VUVHY = 0.12 1+ ;R1= R2 UVHY – 1
R2
0.12
VIN1
R1
–
EN/UVLO
R2
UVLO
1.2V
+
3988 F06
Figure 6. Undervoltage Lockout Circuit
Output Voltage Tracking
The LT3988 allows the user to program how the output
ramps up by means of the TRACK/SS pins. Through these
pins, either channel output can be set up to either coincidently or ratiometrically track the other channel output.
This example will show the channel 2 output tracking the
channel 1 output, as shown in Figure 7.
The TRACK/SS2 pin acts as a clamp on channel 2’s reference voltage. VOUT2 is referenced to the TRACK/SS2
voltage when the TRACK/SS2 < 0.8V and to the internal
precision reference when TRACK/SS2 > 0.8V. To implement the coincident tracking in Figure 7, connect an extra
resistive divider to the output of channel 1 and connect its
midpoint to the TRACK/SS2 pin (Figure 8).
The ratio of this divider should be selected to be the
same as that of channel 2’s feedback divider (R5 = R3
and R6 = R4). In this tracking mode, VOUT1 must be set
higher than VOUT2. To implement the ratiometric tracking
in Figure 6, change the extra divider ratio to R5 = R1 and
R6 = R2 + ΔR. The extra resistance on R6 should be set
so that the TRACK/SS2 voltage is ≥1V when VOUT1 is at
its final value. The need for this extra resistance is best
understood with the help of the equivalent input circuit
shown in Figure 9.
3988f
15
LT3988
Applications Information
OUTPUT VOLTAGE
VOUT1
I
I
1.36µA
VOUT2
TRACK/SS
0.75V
FB
D1
+
D2
gm
–
D3
3988 F09
Figure 9. Equivalent Input Circuit of Error Amplifier
TIME
Coincident Tracking
OUTPUT VOLTAGE
VOUT1
VOUT2
3988 F07
TIME
Ratiometric Tracking
Figure 7. Two Different Modes of Output Voltage Tracking
VOUT1
TO
TRK/SS2
PIN
VOUT2
R5
R1
R6
R2
TO
FB1
PIN
R3
R4
TO
FB2
PIN
SELECTING VALUES FOR R5 AND R6
COINCIDENT RATIOMETRIC
R5 =
R3
R1
R6 =
R4
R1
VOUT1/1V – 1
3988 F08
R1 VOUT1
R3 VOUT2
=
– 1,
=
–1
R2 0.75
R4 0.75
Figure 8. Setup for Coincident and Ratiometric Tracking
At the input stage of the error amplifier, two common anode
diodes are used to clamp the equivalent reference voltage
and an additional diode is used to match the shifted common mode voltage. The top two current sources are of the
same amplitude. In the coincident mode, the TRACK/SS2
voltage is substantially higher than 0.75V at steady state
and effectively turns off D1. D2 and D3 will therefore conduct the same current and offer tight matching between
VFB2 and the internal precision 0.75V reference. In the
ratiometric mode with R6 = R2, TRACK/SS2 equals 0.75V
at steady state. D1 will divert part of the bias current and
make VFB2 slightly lower than 0.75V. Although this error
is minimized by the exponential I-V characteristic of the
diodes, it does impose a finite amount of output voltage
deviation. Further, when channel 1’s output experiences
dynamic excursions (under load transient, for example),
channel 2 will be affected as well. Setting R6 to a value
that pushes the TRK/SS2 voltage to 1V at steady state will
eliminate these problems while providing near ratiometric
tracking. The example shows channel 2 tracking channel 1,
however either channel may be set up to track the other.
Soft-Start
If a capacitor is tied from the TRACK/SS pin to ground,
then the internal pull-up current will generate a voltage
ramp on this pin. This results in a ramp at the output,
limiting the inductor current and therefore input current
during start-up. A good value for the soft-start capacitor
is COUT/10,000, where COUT is the value of the output
capacitor.
3988f
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LT3988
Applications Information
the input current at VIN2 when VOUT2 is at maximum load.
Figure 10 shows a 12V to 5V, and 1.8V 2-stage converter
using this approach.
Independent Input Voltages
VIN1 and VIN2 are independent and can be powered with
different voltages provided VIN1 is present when VIN2 is
present. Each supply must be bypassed as close to the VIN
pins as possible. For applications requiring large inductors
due to high VIN to VOUT ratios, a 2-stage step-down approach may reduce inductor size by allowing an increase
in frequency. A dual step-down application steps down
the input voltage (VIN1) to the highest output voltage, then
uses that voltage to power the other output (VIN2). VOUT1
must be able to provide enough current for its output plus
VIN
12V
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board (PCB) layout. Figure 11
shows the high current paths in the step-down regulator circuit. Note that in the step-down regulators large,
switched currents flow in the power switch, the catch
diode and the input capacitor. The loop formed by these
VOUT1
4.7µF
4.7µF
VIN1
VIN2
EN/UVLO
TRACK/SS1
TRACK/SS2
2200pF
2200pF
RT
SYNC
LT3988
BOOST1
6.8µH
VOUT1
5V, 500mA
10µF
BOOST2
0.22µF
57.6k
40.2k
BD
0.22µF
SW1
SW2
DA1
FB1
DA2
FB2
GND
10.2k
3.3µH
VOUT2
1.8V, 500mA
14k
10k
22µF
3988 F10
Figure 10. 1MHz, 2-Stage Step-Down 5V and 1.8V Outputs
VIN
VIN
SW
GND
SW
GND
(11a)
VIN
IC1
(11b)
VSW
C1
L1
SW
D1
GND
(11c)
C2
3988 F11
Figure 11. Subtracting the Current When the Switch Is ON (11a) From the Current When the Switch Is OFF (11b) Reveals the Path of
the High Frequency Switching Current (11c). Keep this Loop Small. The Voltage on the SW and Boost Nodes Will Also Be Switched;
Keep These Nodes as Small as Possible. Finally, Make Sure the Circuit Is Shielded with a Local Ground Plane
3988f
17
LT3988
Applications Information
components should be as small as possible. Place these
components, along with the inductor and output capacitor,
on the same side of the circuit board and connect them
on that layer. Place a local, unbroken ground plane below
these components and tie this ground plane to system
ground at one location, ideally at the ground terminal of
the output capacitor. Additionally, keep the SW and BOOST
nodes as small as possible. Figure 12 shows an example
of proper PCB layout.
for the H-grade). The die temperature is calculated by
multiplying the LT3988 power dissipation by the thermal
resistance from junction to ambient. Power dissipation
within the LT3988 can be estimated by calculating the total
power loss from an efficiency measurement and subtracting the catch diode loss. Thermal resistance depends on
the layout of the circuit board, but values from 30°C/W
to 60°C/W are typical.
Related Linear Technology Publications
Thermal Considerations
Application Notes 19, 35, 44, 76 and 88 contain more
detailed descriptions and design information for buck
regulators and other switching regulators. The LT1375
data sheet has a more extensive discussion of output
ripple, loop compensation, and stability testing. Design
Note 318 shows how to generate a dual polarity output
supply using a buck regulator.
The die temperature of the LT3988 must be lower than the
maximum rating of 125°C (150°C for the H-grade). This is
generally not a concern unless the ambient temperature is
above 85°C. For higher temperatures, care should be taken
in the layout of the circuit to ensure good heat sinking of
the LT3988. The maximum load current should be derated
as the ambient temperature approaches 125°C (150°C
C3
R4
R3 R5
R6
R7
C1
C4
C2
U1
D1
D2
C7
C9
L1
C8
L2
C10
3988 F12
Figure 12. Sample PC Board Layout
3988f
18
LT3988
Typical Applications
400kHz, 5V and 3.3V Outputs
VIN
7V TO 40V
80V TRANSIENT
C1
4.7µF
C2
2200pF
VIN1
VIN2
EN/UVLO
TRACK/SS1
TRACK/SS2
RT
SYNC
LT3988
BD
VOUT1
5V, 1A
L1
22µH
R2
57.6k
C6
47µF
C4
0.22µF
BOOST1
BOOST2
SW1
SW2
DA1
FB1
DA2
FB2
D1
R3
10.2k
C3
2200pF
R1
118k
fSW = 400kHz
C5
0.22µF
L2
15µH
D2
GND
VOUT2
3.3V, 1A
R4
34k
R5
10k
C7
47µF
3988 TA02
C1 TO C7: X5R OR X7R
D1, D2: DIODES, INC. B160
3988f
19
LT3988
Typical Applications
1MHz, Wide Input Range 5V and 1.8V Outputs
VIN
7V TO 24V
80V TRANSIENT
VOUT1
C1
4.7µF
C2
2200pF
4.7µF
VIN1
VIN2
EN/UVLO
TRACK/SS1
TRACK/SS2
RT
SYNC
LT3988
BD
VOUT1
5V, 0.5A
L1
6.8µH
R2
57.6k
C6
22µF
C4
0.22µF
BOOST1
BOOST2
SW1
SW2
DA1
FB1
DA2
FB2
D1
R3
10.2k
C3
2200pF
R1
40.2k
fSW = 1MHz
C5
0.22µF
L2
3.3µH
D2
GND
VOUT2
1.8V, 0.5A
R4
14k
R5
10k
C7
22µF
3988 TA03
C1 TO C7: X5R OR X7R
D1: DIODES, INC. B160
D2: DIODES, INC. B120
3988f
20
LT3988
Typical Applications
700kHz, 24V and 12V Outputs with Coincident Tracking
VIN1
26V TO 60V
80V TRANSIENT
C1
4.7µF
C2
4.7µF
VIN2
VIN1
EN/UVLO
TRACK/SS1
C3
2200pF
RT
SYNC
R2
10k
LT3988
TRACK/SS2
R4
4.7k
R3
309k
VOUT1
24V, 1A
L1
47µH
C6
10µF
VIN2
14V TO 60V
C4
0.22µF
BOOST1
BD
BOOST2
SW1
SW2
DA1
FB1
DA2
FB2
D1
R5
309k
R6
10k
R1
61.9k
fSW = 700kHz
C5
0.22µF
D2
GND
L2
22µH
VOUT2
12V, 1A
R7
150k
R8
10k
C7
10µF
3988 TA04
C1 TO C7: X5R OR X7R
D1, D2: DIODES, INC. B160
R4: USE 0.25W RESISTOR
DERATE OUTPUT CURRENT AT HIGHER AMBIENT TEMPERATURES AND INPUT VOLTAGES
TO MAINTAIN JUNCTION TEMPERATURE BELOW THE ABSOLUTE MAXIMUM.
3988f
21
LT3988
Typical Applications
400kHz, 3.3V and 2.5V Outputs
VIN
5.5V TO 32V
80V TRANSIENT
C1
4.7µF
C2
2200pF
VIN1
VIN2
EN/UVLO
TRACK/SS1
TRACK/SS2
RT
SYNC
LT3988
BD
VOUT1
3.3V, 1A
L1
10µH
R2
34k
C6
47µF
C4
0.22µF
BOOST1
BOOST2
SW1
SW2
DA1
FB1
DA2
FB2
D1
R3
10k
C3
2200pF
R1
118k
fSW = 400kHz
C5
0.22µF
D2
GND
VOUT1
L2
10µH
VOUT2
2.5V, 1A
R4
23.2k
R5
10k
C7
47µF
3988 TA05
C1 TO C7: X5R OR X7R
D1, D2: DIODES, INC. B180
3988f
22
LT3988
Package Description
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev E)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ±0.102
(.112 ±.004)
5.23
(.206)
MIN
2.845 ±0.102
(.112 ±.004)
0.889 ±0.127
(.035 ±.005)
8
1
1.651 ±0.102
(.065 ±.004)
1.651 ±0.102 3.20 – 3.45
(.065 ±.004) (.126 – .136)
0.305 ±0.038
(.0120 ±.0015)
TYP
16
0.50
(.0197)
BSC
4.039 ±0.102
(.159 ±.004)
(NOTE 3)
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
9
NO MEASUREMENT PURPOSE
0.280 ±0.076
(.011 ±.003)
REF
16151413121110 9
DETAIL “A”
0° – 6° TYP
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
4.90 ±0.152
(.193 ±.006)
GAUGE PLANE
0.53 ±0.152
(.021 ±.006)
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
1234567 8
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.86
(.034)
REF
0.1016 ±0.0508
(.004 ±.002)
MSOP (MSE16) 0911 REV E
3988f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT3988
Typical Application
500kHz External Sync, 5V and 3.3V Outputs with 6V UVLO
VIN
7V TO 30V
80V TRANSIENT
R6
40.2k
R7
10k
C1
4.7µF
C2
1000pF
VIN1
VIN2
EN/UVLO
TRACK/SS1
TRACK/SS2
RT
LT3988
500kHz
CLOCK
SYNC
C3
1000pF
R1
100k
BD
L1
15µH
VOUT1
5V, 1A
R2
57.6k
C6
22µF
C4
0.22µF
BOOST1
BOOST2
SW1
SW2
DA1
FB1
DA2
FB2
D1
C5
0.22µF
L2
10µH
D2
R3
10.2k
GND
22pF
VOUT2
3.3V, 1A
R4
34k
R5
10k
C7
47µF
3988 TA06
C1 TO C7: X5R OR X7R
D1, D2: DIODES, INC. B180
EN/UVLO THRESHOLD = 6.02V
Related Parts
PART NUMBER
DESCRIPTION
COMMENTS
LT3509
36V with Transient Protection to 60V, Dual 700mA (IOUT), 2.2MHz,
High Efficiency Step-Down DC/DC Converter
VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 1.9mA, ISD = 1µA,
3mm × 4mm DFN-14, MSOP-16E
LT3508
36V with Transient Protection to 40V, Dual 1.4A (IOUT), 2.5MHz, High
Efficiency Step-Down DC/DC Converter
VIN: 3.7V to 36V, VOUT(MIN) = 0.8V, IQ = 4.6mA, ISD = 1µA,
4mm × 4mm QFN-24, TSSOP-16E
LT3980
58V with Transient Protection to 80V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode® Operation
VIN: 3.6V to 58V, Transient to 80V, VOUT(MIN) = 0.79V, IQ = 75µA,
ISD < 1µA, 3mm × 4mm DFN-16, MSOP-16E
LT3970
40V, 350mA (IOUT), 2MHz, High Efficiency Step-Down DC/DC
Converter with Only 2.5µA of Quiescent Current
VIN: 4.2V to 40V, VOUT(MIN) = 1.2V, IQ = 2.5µA, ISD < 1µA,
2mm × 3mm DFN-10, MSOP-10
LT3990
60V, 350mA (IOUT), 2MHz, High Efficiency Step-Down DC/DC
Converter with only 2.5µA of Quiescent Current
VIN: 4.2V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5µA, ISD < 1µA,
3mm × 3mm DFN-10, MSOP-16E
LT3971
38V, 1.2A (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter
with Only 2.8µA of Quiescent Current
VIN: 4.2V to 38V, VOUT(MIN) = 1.2V, IQ = 2.8µA, ISD < 1µA,
3mm × 3mm DFN-10, MSOP-10E
LT3991
55V, 1.2A (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter
with Only 2.8µA of Quiescent Current
VIN: 4.2V to 55V, VOUT(MIN) = 1.2V, IQ = 2.8µA, ISD < 1µA,
3mm × 3mm DFN-10, MSOP-10E
LT3507/LT3507A 36V, Triple 2.4A,1.4A, and 1.4A (IOUT), 2.5MHz, High Efficiency
Step-Down DC/DC Converter with LDO Controller
VIN: 4V to 36V, VOUT(MIN) = 0.8V, IQ = 7mA, ISD = 1µA,
5mm × 7mm QFN-38
LT3680
36V, 3A, 2.4MHz High Efficiency MicroPower Step-Down DC/DC
Converter
VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 75µA, ISD < 1µA,
3mm × 3mm DFN-10, MSOP-10E
LT3693
36V, 3A, 2.4MHz High Efficiency Step-Down DC/DC Converter
VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 1.3mA, ISD < 1µA,
3mm × 3mm DFN-10, MSOP-10E
LT3480
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode Operation
VIN: 3.6V to 38V, Transient to 60V, VOUT(MIN) = 0.78V, IQ = 70µA,
ISD < 1µA, 3mm × 3mm DFN-10, MSOP-10E
3988f
24 Linear Technology Corporation
LT 0412 • PRINTED IN USA
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