LTC1266
LTC1266-3.3/LTC1266-5
Synchronous Regulator
Controller for
N- or P-Channel MOSFETs
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FEATURES
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DESCRIPTIO
Ultrahigh Efficiency: Over 95% Possible
Drives N-Channel MOSFET for High Current or
P-Channel MOSFET for Low Dropout
Pin Selectable Burst Mode Operation
1% Output Accuracy (LTC1266A)
Pin Selectable Phase of Topside Driver for Boost
or Step-Down Operation
Wide VIN Range: 3.5V to 20V
On-Chip Low-Battery Detector
High Efficiency Maintained Over Large Current Range
Low 170µA Standby Current at Light Loads
Current Mode Operation for Excellent Line and Load
Transient Response
Logic Controlled Micropower Shutdown: IQ < 40µA
Short-Circuit Protection
Synchronous Switching with Nonoverlaping Gate Drives
Available in 16-Pin Narrow SO Package
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APPLICATIO S
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Notebook and Palmtop Computers
Portable Instruments
Cellular Telephones
DC Power Distribution Systems
GPS Systems
The operating current level is user-programmable via an
external current sense resistor. Wide input supply range
allows operation from 3.5V to 18V (20V maximum).
Constant off-time architecture provides low dropout regulation limited only by the RDS(ON) of the topside MOSFET
(when using the P-channel) and the resistance of the
inductor and current sense resistor.
The LTC1266 series combines synchronous switching for
maximum efficiency at high currents with an automatic
low current operating mode, called Burst Mode operation,
which reduces switching losses. Standby power is reduced to only 1mW at VIN = 5V (at IOUT = 0). Load currents
in Burst Mode operation are typically 0mA to 500mA.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
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The LTC®1266 series is a family of synchronous switching
regulator controllers featuring automatic Burst ModeTM
operation to maintain high efficiencies at low output
currents. These devices drive external power MOSFETs at
switching frequencies up to 400kHz using a constant offtime current mode architecture providing constant ripple
current in the inductor. They can drive either an N-channel
or a P-channel topside MOSFET.
TYPICAL APPLICATIO
D2
MBR0530T1
+
CIN
100µF
×2
LOW BAT IN
LTC1266-3.3 Efficiency
VIN
LBIN
PINV
PWR VIN
LBOUT
CB
0.1µF
100k
L*
5µH
RSENSE
0.02Ω
RC
470Ω
CC
3300pF
CT
180pF
SHDN
ITH
CT
BINH
SGND
SENSE
+
SENSE –
BDRIVE
PGND
VIN = 5V
95
N-CHANNEL
Si9410
TDRIVE
LTC1266-3.3
0V = NORMAL
>1.5V = SHUTDOWN
100
LOW BAT OUT
VOUT
3.3V
5A
1000pF
N-CHANNEL
Si9410
D1
MBRS130LT3
+
COUT
330µF
×2
EFFICIENCY (%)
VIN
4V TO 9V
90
85
1266 TA01
80
0.01
*TOKO 919AS-4R7M
Figure 1. High Efficiency Step-Down Converter
0.1
1
LOAD CURRENT (A)
5
1266 TA02
1
LTC1266
LTC1266-3.3/LTC1266-5
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AXI U
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ABSOLUTE
RATI GS
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PACKAGE/ORDER I FOR ATIO
(Note 1)
Input Supply Voltage (Pins 2, 5) ............... 20V to – 0.3V
Continuous Output Current (Pins 1, 16) .............. 50mA
Sense Voltages (Pins 8, 9) ....................... 13V to – 0.3V
SHDN Voltage (Pin 11) ............................. 12V to – 0.3V
PINV, BINH, LBIN (Pins 3, 4, 13)................20V to – 0.3V
LBOUT Output Current ........................................... 12mA
Operating Ambient Temperature Range ...... 0°C to 70°C
Industrial Temperature Range ................ – 40°C to 85°C
Extended Commercial
Temperature Range ........................... – 40°C to 85°C
Junction Temperature (Note 2) ............................ 125°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
ORDER PART
NUMBER
TOP VIEW
TDRIVE 1
16 BDRIVE
PWR VIN 2
15 PGND
PINV 3
14 LBOUT
BINH 4
13 LBIN
VIN 5
12 SGND
CT 6
11 SHDN
ITH 7
10 VFB (NC*)
SENSE – 8
9 SENSE +
S PACKAGE
16-LEAD PLASTIC SO
LTC1266CS
LTC1266CS-3.3
LTC1266CS-5
LTC1266ACS
LTC1266IS
LTC1266IS-3.3
LTC1266IS-5
LTC1266AIS
*FIXED OUTPUT VERSIONS
TJMAX = 125°C, θJA = 110°C/ W
Consult factory for Industrial and Military grade parts.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VSHDN = VBINH = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VFB
Feedback Voltage
VIN = 9V, ILOAD = 700mA, VPINV = VPWR = 14V,
Topside Switch = N-Ch
LTC1266ACS
LTC1266CS
IFB
Feedback Current (LTC1266 Only)
VOUT
Regulated Output Voltage
●
●
IQ1
IQ2
2
TYP
MAX
UNITS
1.252
1.210
1.265
1.265
1.278
1.290
V
V
0.2
1
µA
3.33
5.05
3.43
5.20
V
V
●
VIN = 9V, ILOAD = 700mA, VPINV = VPWR = 14V,
Topside Switch = N-Ch
LTC1266CS-3.3
LTC1266CS-5
∆VOUT
MIN
●
●
Output Ripple (Burst Mode Operation) ILOAD = 150mA
Output Voltage Line Regulation
ILOAD = 50mA
VPINV = 0V, Topside Switch = P-Ch, VIN = 7V to 12V
VPINV = VPWR, Topside Switch = N-Ch, VIN = 7V to 12V
3.23
4.90
50
– 40
– 40
mVP-P
0
0
40
40
mV
mV
40
15
60
25
65
25
100
40
mV
mV
mV
mV
Output Voltage Load Regulation
LTC1266-3.3
LTC1266-3.3
LTC1266-5
LTC1266-5
5mA < ILOAD < 2A, RSENSE = 0.05Ω
Burst Mode Operation Enabled, VBINH = 0V
Burst Mode Operation Inhibited, VBINH = 2V
Burst Mode Operation Enabled, VBINH = 0V
Burst Mode Operation Inhibited, VBINH = 2V
VIN Pin DC Supply Current (Note 3)
Normal Mode
Sleep Mode
Shutdown
3.5V < VIN < 18V
3.5V < VIN < 18V
VSHDN = 2.1V, 3.5V < VIN < 18V
2.1
170
25
3.0
250
50
mA
µA
µA
PWR VIN DC Supply Current (Note 3)
Normal Mode
Sleep Mode
Shutdown
3.5V < PWR VIN < 18V
3.5V < PWR VIN < 18V
VSHDN = 2.1V, 3.5V < PWR VIN < 18V
20
1
1
40
5
5
µA
µA
µA
●
●
●
●
LTC1266
LTC1266-3.3/LTC1266-5
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VSHDN = VBINH = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VSENSE 1
Current Sense Threshold
(Burst Mode Operation Enabled)
LTC1266
VBINH = 0V
LTC1266-3.3
LTC1266-5
VSENSE 2
Current Sense Threshold
(Burst Mode Operation Disabled)
LTC1266
LTC1266-3.3
LTC1266-5
VSENSE – = 3.3V, VFB = VOUT/2.64 + 25mV (Forced)
VSENSE – = 3.3V, VFB = VOUT/2.64 – 25mV (Forced)
VSENSE – = VOUT + 100mV (Forced)
VSENSE – = VOUT – 100mV (Forced)
VSENSE – = VOUT + 100mV (Forced)
VSENSE – = VOUT – 100mV (Forced)
MIN
TYP
25
155
25
155
25
155
MAX
●
135
175
●
135
●
135
●
135
●
135
●
135
– 20
155
– 20
155
– 20
155
0.6
0.8
2
175
175
UNITS
mV
mV
mV
mV
mV
mV
VBINH = 2.1V
VSENSE – = 3.3V, VFB = VOUT/2.64 + 25mV (Forced)
VSENSE – = 3.3V, VFB = VOUT/2.64 – 25mV (Forced)
VSENSE – = VOUT + 100mV (Forced)
VSENSE – = VOUT – 100mV (Forced)
VSENSE – = VOUT + 100mV (Forced)
VSENSE – = VOUT – 100mV (Forced)
175
175
175
mV
mV
mV
mV
mV
mV
VSHDN
Shutdown Pin Threshold
ISHDN
Shutdown Pin Input Current
0V < VSHDN < 8V, VIN = 16V
1.2
5
µA
IPINV
Phase Invert Pin Input Current
0V < VPINV < 18V, VIN = 18V
0.2
1
µA
VBINH
Burst Mode Operation
Inhibit Pin Threshold
VIN = 7V
1.2
2
V
IBINH
Burst Mode Operation
Inhibit Pin Input Current
0V < VBINH < 18V, VIN = 18V
0.2
1
µA
ICT
CT Pin Discharge Current
VSENSE + = VOUT – 100mV, VSENSE – = VOUT – 300mV
VOUT = 0V
50
70
2
90
10
µA
µA
tOFF
Off-Time (Note 4)
CT = 390pF, ILOAD = 700mA
4
5
6
µs
tMAX
Max On-Time
VOUT = 0V, VIN = 18V
60
tr, tf
Driver Output Transition Times (Note 7)
CL = 3000pF (Pins 1, 16), VIN = 6V
100
200
ns
VCLAMP
VBINH = 2.1V
VLBTRIP
Output Voltage Clamp in
Burst Mode Operation Inhibit
LTC1266
LTC1266-3.3
LTC1266-5
Low-Battery Trip Point
ILBLEAK
Max Leakage Current Into Pin 14
VLBOUT = 18V, VLBIN = 2V
ILBSINK
Max Sink Current Into Pin 14
VLBOUT = 1V, VLBIN = 0V, 2.5V < VIN < 18V
ILBIN
Max Leakage Current Into Pin 13
VLBIN = 18V
Measured at VFB
Measured at VSENSE –
Measured at VSENSE –
VIN = 5V
VIN = 12V
0.8
µs
1.30
3.43
5.20
1.14
1.17
1
V
V
V
V
1.25
1.30
1.35
1.42
V
V
25
200
nA
8
mA
µA
0.2
1
MIN
TYP
MAX
UNITS
1.246
1.210
1.265
1.265
1.290
1.290
V
V
3.23
4.90
3.33
5.05
3.43
5.20
V
V
– 40°C < TA < 85°C (Note 5), VIN = 10V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VFB
Feedback Voltage
VIN = 9V, ILOAD = 700mA, VPINV = VPWR = 14V,
Topside Switch = N-Ch
LTC1266AIS
LTC1266CS, LTC1266IS
VOUT
Regulated Output Voltage
LTC1266CS-3.3, LTC1266IS-3.3
LTC1266CS-5, LTC1266IS-5
VIN = 9V, ILOAD = 700mA, VPINV = VPWR = 14,
Topside Switch = N-Ch
3
LTC1266
LTC1266-3.3/LTC1266-5
ELECTRICAL CHARACTERISTICS
SYMBOL
PARAMETER
CONDITIONS
IQ1
VIN Pin DC Supply Current (Note 3)
Normal Mode
Sleep Mode
Shutdown
PWR VIN DC Supply Current (Note 3)
Normal Mode
Sleep Mode
Shutdown
IQ2
VSENSE1
VSENSE2
Current Sense Threshold
(Burst Mode Operation Enabled)
LTC1266CS, LTC1266IS
LTC1266CS
LTC1266IS
LTC1266-3.3, LTC1266-5 (I and C)
LTC1266CS-3.3, LTC1266CS-5
LTC1266IS-3.3, LTC1266IS-5
Current Sense Threshold
(Burst Mode Operation Disabled)
LTC1266CS, LTC1266IS
LTC1266CS
LTC1266IS
LTC1266-3.3, LTC1266-5 (I and C)
LTC1266CS-3.3, LTC1266CS-5
LTC1266IS-3.3, LTC1266IS-5
– 40°C < TA < 85°C (Note 5), VIN = 10V, unless otherwise noted.
MIN
TYP
MAX
UNITS
3.5V < VIN < 18V
3.5V < VIN < 18V
VSHUTDOWN = 2.1V, 3.5V < VIN < 18V
2.1
170
25
3.3
260
60
mA
µA
µA
3.5V < PWR VIN < 18V
3.5V < PWR VIN < 18V
VSHUTDOWN = 2.1V, 3.5V < PWR VIN < 18V
20
1
1
50
7
7
µA
µA
µA
VBINH = 0V
VSENSE– = 3.3V, VFB = VOUT/2.64 + 25mV (Forced)
VSENSE– = 3.3V, VFB = VOUT/2.64 – 25mV (Forced)
VSENSE– = 3.3V, VFB = VOUT/2.64 – 25mV (Forced)
VSENSE– = VOUT + 100mV (Forced)
VSENSE– = VOUT – 100mV (Forced)
VSENSE– = VOUT – 100mV (Forced)
135
135
135
135
25
155
155
25
155
155
180
190
180
190
mV
mV
mV
mV
mV
mV
VBINH = 2.1V
VSENSE– 3.3V, VFB = VOUT/2.64 + 25mV (Forced)
VSENSE– 3.3V, VFB = VOUT/2.64 – 25mV (Forced)
VSENSE– 3.3V, VFB = VOUT/2.64 – 25mV (Forced)
VSENSE– = VOUT + 100mV (Forced)
VSENSE– = VOUT – 100mV (Forced)
VSENSE– = VOUT – 100mV (Forced)
130
130
–20
155
155
–20
155
155
185
195
130
130
185
195
mV
mV
mV
mV
mV
mV
VSHDN
Shutdown Pin Threshold
C Grade
I Grade
0.55
0.50
0.8
0.8
2
2
V
V
tOFF
Off-Time (Note 4)
CT = 390pF, ILOAD = 700mA, C Grade
CT = 390pF, ILOAD = 700mA, I Grade
3.8
3.8
5
5
6.5
7.0
µs
µs
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD × 110°C/W)
Note 3: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 4: In applications where RSENSE is placed at ground potential, the offtime increases approximately 40%.
4
Note 5: The LTC1266CS, LTC1266CS-3.3, LTC1266-5 and LTC1266ACS
are not tested and not quality assurance sampled at – 40°C and 85°C.
These specifications are guaranteed by design and/or correlation. The
LTC1266IS, LTC1266IS-3.3, LTC1266IS-5 and LTC1266AIS are
guaranteed and tested over the – 40°C to 85°C operating temperature
range.
Note 6: Unless otherwise noted the specifications for the LTC1266A are
the same as those for the LTC1266.
Note 7: tr and tf are measured at 10% and 90% levels.
LTC1266
LTC1266-3.3/LTC1266-5
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TYPICAL PERFOR A CE CHARACTERISTICS
Line Regulation
Efficiency vs Input Voltage
100
FIGURE 1 CIRCUIT
ILOAD = 2.5A
ILOAD = 5A
85
ILOAD = 100mA
80
10
0
–10
75
8
5
6
7
INPUT VOLTAGE (V)
4
– 40
9
4
5
6
7
INPUT VOLTAGE (V)
ILOAD = 1A
0
–10
16
0
20
4
8
12
VIN = 5V (Burst Mode
OPERATION INHIBITED)
0
2.0
1.5
1.0
LOAD CURRENT (A)
0.5
1.5
1.0
Supply Current in Shutdown
25
50
20
40
SUPPLY CURRENT (µA)
SUPPLY CURRENT (µA)
2.5
ACTIVE MODE
15
10
5
VIN
30
20
10
SLEEP MODE
PWR VIN
SLEEP MODE
0
0
0
4
12
8
INPUT VOLTAGE (V)
16
20
1266 G07
3.0
1266 G06
Power VIN DC Supply Current
ACTIVE MODE
2.5
1266 G05
VIN DC Supply Current
SUPPLY CURRENT (mA)
VIN = 5V
INPUT VOLTAGE (V)
3.0
0.5
–10
16
1266 G04
2.0
VIN = 12V (Burst Mode
OPERATION ENABLED)
–40
– 40
12
8
INPUT VOLTAGE (V)
0
–30
–30
4
FIGURE 11 CIRCUIT
–20
–20
75
5
10
10
∆VOUT (mV)
∆VOUT (mV)
EFFICIENCY (%)
80
0
4
20
20
ILOAD = 2.5A
70
3
2
LOAD CURRENT (A)
Load Regulation
FIGURE 11 CIRCUIT
ILOAD = 1A
30
ILOAD = 100mA
1
1266 G03
30
40
FIGURE 11 CIRCUIT
85
0
9
8
Line Regulation
90
VIN = 5V (Burst Mode
OPERATION INHIBITED)
1266 G02
Efficiency vs Input Voltage
95
–20
–50
3
1266 G01
100
VIN = 5V
–40
–30
3
–10
–30
–20
70
VIN = 9V (Burst Mode
OPERATION ENABLED)
0
∆VOUT (mV)
90
FIGURE 1 CIRCUIT
10
20
∆VOUT (mV)
EFFICIENCY (%)
FIGURE 1 CIRCUIT
ILOAD = 1A
30
95
Load Regulation
20
40
0
4
12
8
INPUT VOLTAGE (V)
16
20
1266 G08
0
0
5
15
10
INPUT VOLTAGE (V)
20
1266 G09
5
LTC1266
LTC1266-3.3/LTC1266-5
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TYPICAL PERFOR A CE CHARACTERISTICS
Operating Frequency
vs (VIN – VOUT) Voltage
100
3.0
VSENSE– = VOUT
NORMALIZED FREQUENCY
OFF-TIME (µs)
VOUT = 3.3V
2.5
80
60
40
20
Current Sense Threshold Voltage
200
MAX THRESHOLD
0°C
150
SENSE VOLTAGE (mV)
Off-Time vs Output Voltage
70°C
2.0
25°C
1.5
1.0
MIN THRESHOLD (Burst Mode
OPERATION ENABLED)
50
MIN THRESHOLD (Burst Mode
OPERATION INHIBIT)
0
0.5
LTC1266-5
100
LTC1266-3.3
0
0
1
3
4
2
OUTPUT VOLTAGE (V)
5
0
0
2
6
8
10 12
4
(VIN – VOUT) VOLTAGE (V)
1266 G10
–50
14
16
1266 G11
0
20
60
40
TEMPERATURE (°C)
80
100
1266 G12
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PI FU CTIO S
TDRIVE (Pin 1): High Current Drive for Topside MOSFET.
This MOSFET can be either P-channel or N-channel, user
selectable by Pin 3. Voltage swing at this pin is from PWR
VIN to ground.
PWR VIN (Pin 2): Power Suppy for Drive Signals. Must be
closely decoupled to power ground (Pin 15).
PINV (Pin 3): Phase Invert. Sets the phase of the topside
driver to drive either a P-channel or an N-channel MOSFET
as follows:
P-channel: Pin 3 = 0V
N-channel: Pin 3 = PWR VIN
BINH (Pin 4): Burst Mode Operation Inhibit. A CMOS logic
high on this pin will disable the Burst Mode operation
feature forcing continuous operation down to zero load.
VIN (Pin 5): Main Supply Pin.
CT (Pin 6): External Capacitor. CT from Pin 4 to ground sets
the operating frequency. The actual frequency is also
dependent on the input voltage.
ITH (Pin 7): Gain Amplifier Decoupling Point. The current
comparator threshold increases with the Pin 7 voltage.
SENSE – (Pin 8): Connects to internal resistive divider
which sets the output voltage in LTC1266-3.3 and
LTC1266-5 versions. Pin 8 is also the (–) input for the
current comparator.
6
SENSE + (Pin 9): The (+) Input to the Current Comparator.
A built-in offset between Pins 8 and 9 in conjunction with
RSENSE sets the current trip threshold.
VFB (Pin 10): For the LTC1266 adjustable version, Pin 10
serves as the feedback pin from an external resistive
divider used to set the output voltage. On LTC1266-3.3
and LTC1266-5 versions this pin is not used.
SHDN (Pin 11): When grounded, the LTC1266 series
operates normally. Pulling Pin 11 high holds both
MOSFETs off and puts the LTC1266 in micropower shutdown mode. Requires CMOS logic signal with tr, tf < 1µs.
Should not be left floating.
SGND (Pin 12): Small-Signal Ground. Must be routed
separately from other grounds to the (–) terminal of COUT.
LBIN (Pin 13): Input to the Low-Battery Comparator. This
input is compared to an internal 1.25V reference.
LBOUT (Pin 14): Open Drain Output of the Low-Battery
Comparator. This pin will sink current when Pin 13 is
below 1.25V.
PGND (Pin 15): Driver Power Ground. Connects to source
of N-channel MOSFET and the (–) terminal of CIN.
BDRIVE (Pin 16): High Current Drive for Bottom
N-Channel MOSFET. Voltage swing at Pin 16 is from
ground to PWR VIN.
LTC1266
LTC1266-3.3/LTC1266-5
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FU CTIO AL DIAGRA
Pin 10 Connection Shown for LTC1266-3.3 and LTC1266-5; Changes Create LTC1266
–
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LBIN 13
+
1.25V
REFERENCE
VIN
PWR VIN
14 LBOUT
LB
2
3 PINV
1 TDRIVE
SIGNAL
GROUND
12
SENSE+
SENSE –
9
8
ADJUSTABLE
VERSION
VFB
16 BDRIVE
BINH
10
4
–
15 PGND
V
+
SLEEP
–
C
R
+
Q
+
S
S
VTH1
13k
+
6
CT
5pF
VOS
–
ITH 7
T
OFF-TIME
CONTROL
VIN
SENSE –
VFB
G
MAX
ON-TIME
CONTROL
ENABLE
100k
+
VTH2
+ VTRIP
–
–
–
PINV
SHDN 11
1.265V
REFERENCE
5 VIN
1266 FD
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OPERATIO
The LTC1266 series uses a current mode, constant offtime architecture to synchronously switch an external pair
of power MOSFETs. Operating frequency is set by an
external capacitor at the timing capacitor Pin 6.
The output voltage is sensed by an internal voltage divider
connected to SENSE –, Pin 8, (LTC1266-3.3 and LTC12665) or external divider returned to VFB, Pin 10, (LTC1266).
A voltage comparator V, and a gain block G, compare the
divided output voltage with a reference voltage of 1.265V.
To optimize efficiency, the LTC1266 automatically switches
between two modes of operation, burst and continuous.
The voltage comparator is the primary control element
when the device is in Burst Mode operation, while the gain
block controls the output voltage in continuous mode.
During the switch ON cycle in continuous mode, current
comparator C monitors the voltage between Pins 8 and 9
connected across an external shunt in series with the
inductor. When the voltage across the shunt reaches its
threshold value, the topside driver output is switched to
turn off the topside MOFSET (Power VIN for P-channel or
ground for N-channel). The timing capacitor connected to
Pin 6 is now allowed to discharge at a rate determined by
the off-time controller. The discharge current is made
proportional to the output voltage (measured by Pin 8) to
model the inductor current, which decays at a rate which
is also proportional to the output voltage. While the timing
capacitor is discharging, the bottom-side drive output is
switched to power VIN to turn on the bottom-side
N-channel MOSFET.
7
LTC1266
LTC1266-3.3/LTC1266-5
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OPERATIO
When the voltage on the timing capacitor has discharged
past VTH1, comparator T trips, setting the flip-flop. This
causes the bottom-side output to switch off and the
topside output to switch on (ground for P-channel and
Power VIN for N-channel). The cycle then repeats.
As the load current increases, the output voltage decreases
slightly. This causes the output of the gain stage (Pin 7) to
increase the current comparator threshold, thus tracking
the load current.
The sequence of events for Burst Mode operation is very
similar to continuous operation with the cycle interrupted
by the voltage comparator. When the output voltage is at
or above the desired regulated value, the topside MOSFET
is held off by comparator V and the timing capacitor
continues to discharge below VTH1. When the timing
capacitor discharges past VTH2, voltage comparator S
trips, causing the internal sleep line to go low and the
bottom-side MOSFET to turn off.
The circuit now enters sleep mode with both power
MOSFETs turned off. In sleep mode, a majority of the
circuitry is turned off, dropping the quiescent current
from 2.1mA to 170µA. The load current is now being
supplied from the output capacitor. When the output
voltage has dropped by the amount of hysteresis in
comparator V, the topside MOSFET is again turned on
and this process repeats.
To avoid the operation of the current loop interfering with
Burst Mode operation, a built-in offset VOS is incorporated
in the gain stage. This prevents the current comparator
threshold from increasing until the output voltage has
dropped below a minimum threshold.
To prevent both the external MOSFETs from ever being
turned on at the same time, feedback is incorporated to
sense the state of the driver output pins. Before the
bottom-side drive output can turn on, the topside output
must be off. Likewise, the topside output is prevented
from turning on while the bottom-side drive output is
still on.
The LTC1266 has two select pins which provide the user
with choice of topside switch and with the option of
inhibiting Burst Mode operation. The phase select pin
allows the user to choose whether the topside MOSFET
is a P-channel or an N-channel. The phase select pin does
two things: sets the proper phase of the drive signal (ON
= Power VIN for N-channel and ON = 0V for P-channel)
and also sets an upper limit for the on-time (60µs) when
set to the N-channel. The on-time limit ensures proper
start-up when used in a single supply bootstrap circuit
configuration (see Applications Information). In P-channel
mode there is no on-time limit and thus, in dropout, the
P-channel MOSFET is turned on continuously (100%
duty cycle).
The Burst Mode operation inhibit (BINH, Pin 4) allows the
Burst Mode operation to be disabled by applying a CMOS
logic high to this pin. With Burst Mode operation disabled,
the LTC1266 will remain in continuous mode down to zero
load. Burst Mode operation is disabled by allowing the
lower current threshold limit to go below zero so that the
voltage comparator will never trip. The voltage comparator
trip point is also raised up so that it will not be tripped by
transients. It is still active to provide a voltage clamp to
prevent the output from overshooting.
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One of the three basic LTC1266 application circuits is
shown in Figure 1. This circuit uses an N-channel
topside driver and a single supply. The other two circuit
configurations (see Typical Applications) use an
N-channel topside driver and dual supply, and a
P-channel topside driver. Selections of other external
components are driven by the load requirement and are
the same for all three circuit configurations. The first
8
step is the selection of RSENSE. Once RSENSE is known,
CT and L can be chosen. Next, the power MOSFETs and
D1 are selected. Finally, CIN and COUT are selected and
the loop is compensated. Using an N-channel topside
switch, input voltages are limited to a maximum of
about 15V. With a P-channel, the input voltage may be
as high as 20V.
LTC1266
LTC1266-3.3/LTC1266-5
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RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
The LTC1266 series current comparator has a threshold
range which extends from a minimum of 25mV/RSENSE
(when Burst Mode operation is enabled) to a maximum of
155mV/RSENSE. The current comparator threshold sets
the peak of the inductor ripple current, yielding a maximum output current IMAX equal to the peak value less half
the peak-to-peak ripple current. For proper Burst Mode
operation, IRIPPLE(P-P) must be less than or equal to the
minimum current comparator threshold.
Since efficiency generally increases with ripple current,
the maximum allowable ripple current is assumed, i.e.,
IRIPPLE(P-P) = 25mV/RSENSE (see CT and L Selection for
Operating Frequency). Solving for RSENSE and allowing
a margin for variations in the LTC1266 series and
external component values yields:
RSENSE = 100mV
IMAX
A graph for selecting RSENSE vs maximum output
current is given in Figure 2.
100
ISC(PK) = 155mV
RSENSE
The LTC1266 series automatically extends tOFF during a
short circuit to allow sufficient time for the inductor
current to decay between switch cycles. The resulting
ripple current causes the average short-circuit current
ISC(AVG) to be reduced to approximately IMAX.
L and CT Selection for Operating Frequency
The LTC1266 series uses a constant off-time architecture
with tOFF determined by an external timing capacitor CT.
Each time the topside MOSFET switch turns on, the
voltage on CT is reset to approximately 3.3V. During the
off-time, CT is discharged by a current which is proportional to VOUT. The voltage on CT is analogous to the
current in inductor L, which likewise decays at a rate
proportional to VOUT. Thus the inductor value must track
the timing capacitor value.
The value of CT is calculated from the desired continuous
mode operating frequency, f:
1
2.6 • 104 • f
assumes VIN = 2VOUT, (Figure 1 circuit).
CT =
75
RSENSE (mΩ)
IBURST ≈ 15mV
RSENSE
50
A graph for selecting CT vs frequency including the effects
of input voltage is given in Figure 3.
25
800
0
4
2
8
6
MAXIMUM OUTPUT CURRENT (A)
10
1266 F02
Figure 2. Selecting RSENSE
The load current, below which Burst Mode operation
commences, (IBURST), and the peak short-circuit current, (ISC(PK)), both track IMAX. Once RSENSE has been
chosen, IBURST and ISC(PK) can be predicted from the
following:
600
CAPACITANCE (pF)
0
VOUT = 3.3V
400
VIN = 12V
200
VIN = 5V
0
0
100
200
300
FREQUENCY (kHz)
400
500
1266 F03
Figure 3. Timing Capacitor Value
9
LTC1266
LTC1266-3.3/LTC1266-5
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As the operating frequency is increased the gate charge
losses will be higher, reducing efficiency (see Efficiency
Considerations). The complete expression for operating
frequency of the circuit in Figure 1 is given by:
f=
1
tOFF
)
1–
VOUT
VIN
)
where:
tOFF = 1.3 • 104 • CT •
) )
VREG
VOUT
VREG is the desired output voltage (i.e., 5V, 3.3V). VOUT is the
measured output voltage. Thus VREG/VOUT = 1 in regulation.
Once the frequency has been set by CT, the inductor L
must be chosen to provide no more than 25mV/RSENSE
of peak-to-peak inductor ripple current. This results in
a minimum required inductor value of:
LMIN = 5.1 • 105 • RSENSE • CT • VREG
As the inductor value is increased from the minimum
value, the ESR requirements for the output capacitor
are eased at the expense of efficiency. If too small an
inductor is used, the inductor current will decrease past
zero and change polarity. A consequence of this is that
the LTC1266 series may not enter Burst Mode operation
and efficiency will be slightly degraded at low currents.
Inductor Core Selection
Once the minimum value for L is known, the type of
inductor must be selected. The highest efficiency will be
obtained using ferrite, Kool Mµ® on molypermalloy (MPP)
cores. Lower cost powdered iron cores provide suitable
performance but cut efficiency by 3% to 7%. Actual core
loss is independent of core size for a fixed inductor value,
but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately,
increased inductance requires more turns of wire and
therefore copper losses increase.
Ferrite designs have very low core loss, so design goals
can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means
that inductance collapses abruptly when the peak design
Kool Mµ is a registered trademark of Magnetics, Inc.
10
current is exceeded. This results in an abrupt increase in
inductor ripple current and consequent output voltage
ripple which can cause Burst Mode operation to be falsely
triggered. Do not allow the core to saturate!
Kool Mµ is a very good, low loss core material for toroids,
with a “soft” saturation characteristic. Molypermalloy is
slightly more efficient at high (> 200kHz) switching frequency. Toroids are very space efficient, especially when
you can use several layers of wire. Because they generally
lack a bobbin, mounting is more difficult. However, new
designs for surface mount are available from Coiltronics
and Beckman Industrial Corp. which do not increase the
height significantly.
Power MOSFET and D1 Selection
Two external power MOSFETs must be selected for use
with the LTC1266 series: either a P-channel MOSFET or an
N-channel MOSFET for the main switch and an N-channel
MOSFET for the synchronous switch. The main selection
criteria for the power MOSFETs are the type of MOSFET,
threshold voltage VGS(TH) and on-resistance RDS(ON).
The cost and maximum output current determine the type
of MOSFET for the topside switch. N-channel MOSFETs
have the advantage of lower cost and lower RDS(ON) at the
expense of slightly increased circuit complexity. For lower
current applications where the losses due to RDS(ON) are
small, a P-channel MOSFET is recommended due to the
lower circuit complexity. However, at load currents in
excess of 3A where the RDS(ON) becomes a significant
portion of the total power loss, an N-channel is strongly
recommended to maximize efficiency.
The maximum output current IMAX determines the RDS(ON)
requirement for the two MOSFETs. When the LTC1266
series is operating in continuous mode, the simplifying
assumption can be made that one of the two MOSFETs is
always conducting the average load current. The duty
cycles for the two MOSFETs are given by:
Topside Duty Cycle =
VOUT
VIN
Bottom-Side Duty Cycle =
VIN – VOUT
VIN
LTC1266
LTC1266-3.3/LTC1266-5
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From the duty cycles, the required RDS(ON) for each
MOSFET can be derived:
TS RDS(ON) =
VIN • PT
VOUT • IMAX2 • (1 + δT)
BS RDS(ON) =
VIN • PB
(VIN – VOUT) • IMAX2 • (1 + δB)
where PT and PB are the allowable power dissipations and
δT and δB are the temperature dependencies of RDS(ON). PT
and PB will be determined by efficiency and/or thermal
requirements (see Efficiency Considerations). For a MOSFET,
(1 + δ) is generally given in the form of a normalized
RDS(ON) vs temperature curve, but δPCH = 0.007/°C and
δNCH = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
The minimum input voltage determines whether standard
threshold or logic-level threshold MOSFETs must be used.
For VIN > 8V, standard threshold MOSFETs (VGS(TH) < 4V)
may be used. If VIN is expected to drop below 8V, logiclevel threshold MOSFETs (VGS(TH) < 2.5V) are strongly
recommended. The LTC1266 series Power VIN must always be less than the absolute maximum VGS ratings for
the MOSFETs.
The Schottky diode D1 shown in Figure 1 only conducts
during the deadtime between the conduction of the two
power MOSFETs. D1’s sole purpose in life is to prevent the
body diode of the bottom-side MOSFET from turning on
and storing charge during the deadtime, which could cost
as much as 1% in efficiency (although there are no other
harmful effects if D1 is omitted). Therefore, D1 should be
selected for a forward voltage of less than 0.7V when
conducting IMAX.
CIN and COUT Selection
In continuous mode, the current through the topside
MOSFET is a square wave of duty cycle VOUT/VIN. To
prevent large voltage transients, a low ESR (Effective
Series Resistance) input capacitor sized for the maximum
RMS current must be used. The maximum RMS capacitor
current is given by:
CIN Required IRMS ≈ IMAX
[VOUT(VIN – VOUT)]1/2
VIN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor
manufacturer’s ripple current ratings are often based on
only 2000 hours of life. This makes it advisable to further
derate the capacitor, or to choose a capacitor rated at a
higher temperature than required. Always consult the
manufacturer if there is any question. An additional 0.1µF
to 1µF ceramic capacitor is also required on Power VIN
(Pin 2) for high frequency decoupling.
The selection of COUT is driven by the required ESR. The
ESR of COUT must be less than twice the value of RSENSE
for proper operation of the LTC1266 series:
COUT Required ESR < 2RSENSE
Optimum efficiency is obtained by making the ESR equal
to RSENSE. As the ESR is increased up to 2RSENSE, the
efficiency degrades by less than 1%. If the ESR is greater
than 2RSENSE, the voltage ripple on the output capacitor
will prematurely trigger Burst Mode operation, resulting in
disruption of continuous mode and an efficiency hit which
can be several percent. If Burst Mode operation is disabled, the ESR requirement can be relaxed and is limited
only by the allowable output voltage ripple.
Manufacturers such as Nichicon and United Chemicon
should be considered for high performance capacitors.
The OS-CON semiconductor dielectric capacitor available
from Sanyo has the lowest ESR/size ratio of any aluminum
electrolytic at a somewhat higher price. Once the ESR
requirement for COUT has been met, the RMS current
rating generally far exceeds the IRIPPLE(P-P) requirement.
In surface mount applications multiple capacitors may
have to be paralleled to meet the capacitance, ESR or RMS
current handling requirements of the application. An
excellent choice is the AVX TPS series of surface mount
tantalums.
At low supply voltages, a minimum capacitance at COUT
is needed to prevent an abnormal low frequency operating mode (see Figure 4). When COUT is made too
small, the output ripple at low frequencies will be large
enough to trip the voltage comparator. This causes
Burst Mode operation to be activated when the LTC1266
11
LTC1266
LTC1266-3.3/LTC1266-5
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Driving N-Channel Topside MOSFETs
1000
L = 50µH
RSENSE = 0.02Ω
COUT (µF)
800
L = 25µH
RSENSE = 0.02Ω
600
400
200
0
L = 50µH
RSENSE = 0.05Ω
0
1
3
4
2
(VIN – VOUT) VOLTAGE (V)
5
1266 F04
Figure 4. Minimum Value of COUT
series would normally be in continuous operation. The
output remains in regulation at all times. This minimum
capacitance requirement may be relaxed if Burst Mode
operation is disabled.
N-Channel vs P-Channel MOSFETs
The LTC1266 has the capability to drive either an
N-channel or a P-channel topside switch to give the user
more flexibility. N-channel MOSFETs are superior in performance to P-channel due to their lower RDS(ON) and
lower gate capacitance and are typically less expensive;
however, they do have a slightly more complicated gate
drive requirement and a more limited input voltage range
(see following sections).
Driving P-Channel Topside MOSFETs
The P-channel topside switch circuit configuration is the
most straightforward due to the requirement of only one
supply voltage level. This is due to the negative gate
threshold of the P-channel MOSFET which allows the
MOSFET to be switched on and off by swinging the gate
between VIN and ground. The phase invert (Pin 3) is tied
to ground to choose this operating mode. Normally, the
converter input (VIN) is connected to the LTC1266 supply
Pins 2 and 5 and can go as high as 20V. Pin 2 supplies the
high frequency current pulses to switch the MOSFETs and
should be decoupled with a 0.1µF to 1µF ceramic capacitor. Pin 5 supplies most of the quiescent power to the rest
of the chip.
12
Driving an N-channel topside MOSFET (PINV, Pin 3, tied to
PWR VIN) is a little trickier than driving a P-channel since
the gate voltage must be positive with respect to the
source to turn it on, which means that the gate voltage
must be higher than VIN. This requires either a second
supply at least VGS(ON) above VIN or a bootstrapping circuit
to boost the VIN to the proper level. The easiest method is
using a higher supply (see Figure 14) but if one is not
available, the bootstrap method can be used at the expense of an additional diode (see Figure 1). The bootstrap
works by charging the bootstrap capacitor to VIN during
the off-time. During the on-time, the bottom plate of the
capacitor is pulled up to VIN so that the voltage at Pin 2 is
now twice VIN (plus any ringing on the switch node).
Since the maximum allowable voltage at Pin 2 is 20V, the
Figure 1 bootstrap circuit limits VIN to less than 10V. A
higher VIN can be achieved if the bootstrap capacitor is
charged to a voltage less than VIN, in which case
VIN(MAX)␣ = 20 – VCAP.
N-channel mode, internal circuitry limits the maximum
on-time to 60µs to guarantee start-up of the bootstrap
circuit. This maximum on-time reduces the maximum
duty cycle to:
Max Duty Cycle =
60µs
60µs + tOFF
which slightly increases the minimum input voltage at
which dropout occurs. However, because of the superior
on-conductance of the N-channel, the dropout performance of an all N-channel regulator is still better (see
Figure 5) even with the duty cycle limitation, except at light
loads.
Low-Battery Comparator
The LTC1266 has an on-chip low-battery comparator
which can be used to sense a low-battery condition when
implemented as shown in Figure 6. The resistor divider
R1, R2 sets the comparator trip point as follows:
)
VTRIP = 1.25 1 + R2
R1
)
LTC1266
LTC1266-3.3/LTC1266-5
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100
VOUT = 3.3V
Burst Mode OPERATION
ENABLED
TOPSIDE
N-CHANNEL WITH
CHARGE PUMP
500
90
TOPSIDE
P-CHANNEL
400
EFFICIENCY (%)
VIN–V0UT (mV) AT DROPOUT
600
300
200
0
0
1
3
2
LOAD CURRENT
Burst Mode OPERATION
INHIBITED
70
TOPSIDE N-CHANNEL
WITH POWER VIN = 12V
100
80
4
60
0.01
5
0.1
1
LOAD CURRENT (A)
1266 F07
1266 F05
Figure 5. Comparison of Dropout Performance
VIN
R2
LTC1266
–
R1
5
LBOUT
+
1.25V
REFERENCE
1266 F06
Figure 6. Low-Battery Comparator
The divided down voltage at the “–” input to the comparator
is compared to an internal 1.25V reference. This reference
is separate from the 1.265V reference used by the voltage
comparator and current comparator for regulation and is
not disabled by the shutdown pin, therefore the low-battery
detection is operational even when the rest of the chip is
shut down. The comparator is functional down to an input
voltage of 2.5V. Thus, the output will provide a valid state
even when the rest of the chip does not have sufficient
voltage to operate. For best performance, the value of the
pull-up resistor should be high enough that the output is
pulled down to ground when sinking 200µA or less.
Figure 7. Effect of Disabling Burst Mode Operation on Efficiency
2. If the load is never expected to drop low enough to
benefit from the efficiency advantages of Burst Mode
operation, the output capacitor ESR and minimum capacitance requirements (which may falsely trigger Burst Mode
operation if not met) can be relaxed if Burst Mode operation is disabled.
3. If an auxiliary winding is used. Disabling Burst Mode
operation guarantees switching independent of the load
on the primary. This allows power to be taken from the
auxiliary winding independently.
4. Tighter load regulation (< 1%).
Burst Mode operation is disabled by applying a CMOS
logic high voltage (> 2.1V) to Pin 4. When it is disabled, the
voltage comparator limit is raised high enough so that it no
longer is involved in regulation; however it is still active
and is useful as a voltage clamp to keep the output from
overshooting.
Note that since the inductor current must reverse to
regulate the output at zero load when Burst Mode operation is disabled, the minimum inductance (LMIN) specified
during Inductor Core Selection is no longer applicable.
Suppressing Burst Mode Operation
Normally, enabling Burst Mode operation is desired due to
its superior efficiency at low load currents (see Figure 7).
However, in certain applications it may be desirable to
inhibit this feature. Some reasons for doing so are:
1. To eliminate audible noise from certain types of inductors at light loads.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, VOUT shifts by an
amount equal to ∆ILOAD (ESR), where ESR is the effective
series resistance of COUT. ∆ILOAD also begins to charge or
13
LTC1266
LTC1266-3.3/LTC1266-5
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Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:
% Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc., are the individual losses as a percentage of input power. (For high efficiency circuits, only small
errors are incurred by expressing losses as a percentage
of output power).
Although all dissipative elements in the circuit produce
losses, three main sources usually account for most of the
losses in LTC1266 series circuits: 1) LTC1266 DC bias
current, 2) MOSFET gate charge current and 3) I2R losses.
1. The DC supply current is the current which flows into
VIN (Pin 2). For VIN = 10V the LTC1266 DC supply current
is 170µA for no load, and increases proportionally with
load up to a constant 2.1mA after the LTC1266 series has
entered continuous mode. Because the DC bias current is
drawn from VIN, the resulting loss increases with input
voltage. For VIN = 5V the DC bias losses are generally less
than 1% for load currents over 30mA. However, at very
low load currents the DC bias current accounts for nearly
all of the loss.
2. MOSFET gate charge current results from switching the
gate capacitance of the power MOSFETs. Each time a
MOSFET gate is switched from low to high to low again, a
packet of charge dQ moves from Power VIN to ground. The
resulting dQ/dt is a current flowing into Power VIN (Pin 5)
which is typically much larger than the DC supply current.
In continuous mode, IGATECHG = f (QN + QP). The typical
gate charge for a 0.05Ω N-channel power MOSFET is
14
15nC. This results in IGATECHG = 6mA in 200kHz continuous operation for a 2% to 3% typical mid-current loss with
VIN = 5V.
Note that the gate charge loss increases directly with
both input voltage and operating frequency. This is the
principal reason why the highest efficiency circuits operate at moderate frequencies. Furthermore, it argues against
using larger MOSFETs than necessary to control I2R
losses, since overkill can cost efficiency as well as money!
3. I2R losses are easily predicted from the DC resistances
of the MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L and
RSENSE, but is “chopped” between the topside and bottom-side MOSFETs. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one
MOSFET can simply be summed with the resistances of L
and RSENSE to obtain I2R losses. For example, if each
RDS(ON) = 0.05Ω, RL = 0.05Ω and RSENSE = 0.02Ω, then
the total resistance is 0.12Ω. This results in losses ranging
from 3.5% to 15% as the output current increases from 1A
to 5A. I2R losses cause the efficiency to roll off at high
output currents.
Figure 8 shows how the efficiency losses in a typical
LTC1266 series regulator end up being apportioned. The
gate charge loss is responsible for the majority of the
efficiency lost in the mid-current region. If Burst Mode
operation was not employed at low currents, the gate
charge loss alone would cause efficiency to drop to
unacceptable levels (see Figure 7). With Burst Mode
100
I2R
GATE CHARGE
EFFICIENCY/LOSS (%)
discharge COUT until the regulator loop adapts to the
current change and returns VOUT to its steady-state value.
During this recovery time VOUT can be monitored for
overshoot or ringing which would indicate a stability
problem. The Pin 7 external components shown in the
Figure 1 circuit will prove adequate compensation for
most applications.
95
LTC1266 IQ
90
85
80
0.01
0.03
0.3
0.1
IOUT (A)
1
5
1266 F08
Figure 8. Efficiency Loss
LTC1266
LTC1266-3.3/LTC1266-5
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operation, the DC supply current represents the lone (and
unavoidable) loss component which continues to become
a higher percentage as output current is reduced. As
expected the I2R losses dominate at high load currents.
Other losses including CIN and COUT ESR dissipative
losses, MOSFET switching losses, Schottky conduction
losses during deadtime and inductor core losses, generally account for less than 2% total additional loss.
Design Example
As a design example, assume VIN = 5V (nominal),
VOUT = 3.3V, IMAX = 5A and f = 200kHz; RSENSE, CT and L
can immediately be calculated:
RSENSE = 100mV/5 = 0.02Ω
tOFF = (1/200kHz) • [1 – (3.3/5)] = 1.7µs
CT = 1.7µs/(1.3 • 104) = 130pF
LMIN = 5.1 • 105 • 0.02Ω • 130pF • 3.3V = 5µH
Assume that the MOSFET dissipations are to be limited to
PT = PB = 2W.
If TA = 40°C and the thermal resistance of each MOSFET
is 50°C/ W, then the junction temperatures will be 140°C
and δT = δB = 0.60. The required RDS(ON) for each MOSFET
can now be calculated:
TS RDS(ON) =
5(2)
= 0.076Ω
3.3(5)2 (1.60)
BS RDS(ON) =
5(2)
= 0.147Ω
1.7(5)2 (1.60)
The topside FET requirement can be met by an N-channel
Si9410DY which has an RDS(ON) of about 0.04Ω at
VGS = 5V. The bottom-side FET requirement is exceeded
by an Si9410DY. Note that the most stringent requirement
for the bottom-side MOSFET is with VOUT = 0 (i.e., short
circuit). During a continuous short circuit, the worst-case
dissipation rises to:
PB = ISC(AVG)2 • RDS(ON) • (1 + δB)
With the 0.02Ω sense resistor, ISC(AVG) ≈ 6A will result,
increasing the 0.04Ω bottom-side FET dissipation to 2.3W.
CIN will require an RMS current rating of at least 2.5A at
temperature and COUT will require an ESR of 0.02Ω for
optimum efficiency.
Now allow VIN to drop to its minimum value. The minimum
VIN can be calculated from the maximum duty cycle and
voltage drop across the topside FET,
VMIN =
VOUT + ILOAD • (RDS(ON) + RL + RSENSE)
DMAX
= 4.0V
At this lower input voltage, the operating frequency decreases and the topside FET will be conducting most of the
time, causing the power dissipation to increase.
At dropout,
fMIN =
1
= 16kHz
tON (MAX) + tOFF
PT = I2LOAD • RDS(ON) • (1 + δT) • DMAX
This last step is necessary to assure that the power
dissipation and junction temperature of the topside FET
are not exceeded.
These last calculations assume that Power VIN is high
enough to keep the topside FET fully turned on at dropout,
as would be the case with the Figure 11circuit. If this isn’t
true (as with the Figure 1 circuit) the RDS(ON) will increase
which in turn increases VMIN and PT.
Adjustable Applications
When an output voltage other than 3.3V or 5V is required,
the LTC1266 adjustable version is used with an external
resistive divider from VOUT to VFB, Pin 10. The regulated
voltage is determined by:
)
VOUT = 1.265 1 + R2
R1
)
To prevent stray pickup a 100pF capacitor is suggested
across R1 located close to the LTC1266.
For Figure 1 applications with VOUT below 2V, or when
RSENSE is moved to ground, the current sense comparator
inputs operate near ground. When the current comparator
is operated at less than 2V common mode, the off-time
increases approximately 40%, requiring the use of a
smaller timing capacitor CT.
15
LTC1266
LTC1266-3.3/LTC1266-5
U
W
U U
APPLICATIO S I FOR ATIO
Troubleshooting Hints
Since efficiency is critical to LTC1266 series applications,
it is very important to verify that the circuit is functioning
correctly in both continuous and Burst Mode operation.
The waveform to monitor is the voltage on the timing
capacitor, Pin 6.
In continuous mode (ILOAD > IBURST) the voltage on the CT
pin should be a sawtooth with a 0.9VP-P swing. This
voltage should never dip below 2V as shown in Figure 9a.
When load currents are low (ILOAD < IBURST) Burst Mode
operation should occur with the CT pin waveform periodically falling to ground for periods of time as shown in
Figure 9b.
3.3V
0V
(a) Continuous Mode Operation
3.3V
0V
(b) Burst Mode Operation
1266 F09
If Pin 6 is observed falling to ground at high output
currents, it indicates poor decoupling or improper grounding. Refer to the Board Layout Checklist.
Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1266 series. These items are also illustrated graphically in the layout diagram of Figure 10. Check the following in your layout:
1. Are the signal and power grounds segregated? The
LTC1266 signal ground (Pin 12) must return to the (–)
plate of COUT. The power ground returns to the source
of the bottom-side MOSFET, anode of the Schottky
diode and (–) plate of CIN, which should have as short
lead lengths as possible.
2. Does the LTC1266 SENSE – (Pin 8) connect to a point
close to RSENSE and the (+) plate of COUT? In adjustable
applications, the resistive divider R1 and R2 must be
connected between the (+) plate of COUT and signal
ground.
Figure 9. CT Waveforms
+
BOLD LINES INDICATE
HIGH CURRENT PATHS
VIN
CIN
CB
+
1
2
3
TDRIVE
BDRIVE
PGND
PWR VIN
LBOUT
LTC1266
4
LBIN
BINH
5
6
7
CT
3300pF 8
PINV
VIN
CT
ITH
SENSE –
SGND
SHDN
VFB
SENSE +
–
L
16
15
14
13
12
11
–
SHUTDOWN
R1
10
9
+
COUT
R2
VOUT
RSENSE
+
470Ω
1000pF
OUTPUT DIVIDER REQUIRED WITH
ADJUSTABLE VERSION ONLY
1266 F10
Figure 10. LTC1266 Layout Diagram (See Layout Checklist)
16
LTC1266
LTC1266-3.3/LTC1266-5
U
W
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APPLICATIO S I FOR ATIO
3. Are the SENSE– and SENSE+ leads routed together with
minimum PC trace spacing? The 1000pF capacitor between Pins 8 and 9 should be as close as possible to the
LTC1266.
helpful in eliminating instabilities at high input voltage and
high output loads.
6. Is the shutdown (Pin 11) actively pulled to ground
during normal operation? The shutdown pin is high impedance and must not be allowed to float. The select (Pins
3 and 4) are also high impedance and must be tied high or
low depending on the application.
4. Does the (+) plate of CIN connect to the source of the
topside MOSFET as closely as possible? This capacitor
provides the AC current to the topside MOSFET.
5. A 0.1µF to 1µF decoupling capacitor connected between VIN (Pin 5) and ground is optional, but is sometimes
U
TYPICAL APPLICATIO S
(Layout Assist Schematics)
VIN
≈ 3.9V TO 18V
(VIN(MIN) = 3.5V IF ILOAD < 0.8A)
1µF
+
1
2
3
4
BINH
5
6
7
CT
220pF
CC
3300pF
RC
1k
8
TDRIVE
BDRIVE
PWR VIN
PGND
PINV
LBOUT
LBIN
BINH
LTC1266-3.3
VIN
SGND
CT
SHDN
ITH
NC
SENSE –
SENSE +
1000pF
16
Si9430DY
+
Si9410DY
D1
MBRS140T3
CIN
100µF
25V
15
14
13
12
11
10
SHUTDOWN
L*
10µH
+
9
COUT
220µF
10V
2×
RSENSE
0.033Ω
*DALE LPT4545-A001
COILTRONICS CTX10-4
VOUT
3.3V
3A 1266 F11
Figure 11. Low Dropout, 3.3V/3A High Efficiency Regulator
17
LTC1266
LTC1266-3.3/LTC1266-5
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TYPICAL APPLICATIO S
(Layout Assist Schematics)
VIN
4.3V TO 10V
(VIN (MIN) = 3.5V IF ILOAD < 100mA
0.068Ω
+
0.1µF
2
3
4
BINH
5
6
7
CT
200pF
CC
3300pF
8
TDRIVE
BDRIVE
PGND
PWR VIN
LBOUT
PINV
LBIN
BINH
LTC1266
VIN
SGND
CT
SHDN
ITH
VFB
SENSE +
SENSE –
RC
1k
VOUT
12V/500mA
CIN
100µF
20V
127k
1%
+
Si9410DY
1M
1
D1
MBRS130LT3
L*
20µH
16
1M
15k
1%
100pF
C0UT
100µF
20V
15
14
Q1**
13
12
11
10
9
180k
1000pF
1N4148
SHUTDOWN
100k
*DALE LPT4545-A002
COILTRONICS CTX20-4
**MMBT2222ALT1
1266 F12
Figure 12. 5V to 12V/500mA High Efficiency Boost Regulator
VIN
4V TO PWR VIN – 4.5V
(VIN(MIN) = 3.5V IF ILOAD < 2.5A)
Si9410DY
1µF
+
PWR VIN
VIN + 4.5V TO 18V
1
2
3
4
BINH
5
6
7
CT
180pF
CC
3300pF
RC
470Ω
*TOKO 919AS-4R7M
8
TDRIVE
BDRIVE
PWR VIN
PGND
PINV
LBOUT
BINH
LBIN
LTC1266-3.3
VIN
SGND
SHDN
CT
NC
ITH
SENSE –
SENSE +
1000pF
16
CIN
100µF
20V
2×
+
D1
MBRS140T3
Si9410DY
15
14
13
12
11
10
SHUTDOWN
L*
5µH
+
9
COUT
220µF
10V
2×
RSENSE
0.02Ω
1266 F13
VOUT
3.3V
5A
Figure 13. All N-Channel 5V to 3.3V/5A Converter with Drivers Powered from External PWR VIN Supply
18
LTC1266
LTC1266-3.3/LTC1266-5
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TYPICAL APPLICATIO S
(Layout Assist Schematics)
VIN
4V TO 9V
0.1µF MBR0530T1
1
2
3
4
BINH
5
6
7
CT
220pF
CC
3300pF
8
TDRIVE
BDRIVE
PWR VIN
PGND
PINV
LBOUT
BINH
LBIN
LTC1266-3.3
VIN
SGND
CT
SHDN
ITH
NC
SENSE –
RC
470Ω
SENSE +
16
Si4410DY
+
Si4410DY
D1
MBRS340T3
47µF
10V
OS-CON
3×
15
14
13
12
11
SHUTDOWN
L*
5µH
10
COUT
330µF
10V
3×
+
9
1000pF
RSENSE
0.01Ω
1266 F14
*MAGNETICS Kool Mµ 77120-A7
VOUT
3.3V
10A
Figure 14. All N-Channel 5V to 3.3V/10A High Efficiency Regulator
VIN
4V TO 9V
(VIN(MIN) = 3.5V IF ILOAD < 1A)
0.1µF MBR0530T1
1
2
3
4
BINH
5
6
7
CT
180pF
CC
3300pF
RC
470Ω
8
TDRIVE
BDRIVE
PWR VIN
PGND
PINV
LBOUT
BINH
LBIN
LTC1266
VIN
SGND
CT
SHDN
ITH
VFB
SENSE –
SENSE +
1000pF
16
Si9410DY
+
Si9410DY
D1
MBRS130T3
100µF
10V
OS-CON
2×
15
14
13
12
11
SHUTDOWN
L*
5µH
100pF
100k
1%
10
+
9
RSENSE
0.02Ω
97.6k
1%
1266 F15
*TOKO 919AS-4R7M
COUT
330µF
10V
2×
VOUT
2.5V
5A
Figure 15. All N-Channel 5V to 2.5V/5A High Efficiency Regulator
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LTC1266
LTC1266-3.3/LTC1266-5
U
PACKAGE DESCRIPTIO
Dimensions in inches (millimeters) unless otherwise noted.
S Package
16-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.386 – 0.394*
(9.804 – 10.008)
16
15
14
13
12
11
10
9
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
2
3
5
4
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
TYP
7
8
0.004 – 0.010
(0.101 – 0.254)
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
6
0.050
(1.270)
BSC
S16 1098
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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VOUT Up to 36V
LTC1628
2-Phase, Dual Synchronous Controller
Minimizes CIN and COUT, Two Outputs 4V ≤ VIN ≤ 36V,
IOUT Up to 20A
LTC1735
High Efficiency Synchronous Controller
Wide Input Range 3.5V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 6V
OPTI-LOOPTM Compensation Minimizes COUT
LTC1772
SOT-23 P-Channel Controller
Tiny Design, 550kHz, 2.5V ≤ VIN ≤ 9.8V, IOUT Up to 4.5A
LTC1929
42A 2-Phase Synchronous Controller for Single Output
IOUT Up to 42A with Single Controller, Minimizes CIN
and COUT, Up to 200A Out
No RSENSE and OPTI-LOOP are trademarks of Linear Technology Corporation.
20
Linear Technology Corporation
1266fa LT/TP 1000 2K REV A • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
LINEAR TECHNOLOGY CORPORATION 1995
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