LTC1435
High Efficiency Low Noise
Synchronous Step-Down
Switching Regulator
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DESCRIPTION
FEATURES
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The LTC®1435 is a synchronous step-down switching
regulator controller that drives external N-channel power
MOSFETs using a fixed frequency architecture. Burst
ModeTM operation provides high efficiency at low load
currents. A maximum duty cycle limit of 99% provides low
dropout operation which extends operating time in battery-operated systems.
Dual N-Channel MOSFET Synchronous Drive
Programmable Fixed Frequency
Wide VIN Range: 3.5V to 36V Operation
Ultrahigh Efficiency
Very Low Dropout Operation: 99% Duty Cycle
Low Standby Current
Secondary Feedback Control
Programmable Soft Start
Remote Output Voltage Sense
Logic Controlled Micropower Shutdown: IQ < 25µA
Foldback Current Limiting (Optional)
Current Mode Operation for Excellent Line and Load
Transient Response
Output Voltages from 1.19V to 9V
Available in 16-Lead Narrow SO and SSOP Packages
The operating frequency is set by an external capacitor
allowing maximum flexibility in optimizing efficiency. A
secondary winding feedback control pin, SFB, guarantees
regulation regardless of load on the main output by
forcing continuous operation. Burst Mode operation is
inhibited when the SFB pin is pulled low which reduces
noise and RF interference.
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APPLICATIONS
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Notebook and Palmtop Computers, PDAs
Cellular Telephones and Wireless Modems
Portable Instruments
Battery-Operated Devices
DC Power Distribution Systems
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
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Soft start is provided by an external capacitor which can
be used to properly sequence supplies. The operating
current level is user-programmable via an external current
sense resistor. Wide input supply range allows operation
from 3.5V to 30V (36V maximum).
TYPICAL APPLICATION
COSC
68pF
CSS
0.1µF
CC
330pF
VIN
4.5V TO 28V
COSC
VIN
RUN/SS
TG
ITH
SW
M1
Si4412DY
BOOST
100pF
VOSENSE
SENSE –
BG
RSENSE
0.033Ω
VOUT
2.9V/3.5A
R1
32.4k
CB
0.1µF
INTVCC
SGND
CIN
22µF
35V
×2
L1
10µH
DB
CMDSH-3
LTC1435
RC
10k
+
+
4.7µF
M2
Si4412DY
R2
22.1k
D1
MBRS140T3
COUT
+ 100µF
10V
×2
PGND
SENSE +
1000pF
1435 F01
Figure 1. High Efficiency Step-Down Converter
1
LTC1435
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SYMBOL
PARAMETER
Main Control Loop
IIN VOSENSE Feedback Current
VOSENSE
Feedback Voltage
∆VLINEREG
Reference Voltage Line Regulation
∆VLOADREG
Output Voltage Load Regulation
VSFB
ISFB
VOVL
IQ
Secondary Feedback Threshold
Secondary Feedback Current
Output Overvoltage Lockout
Input DC Supply Current
Normal Mode
Shutdown
VRUN/SS
Run Pin Threshold
IRUN/SS
Soft Start Current Source
∆VSENSE(MAX) Maximum Current Sense Threshold
TG Transition Time
Rise Time
TG t r
TG t f
Fall Time
BG Transition Time
Rise Time
BG tr
BG t f
Fall Time
Internal VCC Regulator
VINTVCC
Internal VCC Voltage
VLDO INT
INTVCC Load Regulation
VLDO EXT
EXTVCC Voltage Drop
VEXTVCC
EXTVCC Switchover Voltage
Oscillator
fOSC
Oscillator Frequency
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ELECTRICAL CHARACTERISTICS
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Input Supply Voltage (VIN)......................... 36V to – 0.3V
Topside Driver Supply Voltage (Boost) ......42V to – 0.3V
Switch Voltage (SW)............................. VIN + 5V to – 5V
EXTVCC Voltage ........................................ 10V to – 0.3V
Sense+, Sense– Voltages ......... INTVCC + 0.3V to – 0.3V
ITH, VOSENSE Voltages .............................. 2.7V to – 0.3V
SFB, Run/SS Voltages .............................. 10V to – 0.3V
Peak Driver Output Current < 10µs (TG, BG) ............. 2A
INTVCC Output Current ........................................ 50mA
Operating Ambient Temperature Range
LTC1435C............................................... 0°C to 70°C
LTC1435I............................................ – 40°C to 85°C
Junction Temperature (Note 1)............................. 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
ORDER PART
NUMBER
TOP VIEW
COSC 1
RUN/SS 2
16 TG
15 BOOST
ITH 3
14 SW
SFB 4
13 VIN
SGND 5
LTC1435CG
LTC1435CS
LTC1435IG
LTC1435IS
12 INTVCC
VOSENSE 6
11 BG
SENSE– 7
10 PGND
SENSE+ 8
9
EXTVCC
G PACKAGE
S PACKAGE
16-LEAD PLASTIC SSOP 16-LEAD PLASTIC SO
TJMAX = 125°C, θJA = 130°C/ W (G)
TJMAX = 125°C, θJA = 110°C/ W (S)
Consult factory for Military grade parts.
TA = 25°C, VIN = 15V, VRUN/SS = 5V unless otherwise noted.
CONDITIONS
TYP
MAX
UNITS
1.24
10
1.19
0.002
0.5
– 0.5
1.19
–1
1.28
50
1.202
0.01
0.8
– 0.8
1.22
–2
1.32
nA
V
%/V
%
%
V
µA
V
0.8
1.5
130
260
16
1.3
3
150
25
2
4.5
180
µA
µA
V
µA
mV
CLOAD = 3000pF
CLOAD = 3000pF
50
50
150
150
ns
ns
CLOAD = 3000pF
CLOAD = 3000pF
50
40
150
150
ns
ns
5.2
–1
230
V
%
mV
V
138
kHz
(Note 2)
(Note 2)
VIN = 3.6V to 20V (Note 2)
ITH Sinking 5µA (Note 2)
ITH Sourcing 5µA
VSFB Ramping Negative
VSFB = 1.5V
MIN
●
1.178
●
●
●
1.16
EXTVCC = 5V (Note 3)
3.6V < VIN < 30V
VRUN/SS = 0V, 3.6V < VIN < 15V
●
VRUN/SS = 0V
VOSENSE = 0V, 5V
6V < VIN < 30V, VEXTVCC = 4V
IINTVCC = 15mA, VEXTVCC = 4V
IINTVCC = 15mA, VEXTVCC = 5V
IINTVCC = 15mA, VEXTVCC Ramping Positive
COSC = 100pF (Note 4)
●
4.8
●
4.5
5.0
– 0.2
130
4.7
112
125
LTC1435
ELECTRICAL CHARACTERISTICS
TA = 25°C, VIN = 15V, VRUN/SS = 5V unless otherwise noted.
The ● denotes specifications which apply over the full operating
temperature range.
LTC1435CG/LTC1435CS: 0°C ≤ TA ≤ 70°C
LTC1435IG/LTC1435IS: – 40°C ≤ TA ≤ 85°C
Note 1: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
LTC1435CG/LTC1435IG: TJ = TA + (PD)(130°C/W)
LTC1435CS/LTC1435IS: TJ = TA + (PD)(110°C/W)
Note 2: The LTC1435 is tested in a feedback loop which servos VOSENSE
to the balance point for the error amplifier (VITH = 1.19V).
Note 3: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 4: Oscillator frequency is tested by measuring the COSC charge and
discharge currents and applying the formula:
(
)(
)
8.4(108)
1 + 1 –1
fOSC (kHz) = C
(pF)
+
11
I
OSC
CHG IDIS
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TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage
VOUT = 3.3V
Efficiency vs Input Voltage
VOUT = 5V
VOUT = 3.3V
VOUT = 5V
90
ILOAD = 1A
ILOAD = 1A
85
ILOAD = 100mA
80
90
EFFICIENCY (%)
EFFICIENCY (%)
90
ILOAD = 100mA
85
80
75
75
70
70
VIN = 10V
VOUT = 5V
RSENSE = 0.033Ω
95
95
95
EFFICIENCY (%)
Efficiency vs Load Current
100
100
100
85
80
75
Burst Mode
OPERATION
70
CONTINUOUS
MODE
65
60
55
0
5
10
15
20
INPUT VOLTAGE (V)
25
30
0
5
10
15
20
INPUT VOLTAGE (V)
25
30
VIN – VOUT Dropout Voltage
vs Load Current
Load Regulation
VITH Pin Voltage vs Output Current
3.0
RSENSE = 0.033Ω
∆VOUT (%)
0.4
0.3
0.2
0.1
– 0.25
2.5
– 0.50
2.0
VITH (V)
RSENSE = 0.033Ω
VOUT DROP OF 5%
– 0.75
0.5
1.0
1.5
2.0
LOAD CURRENT (A)
2.5
3.0
1435 G04
1.5
Burst Mode
OPERATION
–1.00
1.0
–1.25
0.5
–1.50
0
0
10
1435 G03
0
0.5
1
0.01
0.1
LOAD CURRENT (A)
1435 G02
1435 G01
VIN – VOUT (V)
50
0.001
CONTINUOUS
MODE
0
0
0.5
1.0
1.5
2.0
LOAD CURRENT (A)
2.5
3.0
1435 G05
0
10 20 30 40 50 60 70 80 90 100
OUTPUT CURRENT (%)
1435 G06
3
LTC1435
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TYPICAL PERFORMANCE CHARACTERISTICS
Input Supply and Shutdown
Current vs Input Voltage
100
VOUT = 5V
EXTVCC = VOUT
60
VOUT = 3.3V
EXTVCC = OPEN
1.0
40
0.5
20
200
VEXTVCC = 0V
180
70°C
0
25°C
– 0.3
0
5
10
15
20
INPUT VOLTAGE (V)
140
25°C
120
100
– 55°C
80
60
40
20
SHUTDOWN
0
70°C
160
0.3
EXTVCC – INTVCC (mV)
SUPPLY CURRENT (mA)
80
SHUTDOWN CURRENT (µA)
2.0
0.5
∆INTVCC (%)
2.5
1.5
EXTVCC Switch Drop
vs INTVCC Load Current
INTVCC Regulation
vs INTVCC Load Current
25
30
0
– 0.5
0
0
10
15
5
INTVCC LOAD CURRENT (mA)
1435 G07
20
0
2
4 6 8 10 12 14 16 18 20
INTVCC LOAD CURRENT (mA)
1435 G09
1435 G08
RUN/SS Pin Current
vs Temperature
Normalized Oscillator Frequency
vs Temperature
10
4
5
3
SFB Pin Current vs Temperature
0
fO
–5
SFB CURRENT (µA)
RUN/SS CURRENT (µA)
FREQUENCY (%)
– 0.25
2
– 0.50
– 0.75
–1.00
1
–1.25
–10
– 40 –15
60
35
85
10
TEMPERATURE (°C)
110
135
0
– 40 –15
85
10
35
60
TEMPERATURE (°C)
110
135
–1.50
– 40 –15
60
35
85
10
TEMPERATURE (°C)
1435 G11
1435 G10
Maximum Current Sense
Threshold Voltage vs Temperature
110
135
1435 G12
Transient Response
Transient Response
CURRENT SENSE THRESHOLD (mV)
154
152
VOUT
50mV/DIV
VOUT
50mV/DIV
150
148
ILOAD = 50mA to 1A
146
– 40 –15
85
10
35
60
TEMPERATURE (°C)
110
135
1435 G13
4
1435 G14
ILOAD = 1A to 3A
1435 G15
LTC1435
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TYPICAL PERFORMANCE CHARACTERISTICS
Soft Start: Load Current vs Time
Burst Mode Operation
VOUT
20mV/DIV
RUN/SS
5V/DIV
INDUCTOR
CURRENT
1A/DIV
VITH
200mV/DIV
ILOAD = 50mA
1435 G16
1435 G17
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PIN FUNCTIONS
COSC (Pin 1): External capacitor COSC from this pin to
ground sets the operating frequency.
RUN/SS (Pin 2): Combination of Soft Start and Run
Control Inputs. A capacitor to ground at this pin sets the
ramp time to full current output. The time is approximately
0.5s/µF. Forcing this pin below 1.3V causes the device to
be shut down. In shutdown all functions are disabled.
ITH (Pin 3): Error Amplifier Compensation Point. The
current comparator threshold increases with this control
voltage. Nominal voltage range for this pin is 0V to 2.5V.
SFB (Pin 4): Secondary Winding Feedback Input. Normally connected to a feedback resistive divider from the
secondary winding. This pin should be tied to: ground to
force continuous operation; INTVCC in applications that
don’t use a secondary winding; and a resistive divider from
the output in applications using a secondary winding.
SGND (Pin 5): Small-Signal Ground. Must be routed
separately from other grounds to the (–) terminal of COUT.
VOSENSE (Pin 6): Receives the feedback voltage from an
external resistive divider across the output.
SENSE – (Pin 7): The (–) Input to the Current Comparator.
SENSE + (Pin 8): The (+) Input to the Current Comparator.
Built-in offsets between SENSE– and SENSE+ pins in
conjunction with RSENSE set the current trip thresholds.
EXTVCC (Pin 9): Input to the Internal Switch Connected to
INTVCC. This switch closes and supplies VCC power when-
ever EXTVCC is higher than 4.7V. See EXTVCC connection
in Applications Information section. Do not exceed 10V on
this pin. Connect to VOUT if VOUT ≥ 5V.
PGND (Pin 10): Driver Power Ground. Connects to source
of bottom N-channel MOSFET and the (–) terminal of CIN.
BG (Pin 11): High Current Gate Drive for Bottom
N-Channel MOSFET. Voltage swing at this pin is from
ground to INTVCC.
INTVCC (Pin 12): Output of the Internal 5V Regulator and
EXTVCC Switch. The driver and control circuits are powered from this voltage. Must be closely decoupled to power
ground with a minimum of 2.2µF tantalum or electrolytic
capacitor.
VIN (Pin 13): Main Supply Pin. Must be closely decoupled
to the IC’s signal ground pin.
SW (Pin 14): Switch Node Connection to Inductor. Voltage swing at this pin is from a Schottky diode (external)
voltage drop below ground to VIN.
BOOST (Pin 15): Supply to Topside Floating Driver. The
bootstrap capacitor is returned to this pin. Voltage swing
at this pin is from INTVCC to VIN + INTVCC.
TG (Pin 16): High Current Gate Drive for Top N-Channel
MOSFET. This is the output of a floating driver with a
voltage swing equal to INTVCC superimposed on the
switch node voltage SW.
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LTC1435
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FUNCTIONAL DIAGRA
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VIN
COSC
+
CIN
1 COSC
4 SFB
13 VIN
SGND 5
INTVCC
1.19V
REF
1µA
DB
BOOST
15
–
1.19V
TG
16
SHUTDOWN
OSC
+
CB
+
DROP
OUT
DET
OV
S
Q
R
–
1.28V
0.6V
SWITCH
LOGIC
+
–
SW
14
VOSENSE
6
VFB
–
–
I1
EA
+
1.19V
Ω
R2
gm = 1m
+
180k
I2
–
4k
D1
+
VIN
+
VSEC
INTVCC
INTVCC
CSEC
+
12
+
–
SHUTDOWN
R1
5V
LDO
REG
3µA
6V
RUN
SOFT
START
30k
+
4.8V
BG
11
8k
VOUT
–
RC
2 RUN/SS
CSS
3 ITH
CC
DFB*
SENSE+ 8
7 SENSE
–
9 EXTVCC
COUT
PGND
10
+
RSENSE
1435 • FD
* FOLDBACK CURRENT LIMITING OPTION
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LTC1435
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OPERATION
(Refer to Functional Diagram)
Main Control Loop
Low Current Operation
The LTC1435 uses a constant frequency, current mode
step-down architecture. During normal operation, the top
MOSFET is turned on each cycle when the oscillator sets
the RS latch, and turned off when the main current
comparator I1 resets the RS latch. The peak inductor
current at which I1 resets the RS latch is controlled by the
voltage on the ITH pin , which is the output of error amplifier
EA. The VOSENSE pin, described in the Pin Functions
section, allows EA to receive an output feedback voltage
VFB from an external resistive divider. When the load
current increases, it causes a slight decrease in VFB
relative to the 1.19V reference, which in turn causes the ITH
voltage to increase until the average inductor current
matches the new load current. While the top MOSFET is
off, the bottom MOSFET is turned on until either the
inductor current starts to reverse, as indicated by current
comparator I2, or the beginning of the next cycle.
The LTC1435 is capable of Burst Mode operation in which
the external MOSFETs operate intermittently based on
load demand. The transition to low current operation
begins when comparator I2 detects current reversal and
turns off the bottom MOSFET. If the voltage across RSENSE
does not exceed the hysteresis of I2 (approximately 20mV)
for one full cycle, then on following cycles the top and
bottom drives are disabled. This continues until an inductor current peak exceeds 20mV/RSENSE or the ITH voltage
exceeds 0.6V, either of which causes drive to be returned
to the TG pin on the next cycle.
The top MOSFET driver is biased from floating bootstrap
capacitor CB, which normally is recharged during each off
cycle. However, when VIN decreases to a voltage close to
VOUT, the loop may enter dropout and attempt to turn on
the top MOSFET continuously. The dropout detector counts
the number of oscillator cycles that the top MOSFET
remains on and periodically forces a brief off period to
allow CB to recharge.
The main control loop is shut down by pulling the RUN/SS
pin low. Releasing RUN/SS allows an internal 3µA current
source to charge soft start capacitor CSS. When CSS
reaches 1.3V, the main control loop is enabled with the ITH
voltage clamped at approximately 30% of its maximum
value. As CSS continues to charge, ITH is gradually released allowing normal operation to resume.
Two conditions can force continuous synchronous operation, even when the load current would otherwise dictate
low current operation. One is when the common mode
voltage of the SENSE+ and SENSE – pins is below 1.4V and
the other is when the SFB pin is below 1.19V. The latter
condition is used to assist in secondary winding regulation
as described in the Applications Information section.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
of the other LTC1435 circuitry is derived from the INTVCC
pin. The bottom MOSFET driver supply pin is internally
connected to INTVCC in the LTC1435. When the EXTVCC
pin is left open, an internal 5V low dropout regulator
supplies INTVCC power. If EXTVCC is taken above 4.8V,
the 5V regulator is turned off and an internal switch is
turned on to connect EXTVCC to INTVCC. This allows the
INTVCC power to be derived from a high efficiency
external source such as the output of the regulator itself
or a secondary winding, as described in the Applications
Information section.
Comparator OV guards against transient overshoots
> 7.5% by turning off the top MOSFET and keeping it off
until the fault is removed.
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LTC1435
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APPLICATIONS INFORMATION
300
The basic LTC1435 application circuit is shown in Figure
1, High Efficiency Step-Down Converter. External component selection is driven by the load requirement and
begins with the selection of RSENSE. Once RSENSE is
known, COSC and L can be chosen. Next, the power
MOSFETs and D1 are selected. Finally, CIN and COUT are
selected. The circuit shown in Figure 1 can be configured
for operation up to an input voltage of 28V (limited by the
external MOSFETs).
COSC VALUE (pF)
250
200
150
100
50
0
RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
The LTC1435 current comparator has a maximum threshold of 150mV/RSENSE and an input common mode range
of SGND to INTVCC. The current comparator threshold
sets the peak of the inductor current, yielding a maximum
average output current IMAX equal to the peak value less
half the peak-to-peak ripple current ∆IL.
Allowing a margin for variations in the LTC1435 and
external component values yields:
RSENSE =
100mV
IMAX
The LTC1435 works well with values of RSENSE from
0.005Ω to 0.2Ω.
COSC Selection for Operating Frequency
The LTC1435 uses a constant frequency architecture with
the frequency determined by an external oscillator capacitor COSC. Each time the topside MOSFET turns on, the
voltage COSC is reset to ground. During the on-time, COSC
is charged by a fixed current. When the voltage on the
capacitor reaches 1.19V, COSC is reset to ground. The
process then repeats.
The value of COSC is calculated from the desired operating
frequency:
1.37(104 )
– 11
COSC (pF) =
Frequency (kHz)
A graph for selecting COSC vs frequency is given in Figure
2. As the operating frequency is increased the gate charge
8
0
100
200
300
400
OPERATING FREQUENCY (kHz)
500
LTC1435 • F02
Figure 2. Timing Capacitor Value
losses will be higher, reducing efficiency (see Efficiency
Considerations). The maximum recommended switching
frequency is 400kHz.
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
MOSFET gate charge losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
The inductor value has a direct effect on ripple current. The
inductor ripple current ∆IL decreases with higher inductance or frequency and increases with higher VIN or VOUT:
∆IL =
V
1
VOUT 1– OUT
VIN
( f)(L)
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.4(IMAX). Remember, the
maximum ∆IL occurs at the maximum input voltage.
The inductor value also has an effect on low current
operation. The transition to low current operation begins
when the inductor current reaches zero while the bottom
LTC1435
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APPLICATIONS INFORMATION
MOSFET is on. Lower inductor values (higher ∆IL) will
cause this to occur at higher load currents, which can
cause a dip in efficiency in the upper range of low current
operation. In Burst Mode operation, lower inductance
values will cause the burst frequency to decrease.
The Figure 3 graph gives a range of recommended inductor values vs operating frequency and VOUT.
60
VOUT = 5.0V
VOUT = 3.3V
VOUT = 2.5V
INDUCTOR VALUE (µH)
50
40
30
20
10
0
0
100
150
200
250
50
OPERATING FREQUENCY (kHz)
300
1435 F03
Figure 3. Recommended Inductor Values
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires more
turns of wire and therefore copper losses will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
Kool Mµ is a registered trademark of Magnetics, Inc.
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more
difficult. However, designs for surface mount are available
which do not increase the height significantly.
Power MOSFET and D1 Selection
Two external power MOSFETs must be selected for use
with the LTC1435: an N-channel MOSFET for the top
(main) switch and an N-channel MOSFET for the bottom
(synchronous) switch.
The peak-to-peak gate drive levels are set by the INTVCC
voltage. This voltage is typically 5V during start-up (see
EXTVCC Pin Connection). Consequently, logic level threshold MOSFETs must be used in most LTC1435 applications. The only exception is applications in which EXTVCC
is powered from an external supply greater than 8V (must
be less than 10V), in which standard threshold MOSFETs
(VGS(TH) < 4V) may be used. Pay close attention to the
BVDSS specification for the MOSFETs as well; many of the
logic level MOSFETs are limited to 30V or less.
Selection criteria for the power MOSFETs include the “ON”
resistance RSD(ON), reverse transfer capacitance CRSS,
input voltage and maximum output current. When the
LTC1435 is operating in continuous mode the duty cycles
for the top and bottom MOSFETs are given by:
V
Main Switch Duty Cycle = OUT
VIN
(V − V )
Synchronous Switch Duty Cycle = IN OUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
V
2
PMAIN = OUT (IMAX ) (1 + δ )RDS(ON) +
VIN
k(VIN )
1.85
(IMAX )(CRSS )( f)
V −V
2
PSYNC = IN OUT (IMAX ) (1 + δ )RDS(ON)
VIN
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where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
Both MOSFETs have I2R losses while the topside
N-channel equation includes an additional term for transition losses, which are highest at high input voltages.
For VIN < 20V the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V the
transition losses rapidly increase to the point that the use
of a higher RDS(ON) device with lower CRSS actual provides higher efficiency. The synchronous MOSFET losses
are greatest at high input voltage or during a short circuit
when the duty cycle in this switch is nearly 100%. Refer
to the Foldback Current Limiting section for further
applications information.
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. CRSS is usually specified in the MOSFET
characteristics. The constant k = 2.5 can be used to
estimate the contributions of the two terms in the main
switch dissipation equation.
The Schottky diode D1 shown in Figure 1 conducts during
the dead-time between the conduction of the two large
power MOSFETs. This prevents the body diode of the
bottom MOSFET from turning on and storing charge
during the dead-time, which could cost as much as 1% in
efficiency. A 1A Schottky is generally a good size for 3A
regulators.
CIN and COUT Selection
In continuous mode, the source current of the top
N-channel MOSFET is a square wave of duty cycle VOUT/
VIN. To prevent large voltage transients, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current is given by:
CIN required IRMS ≈ IMAX
[V (V
OUT
IN − VOUT
)]
1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations
10
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on only 2000 hours
of life. This makes it advisable to further derate the
capacitor or to choose a capacitor rated at a higher
temperature than required. Several capacitors may also be
paralleled to meet size or height requirements in the
design. Always consult the manufacturer if there is any
question.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement is satisfied the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:
1
∆VOUT ≈ ∆IL ESR +
4 fC OUT
where f = operating frequency, COUT = output capacitance
and ∆IL= ripple current in the inductor. The output ripple
is highest at maximum input voltage since ∆IL increases
with input voltage. With ∆IL = 0.4IOUT(MAX) the output
ripple will be less than 100mV at max VIN assuming:
COUT required ESR < 2RSENSE
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR(size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement.
In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalum, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include Sanyo OS-CON, Nichicon PL series and Sprague
593D and 595D series. Consult the manufacturer for other
specific recommendations.
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INTVCC Regulator
An internal P-channel low dropout regulator produces the
5V supply which powers the drivers and internal circuitry
within the LTC1435. The INTVCC pin can supply up to
15mA and must be bypassed to ground with a minimum
of 2.2µF tantalum or low ESR electrolytic. Good bypassing
is necessary to supply the high transient currents required
by the MOSFET gate drivers.
High input voltage applications, in which large MOSFETs
are being driven at high frequencies, may cause the
maximum junction temperature rating for the LTC1435 to
be exceeded. The IC supply current is dominated by the
gate charge supply current when not using an output
derived EXTVCC source. The gate charge is dependent on
operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 1 of the
Electrical Characteristics. For example, the LTC1435 is
limited to less than 17mA from a 30V supply:
TJ = 70°C + (17mA)(30V)(100°C/W) = 126°C
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked when
operating in continuous mode at maximum VIN.
EXTVCC Connection
The LTC1435 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
The switch closes and supplies the INTVCC power whenever the EXTVCC pin is above 4.8V, and remains closed
until EXTVCC drops below 4.5V. This allows the MOSFET
driver and control power to be derived from the output
during normal operation (4.8V < VOUT < 9V) and from the
internal regulator when the output is out of regulation
(start-up, short circuit). Do not apply greater than 10V to
the EXTVCC pin and ensure that EXTVCC < VIN.
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by a
factor of Duty Cycle/Efficiency. For 5V regulators this
supply means connecting the EXTVCC pin directly to VOUT.
However, for 3.3V and other lower voltage regulators,
additional circuitry is required to derive INTVCC power
from the output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5V regulator resulting
in an efficiency penalty of up to 10% at high input
voltages.
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
3. EXTVCC connected to an output-derived boost network.
For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage which has been boosted to
greater than 4.8V. This can be done with either the
inductive boost winding as shown in Figure 4a or the
capacitive charge pump shown in Figure 4b. The charge
pump has the advantage of simple magnetics.
4. EXTVCC connected to an external supply. If an external
supply is available in the 5V to 10V range (EXTVCC ≤
VIN), it may be used to power EXTVCC providing it is
compatible with the MOSFET gate drive requirements.
When driving standard threshold MOSFETs, the external supply must always be present during operation to
prevent MOSFET failure due to insufficient gate drive.
+
VIN
CIN
1N4148
VIN
OPTIONAL
EXT VCC
CONNECTION
5V ≤ VSEC ≤ 9V
TG
N-CH
VOUT
COUT
SW
BG
R5
SGND
1µF
+
LTC1435
SFB
+
RSENSE
EXTVCC
R6
L1
1:N
VSEC
N-CH
PGND
LTC1435 • F04a
Figure 4a. Secondary Output Loop and EXTVCC Connection
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+
1.19V ≤ VOUT ≤ 9V
+
VIN
1µF
CIN
BAT85
0.22µF
R2
VOSENSE
BAT85
TG
SGND
BAT85
N-CH
EXTVCC
R1
LTC1435 • F05
VN2222LL
L1
RSENSE
VOUT
+
LTC1435
Figure 5. Setting the LTC1435 Output Voltage
COUT
SW
BG
100pF
LTC1435
VIN
3.3V OR 5V
N-CH
RUN/SS
RUN/SS
D1
PGND
CSS
LTC1435 • F04b
Figure 4b. Capacitive Charge Pump for EXTVCC
CSS
LTC1435 • F06
Figure 6. RUN/SS Pin Interfacing
Topside MOSFET Driver Supply (CB, DB)
An external bootstrap capacitor CB connected to the Boost
pin supplies the gate drive voltage for the topside MOSFET.
Capacitor CB in the Functional Diagram is charged through
diode DB from INTVCC when the SW pin is low. When the
topside MOSFET is to be turned on, the driver places the
CB voltage across the gate source of the MOSFET. This
enhances the MOSFET and turns on the topside switch.
The switch node voltage SW rises to VIN and the Boost pin
rises to VIN + INTVCC. The value of the boost capacitor CB
needs to be 100 times greater than the total input capacitance of the topside MOSFET. In most applications 0.1µF
is adequate. The reverse breakdown on DB must be greater
than VIN(MAX).
Output Voltage Programming
The output voltage is set by a resistive divider according
to the following formula:
R2
VOUT = 1.19V 1 +
R1
The external resistor divider is connected to the output as
shown in Figure 5 allowing remote voltage sensing.
Run/ Soft Start Function
The RUN/SS pin is a dual purpose pin which provides the
soft start function and a means to shut down the LTC1435.
12
Soft start reduces surge currents from VIN by gradually
increasing the internal current limit. Power supply sequencing can also be accomplished using this pin.
An internal 3µA current source charges up an external
capacitor CSS. When the voltage on RUN/SS reaches 1.3V
the LTC1435 begins operating. As the voltage on RUN/SS
continues to ramp from 1.3V to 2.4V, the internal current
limit is also ramped at a proportional linear rate. The
current limit begins at approximately 50mV/RSENSE (at
VRUN/SS = 1.3V) and ends at 150mV/RSENSE (VRUN/SS >
2.7V). The output current thus ramps up slowly, charging
the output capacitor. If RUN/SS has been pulled all the way
to ground there is a delay before starting of approximately
500ms/µF, followed by an additional 500ms/µF to reach
full current.
tDELAY = 5(10 5)CSS Seconds
Pulling the RUN/SS pin below 1.3V puts the LTC1435 into
a low quiescent current shutdown (IQ < 25µA). This pin
can be driven directly from logic as shown in Figure 6.
Diode D1 in Figure 6 reduces the start delay but allows
CSS to ramp up slowly for the soft start function; this
diode and CSS can be deleted if soft start is not needed.
The RUN/SS pin has an internal 6V Zener clamp (See
Functional Diagram).
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Foldback Current Limiting
Efficiency Considerations
As described in Power MOSFET and D1 Selection, the
worst-case dissipation for either MOSFET occurs with a
short-circuited output, when the synchronous MOSFET
conducts the current limit value almost continuously. In
most applications this will not cause excessive heating,
even for extended fault intervals. However, when heat
sinking is at a premium or higher RDS(ON) MOSFETs are
being used, foldback current limiting should be added to
reduce the current in proportion to the severity of the fault.
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Foldback current limiting is implemented by adding diode
DFB between the output and the ITH pin as shown in the
Functional Diagram. In a hard short (VOUT = 0V) the
current will be reduced to approximately 25% of the
maximum output current. This technique may be used for
all applications with regulated output voltages of 1.8V or
greater.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1435 circuits. LTC1435 VIN current, INTVCC
current, I2R losses, and topside MOSFET transition losses.
SFB Pin Operation
When the SFB pin drops below its ground referenced
1.19V threshold, continuous mode operation is forced. In
continuous mode, the large N-channel main and synchronous switches are used regardless of the load on the main
output.
In addition to providing a logic input to force continuous
synchronous operation, the SFB pin provides a means to
regulate a flyback winding output. Continuous synchronous operation allows power to be drawn from the auxiliary windings without regard to the primary output load.
The SFB pin provides a way to force continuous synchronous operation as needed by the flyback winding.
The secondary output voltage is set by the turns ratio of the
transformer in conjunction with a pair of external resistors
returned to the SFB pin as shown in Figure 4a. The
secondary regulated voltage, VSEC, in Figure 4a is given by:
R6
VSEC ≈ (N + 1)VOUT > 1.19 1 +
R5
where N is the turns ratio of the transformer and VOUT is
the main output voltage sensed by VOSENSE.
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
1. The VIN current is the DC supply current given in the
electrical characteristics which excludes MOSFET driver
and control currents. VIN current results in a small
(< 1%) loss which increases with VIN.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from INTVCC to ground. The resulting dQ/dt is a current
out of INT VCC which is typically much larger than the
control circuit current. In continuous mode,
IGATECHG = f(QT + QB), where QT and QB are the gate
charges of the topside and bottom side MOSFETs.
By powering EXTVCC from an output-derived source,
the additional VIN current resulting from the driver and
control currents will be scaled by a factor of
Duty Cycle/Efficiency. For example, in a 20V to 5V
application, 10mA of INTVCC current results in approximately 3mA of VIN current. This reduces the midcurrent
loss from 10% or more (if the driver was powered
directly from VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L and
RSENSE, but is “chopped” between the topside main
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MOSFET and the synchronous MOSFET. If the two
MOSFETs have approximately the same RDS(ON), then
the resistance of one MOSFET can simply be summed
with the resistances of L and RSENSE to obtain I2R
losses. For example, if each RDS(ON) = 0.05Ω,
RL = 0.15Ω, and RSENSE = 0.05Ω, then the total
resistance is 0.25Ω. This results in losses ranging
from 3% to 10% as the output current increases from
0.5A to 2A. I2R losses cause the efficiency to drop at
high output currents.
4. Transition losses apply only to the topside MOSFET(s),
and only when operating at high input voltages (typically 20V or greater). Transition losses can be estimated from:
Transition Loss = 2.5 (VIN)1.85(IMAX)(CRSS)(f)
Other losses, including CIN and COUT ESR dissipative
losses, Schottky conduction losses during dead-time,
and inductor core losses, generally account for less
than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, VOUT immediately shifts
by an amount equal to (∆ILOAD)(ESR), where ESR is the
effective series resistance of COUT. ∆ILOAD also begins to
charge or discharge COUT which generates a feedback
error signal. The regulator loop then acts to return VOUT to
its steady-state value. During this recovery time VOUT can
be monitored for overshoot or ringing which would indicate a stability problem. The ITH external components
shown in the Figure 1 circuit will provide adequate compensation for most applications.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator
can deliver enough current to prevent this problem if the
load switch resistance is low and it is driven quickly. The
14
only solution is to limit the rise time of the switch drive so
that the load rise time is limited to approximately
(25)(CLOAD). Thus a 10µF capacitor would require a 250µs
rise time, limiting the charging current to about 200mA.
Automotive Considerations:
Plugging into the Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation.
But before you connect, be advised: you are plugging into
the supply from hell. The main battery line in an automobile is the source of a number of nasty potential transients,
including load dump, reverse battery and double battery.
Load dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse battery is
just what it says, while double battery is a consequence of
tow truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
The network shown in Figure 7 is the most straightforward approach to protect a DC/DC converter from the
ravages of an automotive battery line. The series diode
prevents current from flowing during reverse battery,
while the transient suppressor clamps the input voltage
during load dump. Note that the transient suppressor
should not conduct during double battery operation, but
must still clamp the input voltage below breakdown of the
converter. Although the LT1435 has a maximum input
voltage of 36V, most applications will be limited to 30V
by the MOSFET BVDSS.
12V
50A IPK RATING
VIN
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
LTC1435
1435 F07
Figure 7. Automotive Application Protection
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Design Example
As a design example, assume VIN = 12V(nominal), VIN =
22V(max), VOUT = 3.3V, IMAX = 3A and f = 250kHz, RSENSE
and COSC can immediately be calculated:
RSENSE = 100mV/3A = 0.033Ω
COSC = 1.37(104)/250 – 11 = 43pF
Referring to Figure 3, a 10µH inductor falls within the
recommended range. To check the actual value of the
ripple current the following equation is used:
V
V
∆IL = OUT 1– OUT
( f)(L) VIN
The highest value of the ripple current occurs at the
maximum input voltage:
∆IL =
3.3V
3.3V
1–
= 1.12A
250kHz(10µH)
22V
The power dissipation on the topside MOSFET can be
easily estimated. Choosing a Siliconix Si4412DY results
in: RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input
voltage with T(estimated) = 50°C:
( ) [ ( )(
)](
)
1.85
+ 2.5 (22V ) (3A )(100pF )(250kHz) = 122mW
PMAIN =
3.3V 2
3 1 + 0.005 50°C − 25°C 0.042Ω
22V
The most stringent requirement for the synchronous
N-channel MOSFET occurs when VOUT = 0 (i.e. short
circuit). In this case the worst-case dissipation rises to:
(
PSYNC = ISC( AVG)
) (1+ δ ) RDS(ON)
2
highest at the maximum input voltage. The output voltage
ripple due to ESR is approximately:
VORIPPLE = RESR(∆IL) = 0.03Ω(1.112A) = 34mVP-P
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1435. These items are also illustrated graphically in
the layout diagram of Figure 8. Check the following in your
layout:
1. Are the signal and power grounds segregated? The
LTC1435 signal ground pin must return to the (–) plate
of COUT. The power ground connects to the source of
the bottom N-channel MOSFET, anode of the Schottky
diode, and (–) plate of CIN, which should have as short
lead lengths as possible.
2. Does the VOSENSE pin connect directly to the feedback
resistors? The resistive divider R1, R2 must be connected between the (+) plate of COUT and signal ground.
The 100pF capacitor should be as close as possible to
the LTC1435.
3. Are the SENSE – and SENSE + leads routed together with
minimum PC trace spacing? The filter capacitor between SENSE + and SENSE – should be as close as
possible to the LTC1435.
4. Does the (+) plate of CIN connect to the drain of the
topside MOSFET(s) as closely as possible? This capacitor provides the AC current to the MOSFET(s).
5. Is the INTVCC decoupling capacitor connected closely
between INTVCC and the power ground pin? This capacitor carries the MOSFET driver peak currents.
With the 0.033Ω sense resistor ISC(AVG) = 4A will result,
increasing the Si4412DY dissipation to 950mW at a die
temperature of 105°C.
6. Keep the switching node SW away from sensitive smallsignal nodes. Ideally the switch node should be placed
at the furthest point from the LTC1435.
CIN is chosen for an RMS current rating of at least 1.5A at
temperature. COUT is chosen with an ESR of 0.03Ω for low
output ripple. The output ripple in continuous mode will be
7. SGND should be exclusively used for grounding external components on COSC, ITH, VOSENSE and SFB pins.
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+
M1
1
CSS
2
TG
RUN/SS
BOOST
16
CIN
15
VIN
CC1
RC
COSC
+
COSC
3
CC2
4
5
ITH
SW
LTC1435
SFB
VIN
INTVCC
SGND
14
13
VOSENSE
BG
7
SENSE –
PGND
8
SENSE +
EXTVCC
D1
12
–
100pF
6
CB
0.1µF
DB
11
+
M2
4.7µF
10
1000pF
9
L1
–
R1
+
R2
COUT
VOUT
RSENSE
BOLD LINES INDICATE
HIGH CURRENT PATHS
+
LTC1435 • F08
Figure 8. LTC1435 Layout Diagram
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TYPICAL APPLICATIONS
Dual Output 5V and Synchronous 12V Application
VIN
5.4V TO 28V
COSC
68pF
+
1
CSS
0.1µF
RC
10k
CC1
470pF
CC2
51pF
2
3
4
COSC
TG
RUN/SS
BOOST
ITH
SW
SFB
LTC1435
5
VIN
7
BG
VOSENSE
SENSE
–
PGND
SENSE +
EXTVCC
IRLL014
4.7k
14
T1
10µH
1:1.42
13
0.1µF
12
+
11
4.7µF
M2
Si4412DY
MBRS140T3
+
RSENSE
0.033Ω
CSEC
3.3µF
35V
VOUT
5V/3.5A
R1
35.7k
1%
COUT
100µF
10V
×2
+
10
1000pF
8
0.01µF
15
100pF
6
M1
Si4412DY
CMDSH-3
INTVCC
SGND
16
CIN
22µF
35V
×2
9
100Ω
R2
20k
1%
SGND
100Ω
11.3k
1%
16
100k
1%
T1: DALE LPE6562-A236
LTC1435 • TA04
VOUT2
12V
120mA
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3.3V/4.5A Converter with Foldback Current Limiting
VIN
4.5V TO 28V
COSC
68pF
1
CSS
0.1µF
RC
10k
2
CC1
330pF
CC2
51pF
3
4
INTVCC
COSC
TG
RUN/SS
LTC1435
5
14
SW
SFB
ITH PIN 3
13
VIN
IN4148
7
SENSE –
PGND
SENSE +
EXTVCC
M2
Si4410DY
9
+
MBRS140T3
10
100pF
R2
20k
1%
1000pF
8
VOUT
3.3V/4.5A
R1
35.7k
1%
4.7µF
11
BG
VOSENSE
RSENSE
0.025Ω
+
100pF
6
L1
10µH
0.1µF
CMDSH-3
12
INTVCC
SGND
M1
Si4410DY
15
BOOST
ITH
CIN
22µF
35V
×2
+
16
OPTIONAL:
CONNECT TO 5V
COUT
100µF
10V
×2
SGND
(PIN 5)
LTC1435 • TA01
Dual Output 5V and 12V Application
VIN
5.4V TO 28V
COSC
68pF
+
1
CSS
0.1µF
RC
10k
CC1
510pF
CC2
51pF
2
3
4
COSC
TG
RUN/SS
BOOST
ITH
SW
SFB
LTC1435
5
VIN
15
7
BG
VOSENSE
SENSE –
PGND
SENSE +
EXTVCC
100Ω
24V
T1
10µH
1:2.2
13
+
0.1µF
CSEC
3.3µF
25V
VOUT
5V/3.5A
12
+
11
RSENSE
0.033Ω
4.7µF
M2
IRF7403
MBRS140T3
R1
35.7k
1%
COUT
100µF
10V
×2
+
10
1000pF
8
MBRS1100T3
14
100pF
6
M1
IRF7403
CMDSH-3
INTVCC
SGND
16
CIN
22µF
35V
×2
9
R2
20k
1%
SGND
100Ω
10k
90.9k
T1: DALE LPE6562-A092
LTC1435 • TA02
VOUT2
12V
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Constant-Current/Constant-Voltage High Efficiency Battery Charger
E1
VIN
+
C1*
22µF
35V
E3
GND
E3
SHDN
+
C2*
22µF
35V
1
2
3
C14
1000pF
4
5
C9
100pF
U2
LT1620
1
2
3
4
AVG
SENSE
PROG
GND
VCC
NIN
6
7
8
C15
0.1µF
IOUT
PIN
C4
0.1µF
C11
56pF
C12
0.1µF
C13
0.033µF
R5
1k
R7
1.5M
COSC
TG
RUN/SS BOOST
C5
0.1µF
16
Q1
Si4412DY
15
D1
14
U1 SW
LTC1435
13
VIN
SFB
12
SGND INTVCC
11
VOSENSE
BG
10
SENSE – PGND
9
+
SENSE EXTVCC
ITH
C3
22µF
35V
Q2
Si4412DY
+
7
C8
100pF
C7
4.7µF
16V
6
R2
1M
0.1%
5
R3
105k
0.1%
JP1A
C16
0.33µF
R6
10k
1%
C17
0.01µF
R1
0.025Ω
+
C6
0.33µF
D2
C10
100pF
8
L1
27µH
C18
0.1µF
R4
76.8k
0.1%
JP1B
DC133 F01
E5
GND
RPROG
E4
IPROG
*CONSULT CAPACITOR MANUFACTURER FOR RECOMMENDED
ESR RATING FOR CONTINUOUS 4A OPERATION
Current Programming Equation
)(R6) – 0.04
(I
IBATT = PROG
10(R1)
Efficiency
100
VIN = 24V
VBATT = 16V
95
EFFICIENCY (%)
VBATT = 12V
90
VBATT = 6V
85
80
75
0
1
3
4
2
BATTERY CHARGE CURRENT (A)
5
1435 TA05
18
E6
BATT
E7
GND
LTC1435
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
G Package
16-Lead Plastic SSOP (0.209)
(LTC DWG # 05-08-1640)
0.239 – 0.249*
(6.07 – 7.33)
16 15 14 13 12 11 10 9
0.205 – 0.212**
(5.20 – 5.38)
0.068 – 0.078
(1.73 – 1.99)
0.301 – 0.311
(7.65 – 7.90)
0° – 8°
0.0256
(0.65)
BSC
0.022 – 0.037
(0.55 – 0.95)
0.005 – 0.009
(0.13 – 0.22)
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
0.002 – 0.008
(0.05 – 0.21)
0.010 – 0.015
(0.25 – 0.38)
1 2 3 4 5 6 7 8
G16 SSOP 0795
S Package
16-Lead Plastic Small Outline
(Narrow 0.150)
(LTC DWG # 05-08-1610)
0.386 – 0.394*
(9.804 – 10.008)
16
15
14
13
12
11
10
9
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
2
3
4
5
6
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
8
0.004 – 0.010
(0.101 – 0.254)
0° – 8° TYP
0.016 – 0.050
0.406 – 1.270
7
0.050
(1.270)
TYP
S16 0695
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LTC1435
U
TYPICAL APPLICATION
Low Dropout 2.9V/3A Converter
VIN
3.5V TO 25V
COSC
68pF
1
CSS
0.1µF
RC
10k
2
CC1
330pF
CC2
51pF
INTVCC
3
4
COSC
TG
RUN/SS
BOOST
ITH
SW
SFB
LTC1435
5
SGND
VIN
16
14
13
CMDSH-3
INTVCC
7
VOSENSE
SENSE –
BG
PGND
CIN
22µF
35V
×2
+
15
L1
10µH
0.1µF
12
RSENSE
0.033Ω
VOUT
2.9V/3A
+
100pF
6
M1
1/2 Si9925DY
11
4.7µF
M2
1/2 Si9925DY
100pF
MBRS140T3
SENSE +
EXTVCC
+
10
R2
24.9k
1%
1000pF
8
R1
35.7k
1%
9
OPTIONAL:
CONNECT TO 5V
COUT
100µF
10V
×2
SGND
LTC1435 • TA03
L1: SUMIDA CDRH125-10
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20
Linear Technology Corporation
LT/GP 0896 7K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977
LINEAR TECHNOLOGY CORPORATION 1996
工商网监
湘ICP备2023018690号
