LTC1628CG#TRPBF

LTC1628/LTC1628-PG High Efficiency, 2-Phase Synchronous Step-Down Switching Regulators U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LTC®1628/LTC1628-PG are high performance dual step-down switching regulator controllers that drive all N-channel synchronous power MOSFET stages. A constant frequency current mode architecture allows adjustment of the frequency up to 300kHz. Power loss and noise due to the ESR of the input capacitors are minimized by operating the two controller output stages out of phase. Out-of-Phase Controllers Reduce Required Input Capacitance and Power Supply Induced Noise OPTI-LOOP® Compensation Minimizes COUT ±1% Output Voltage Accuracy Dual N-Channel MOSFET Synchronous Drive Power Good Output Voltage Monitor (LTC1628-PG) DC Programmed Fixed Frequency 150kHz to 300kHz Wide VIN Range: 3.5V to 36V Operation Very Low Dropout Operation: 99% Duty Cycle Adjustable Soft-Start Current Ramping Foldback Output Current Limiting Latched Short-Circuit Shutdown with Defeat Option Output Overvoltage Protection Remote Output Voltage Sense Low Shutdown IQ: 20µA 5V and 3.3V Standby Regulators Selectable Constant Frequency or Burst Mode® Operation Available in 5mm × 5mm QFN and 28-Pin SSOP Packages OPTI-LOOP compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. The precision 0.8V reference and power good output indicator are compatible with future microprocessor generations, and a wide 3.5V to 30V (36V maximum) input supply range encompasses all battery chemistries. A RUN/SS pin for each controller provides both soft-start and optional timed, short-circuit shutdown. Current foldback limits MOSFET dissipation during short-circuit conditions when overcurrent latchoff is disabled. Output overvoltage protection circuitry latches on the bottom MOSFET until VOUT returns to normal. The FCB mode pin can select among Burst Mode, constant frequency mode and continuous inductor current mode or regulate a secondary winding. The LTC1628-PG includes a power good output pin that replaces the FLTCPL, fault coupling control pin of the LTC1628. U APPLICATIO S ■ ■ ■ ■ Notebook and Palmtop Computers, PDAs Battery Chargers Portable Instruments Battery-Operated Digital Devices DC Power Distribution Systems U ■ TYPICAL APPLICATIO , LTC and LT are registered trademarks of Linear Technology Corporation. OPTI-LOOP and Burst Mode are registered trademarks of Linear Technology Corporation. + 4.7µF D3 VIN M1 L1 6.3µH CB1, 0.1µF BOOST1 BG1 BOOST2 VOUT1 5V 5A + COUT1 47µF 6V SP R1 20k 1% CC1 220pF RC1 15k L2 6.3µH CB2, 0.1µF M4 BG2 D2 PGND SENSE1+ SENSE2 + SENSE1– VOSENSE1 ITH1 SENSE2 – VOSENSE2 ITH2 1000pF R2 105k 1% VIN 5.2V TO 28V CIN 22µF 50V CERAMIC SW2 LTC1628 SGND RSENSE1 0.01Ω M3 TG2 SW1 M2 D1 D4 INTVCC TG1 1µF CERAMIC RSENSE2 0.01Ω 1000pF RUN/SS1 CSS1 0.1µF RUN/SS2 CSS2 0.1µF CC2 220pF RC2 15k R3 20k 1% M1, M2, M3, M4: FDS6680A R4 63.4k 1% VOUT2 3.3V 5A COUT 56µF 6V SP + 1628 F01 Figure 1. High Efficiency Dual 5V/3.3V Step-Down Converter 1628fb 1 LTC1628/LTC1628-PG W W W AXI U U ABSOLUTE RATI GS (Note 1) Input Supply Voltage (VIN).........................36V to – 0.3V Top Side Driver Voltages (BOOST1, BOOST2) ...................................42V to – 0.3V Switch Voltage (SW1, SW2) .........................36V to – 5V INTVCC, EXTVCC, RUN/SS1, RUN/SS2, (BOOST1-SW1), (BOOST2-SW2), PGOOD .............................7V to – 0.3V SENSE1+, SENSE2 +, SENSE1–, SENSE2 – Voltages ........................ (1.1)INTVCC to – 0.3V FREQSET, STBYMD, FCB, FLTCPL Voltage ................................... INTVCC to – 0.3V ITH1, ITH2, VOSENSE1, VOSENSE2 Voltages ...2.7V to – 0.3V Peak Output Current 2V VRUN/SS1, 2 = 0V, VSTBYMD = Open; MIN TYP MAX 1.3 UNITS mmho 3 MHz 350 125 20 35 µA µA µA 0.76 0.800 0.84 V – 0.30 – 0.18 – 0.1 µA 4.3 4.8 V 3.5 4 V 0.88 VFCB Forced Continuous Threshold IFCB Forced Continuous Pin Current VFCB = 0.85V VBINHIBIT Burst Inhibit (Constant Frequency) Threshold Measured at FCB pin UVLO Undervoltage Lockout VIN Ramping Down ● VOVL Feedback Overvoltage Lockout Measured at VOSENSE1, 2 ● 0.84 0.86 ISENSE Sense Pins Total Source Current (Each Channel); VSENSE1–, 2 – = VSENSE1+, 2+ = 0V – 85 – 60 VSTBYMD MS Master Shutdown Threshold VSTBYMD Ramping Down 0.4 VSTBYMD KA Keep-Alive Power On-Threshold VSTBYMD Ramping Up, RUNSS1, 2 = 0V DFMAX Maximum Duty Factor In Dropout 99.4 % IFLTCPL VFLTCPL Input Current LTC1628 Only 0.5V > VFLTCPL INTVCC – 0.5V < VFLTCPL < INTVCC –3 3 µA µA VFLTCPL Fault Coupling Threshold; LTC1628 Only For FCB Signal and Individual Overcurrent Faults to Affect Both Controllers 2 V IRUN/SS1, 2 Soft-Start Charge Current VRUN/SS1, 2 = 1.9V 0.5 1.2 µA VRUN/SS1, VRUN/SS2 Rising 1.0 1.5 1.9 V 4.1 4.5 V 2 4 µA VRUN/SS1, 2 ON RUN/SS Pin ON Threshold ● 98 VRUN/SS1, 2 LT RUN/SS Pin Latchoff Arming Threshold VRUN/SS1, VRUN/SS2 Rising from 3V ISCL1, 2 RUN/SS Discharge Current Soft Short Condition VOSENSE1, 2 = 0.5V; VRUN/SS1, 2 = 4.5V 0.5 5 µA 88 85 mV mV (Note 5) CLOAD = 3300pF CLOAD = 3300pF 50 50 90 90 ns ns BG Transition Time: Rise Time Fall Time (Note 5) CLOAD = 3300pF CLOAD = 3300pF 40 40 90 80 ns ns Top Gate Off to Bottom Gate On Delay Synchronous Switch-On Delay Time CLOAD = 3300pF Each Driver 90 ns Bottom Gate Off to Top Gate On Delay Top Switch-On Delay Time CLOAD = 3300pF Each Driver 90 ns Minimum On-Time Tested with a Square Wave (Note 6) 180 ns VOSENSE1, 2 = 0.5V Maximum Current Sense Threshold VOSENSE1, 2 = 0.7V,VSENSE1–, 2– = 5V ● VOSENSE1, 2 = 0.7V,VSENSE1–, 2– = 5V, LTC1628 Only TG1, 2 tr TG1, 2 tf TG Transition Time: Rise Time Fall Time BG1, 2 tr BG1, 2 tf tON(MIN) V 75 75 Shutdown Latch Disable Current VSENSE(MAX) BG/TG t2D V 2 1.6 ISDLHO TG/BG t1D 0.6 1.5 V µA 62 65 INTVCC Linear Regulator VINTVCC Internal VCC Voltage 6V < VIN < 30V, VEXTVCC = 4V 5.0 5.2 V VLDO INT INTVCC Load Regulation ICC = 0 to 20mA, VEXTVCC = 4V 4.8 0.2 1.0 % VLDO EXT EXTVCC Voltage Drop ICC = 20mA, VEXTVCC = 5V, LTC1628 120 240 mV VLDO EXT-PG EXTVCC Voltage Drop ICC = 20mA, VEXTVCC = 5V, LTC1628-PG 80 160 mV VEXTVCC EXTVCC Switchover Voltage ICC = 20mA, EXTVCC Ramping Positive VLDOHYS EXTVCC Hysteresis ● 4.5 4.7 V 0.2 V 1628fb 3 LTC1628/LTC1628-PG ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Oscillator fOSC Oscillator frequency VFREQSET = Open (Note 7) 190 220 250 kHz fLOW Lowest Frequency VFREQSET = 0V 120 140 160 kHz fHIGH Highest Frequency VFREQSET = 2.4V 280 310 360 kHz IFREQSET FREQSET Input Current VFREQSET = 0V –2 –1 µA 3.3V Linear Regulator V3.3OUT 3.3V Regulator Output Voltage No Load 3.35 3.45 V V3.3IL 3.3V Regulator Load Regulation I3.3 = 0 to 10mA 0.5 2 % V3.3VL 3.3V Regulator Line Regulation 6V < VIN < 30V 0.05 0.2 % 0.1 0.3 V ±1 µA – 9.5 9.5 % % ● 3.25 PGOOD Output (LTC1628-PG Only) VPGL PGOOD Voltage Low IPGOOD = 2mA IPGOOD PGOOD Leakage Current VPGOOD = 5V VPG PGOOD Trip Level, Either Controller VOSENSE Respect to Set Output Voltage VOSENSE Ramping Negative VOSENSE Ramping Positive Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC1628G/LTC1628G-PG: TJ = TA + (PD • 95°C/W) LTC1628CUH: TJ = TA + (PD • 34°C/W) Note 3: The LTC1628/LTC1628-PG are tested in a feedback loop that servos VITH1, 2 to a specified voltage and measures the resultant VOSENSE1, 2. –6 6 –7.5 7.5 Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 5: Rise and fall times are measured using 10% and 90% levels. Delay times are measured using 50% levels. Note 6: The minimum on-time condition is specified for an inductor peak-to-peak ripple current ≥ 40% of IMAX (see minimum on-time considerations in the Applications Information section). Note 7: VFREQSET pin internally tied to 1.19V reference through a large resistance. U W TYPICAL PERFOR A CE CHARACTERISTICS Efficiency vs Output Current (Figure 13) Efficiency vs Output Current and Mode (Figure 13) 100 100 Burst Mode OPERATION 70 60 50 40 30 FORCED CONTINUOUS MODE CONSTANT FREQUENCY (BURST DISABLE) 20 80 0.1 0.01 1 OUTPUT CURRENT (A) 10 VIN = 20V 70 1628 G01 50 0.001 VOUT = 5V IOUT = 3A 90 VIN = 10V VIN = 15V 80 70 60 60 VIN = 15V VOUT = 5V 10 0 0.001 100 VIN = 7V 90 EFFICIENCY (%) EFFICIENCY (%) 80 VIN = 15V VOUT = 5V EFFICIENCY (%) 90 Efficiency vs Input Voltage (Figure 13) 0.1 0.01 1 OUTPUT CURRENT (A) 50 10 1628 G02 5 25 15 INPUT VOLTAGE (V) 35 1628 G03 1628fb 4 LTC1628/LTC1628-PG U W TYPICAL PERFOR A CE CHARACTERISTICS Supply Current vs Input Voltage and Mode (Figure 13) 1000 600 BOTH CONTROLLERS ON 400 200 STANDBY 5.05 INTVCC AND EXTVCC SWITCH VOLTAGE (V) EXTVCC VOLTAGE DROP (mV) 250 800 SUPPLY CURRENT (µA) INTVCC and EXTVCC Switch Voltage vs Temperature EXTVCC Voltage Drop 200 150 100 50 SHUTDOWN 0 0 0 5 20 15 10 25 INPUT VOLTAGE (V) 30 35 0 10 30 20 CURRENT (mA) 40 4.85 4.80 EXTVCC SWITCHOVER THRESHOLD 4.75 4.70 – 50 – 25 50 50 25 75 0 TEMPERATURE (°C) 100 125 1628 G06 Maximum Current Sense Threshold vs Percent of Nominal Output Voltage (Foldback) 75 80 ILOAD = 1mA 70 5.0 60 4.8 4.7 50 VSENSE (mV) 4.9 VSENSE (mV) INTVCC VOLTAGE (V) 4.90 Maximum Current Sense Threshold vs Duty Factor Internal 5V LDO Line Reg 25 4.6 50 40 30 20 4.5 10 0 4.4 0 5 20 15 25 10 INPUT VOLTAGE (V) 30 0 35 20 40 60 DUTY FACTOR (%) 80 50 100 0 25 75 PERCENT ON NOMINAL OUTPUT VOLTAGE (%) 1628 G09 Maximum Current Sense Threshold vs Sense Common Mode Voltage Maximum Current Sense Threshold vs VRUN/SS (Soft-Start) 80 0 100 1628 G08 1628 G07 Current Sense Threshold vs ITH Voltage 90 80 VSENSE(CM) = 1.6V 80 70 76 40 60 VSENSE (mV) VSENSE (mV) 60 VSENSE (mV) 4.95 1628 G05 1628 G04 5.1 INTVCC VOLTAGE 5.00 72 68 50 40 30 20 10 20 0 64 –10 –20 0 0 1 2 3 4 5 6 VRUN/SS (V) 1628 G10 60 0 1 3 4 2 COMMON MODE VOLTAGE (V) 5 1628 G11 –30 0 0.5 1 1.5 VITH (V) 2 2.5 1628 G12 1628fb 5 LTC1628/LTC1628-PG U W TYPICAL PERFOR A CE CHARACTERISTICS Load Regulation FCB = 0V VIN = 15V FIGURE 1 SENSE Pins Total Source Current 100 VOSENSE = 0.7V 2.0 –0.2 50 ISENSE (µA) –0.1 VITH (V) NORMALIZED VOUT (%) VITH vs VRUN/SS 2.5 0.0 1.5 1.0 –0.3 0 –50 0.5 –0.4 0 1 0 3 2 LOAD CURRENT (A) 4 5 0 2 1 3 4 5 –100 6 2 0 VRUN/SS (V) 4 1628 G14 1628 G13 Maximum Current Sense Threshold vs Temperature 1628 G15 Dropout Voltage vs Output Current (Figure 13) 80 4 6 VSENSE COMMON MODE VOLTAGE (V) RUN/SS Current vs Temperature 1.8 VOUT = 5V 1.6 76 74 3 RUN/SS CURRENT (µA) DROPOUT VOLTAGE (V) VSENSE (mV) 78 2 RSENSE = 0.015Ω 1 72 1.4 1.2 1.0 0.8 0.6 0.4 RSENSE = 0.010Ω 0.2 70 –50 –25 0 50 25 0 75 TEMPERATURE (°C) 100 125 0 0.5 1.0 1.5 2.0 2.5 3.0 OUTPUT CURRENT (A) 3.5 4.0 0 –50 0 25 50 75 TEMPERATURE (°C) 1628 G18 1628 G17 Soft-Start Up (Figure 13) 100 125 1628 G25 Load Step (Figure 13) VOUT 5V/DIV –25 Load Step (Figure 13) VOUT 200mV/DIV VOUT 200mV/DIV IOUT 2A/DIV IOUT 2A/DIV VRUN/SS 5V/DIV IOUT 2A/DIV VIN = 15V VOUT = 5V 5ms/DIV 1628 G19 VIN = 15V 20µs/DIV VOUT = 5V LOAD STEP = 0A TO 3A Burst Mode OPERATION 1628 G20 20µs/DIV VIN = 15V VOUT = 5V LOAD STEP = 0A TO 3A CONTINUOUS MODE 1628 G21 1628fb 6 LTC1628/LTC1628-PG U W TYPICAL PERFOR A CE CHARACTERISTICS Input Source/Capacitor Instantaneous Current (Figure 13) IIN 2A/DIV VIN 200mV/DIV Constant Frequency (Burst Inhibit) Operation (Figure 13) Burst Mode Operation (Figure 13) VOUT 20mV/DIV VOUT 20mV/DIV VSW1 10V/DIV VSW2 10V/DIV IOUT 0.5A/DIV VIN = 15V 1µs/DIV VOUT = 5V IOUT5 = IOUT3.3 = 2A VIN = 15V VOUT = 5V VFCB = OPEN IOUT = 20mA 1628 G22 Current Sense Pin Input Current vs Temperature 10µs/DIV VIN = 15V VOUT = 5V VFCB = 5V IOUT = 20mA 1628 G23 EXTVCC Switch Resistance vs Temperature 35 10 1628 G24 350 31 29 27 50 25 0 75 TEMPERATURE (°C) 100 125 300 8 FREQUENCY (kHz) EXTVCC SWITCH RESISTANCE (Ω) VFREQSET = 5V 33 6 4 VFREQSET = OPEN 200 VFREQSET = 0V 150 100 50 0 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 1628 G26 125 0 – 50 – 25 50 25 75 0 TEMPERATURE (°C) 1628 G27 Undervoltage Lockout vs Temperature 100 125 1628 G28 Shutdown Latch Thresholds vs Temperature 4.5 3.50 3.45 3.40 3.35 3.30 3.25 3.20 –50 –25 250 2 SHUTDOWN LATCH THRESHOLDS (V) 25 –50 –25 2µs/DIV Oscillator Frequency vs Temperature VOUT = 5V UNDERVOLTAGE LOCKOUT (V) CURRENT SENSE INPUT CURRENT (µA) IOUT 0.5A/DIV 50 25 75 0 TEMPERATURE (°C) 100 125 1628 G29 LATCH ARMING 4.0 3.5 3.0 LATCHOFF THRESHOLD 2.5 2.0 1.5 1.0 0.5 0 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 1628 G30 1628fb 7 LTC1628/LTC1628-PG U U U PI FU CTIO S RUN/SS1, RUN/SS2: Combination of soft-start, run control inputs and short-circuit detection timers. A capacitor to ground at each of these pins sets the ramp time to full output current. Forcing either of these pins back below 1.0V causes the IC to shut down the circuitry required for that particular controller. Latchoff overcurrent protection is also invoked via this pin as described in the Applications Information section. SENSE1+, SENSE2+: The (+) Input to the Differential Current Comparators. The Ith pin voltage and controlled offsets between the SENSE– and SENSE+ pins in conjunction with RSENSE set the current trip threshold. SENSE1–, SENSE2–: The (–) Input to the Differential Current Comparators. VOSENSE1, VOSENSE2: Receives the remotely-sensed feedback voltage for each controller from an external resistive divider across the output. FREQSET: Frequency Control Input to the Oscillator. This pin can be left open, tied to ground, tied to INTVCC or driven by an external voltage source. This pin can also be used with an external phase detector to build a true phaselocked loop. STBYMD: Control pin that determines which circuitry remains active when the controllers are shut down and/or provides a common control point to shut down both controllers. See the Operation section for details. FCB: Forced Continuous Control Input. This input acts on the first controller (or both controllers depending upon the FLTCPL pin—see pin description), and is normally used to regulate a secondary winding. Pulling this pin below 0.8V will force continuous synchronous operation for the first and optionally the second controller. Do not leave this pin floating. ITH1, ITH2: Error Amplifier Output and Switching Regulator Compensation Point. Each associated channels’ current comparator trip point increases with this control voltage. SGND: Small Signal Ground common to both controllers, must be routed separately from high current grounds to the common (–) terminals of the COUT capacitors. 3.3VOUT: Output of a linear regulator capable of supplying 10mA DC with peak currents as high as 50mA. PGND: Driver Power Ground. Connects to the sources of bottom (synchronous) N-channel MOSFETs, anodes of the Schottky rectifiers and the (–) terminal(s) of CIN. INTVCC: Output of the Internal 5V Linear Low Dropout Regulator and the EXTVCC Switch. The driver and control circuits are powered from this voltage source. Must be decoupled to power ground with a minimum of 4.7µF tantalum or other low ESR capacitor. The INTVCC regulator standby function is determined by the STBYMD pin. EXTVCC: External Power Input to an Internal Switch Connected to INTVCC. This switch closes and supplies VCC power, bypassing the internal low dropout regulator, whenever EXTVCC is higher than 4.7V. See EXTVCC connection in Applications section. Do not exceed 7V on this pin. BG1, BG2: High Current Gate Drives for Bottom (Synchronous) N-Channel MOSFETs. Voltage swing at these pins is from ground to INTVCC. VIN: Main Supply Pin. A bypass capacitor should be tied between this pin and the signal ground pin. BOOST1, BOOST2: Bootstrapped Supplies to the Top Side Floating Drivers. Capacitors are connected between the boost and switch pins and Schottky diodes are tied between the boost and INTVCC pins. Voltage swing at the boost pins is from INTVCC to (VIN + INTVCC). SW1, SW2: Switch Node Connections to Inductors. Voltage swing at these pins is from a Schottky diode (external) voltage drop below ground to VIN. TG1, TG2: High Current Gate Drives for Top N-Channel MOSFETs. These are the outputs of floating drivers with a voltage swing equal to INTVCC – 0.5V superimposed on the switch node voltage SW. FLTCPL: (LTC1628 Only) Fault Coupling Control Pin that determines if fault/normal conditions on one controller will act on the other controller. FLTCPL = INTVCC to couple channels; FLTCPL = 0V to decouple. 1628fb 8 LTC1628/LTC1628-PG U U U PI FU CTIO S PGOOD: (LTC1628-PG Only) Open-Drain Logic Output. PGOOD is pulled to ground when the voltage on either VOSENSE pin is not within ±7.5% of its set point. NC: These “No Connect” pins are not tied internally to anything. On the PC layout, these pin landings should be connected to the SGND plane under the IC. W FU CTIO AL DIAGRA U U VIN INTVCC DUPLICATE FOR SECOND CONTROLLER CHANNEL 1.19V BOOST 1M FREQSET DROP OUT DET CLK1 OSCILLATOR CLK2 FLTCPL Q R Q BOT BOT – + INTVCC BINH BG COUT PGND B + 4.5V CIN SW SWITCH LOGIC + + D1 TOP ON 0.55V CB FCB RUN/SS2 3V 0.18µA R6 S TG TOP RUN/SS1 MERGE LOGIC VSEC DB – SHDN VOUT RSENSE FCB + – FCB I1 + – 3.3VOUT + 0.8V VREF – ++ SLOPE COMP VIN + 30k SENSE + 45k EXTVCC + – – EA + 5V LDO REG OV INTVCC STBYMD INTERNAL SUPPLY 6V VOSENSE 0.80V SHDN RST 4(VFB) R2 R1 0.86V ITH 1.2µA SGND VFB + – + CSEC 45k 2.4V 4.8V DSEC – 30k SENSE VIN 5V INTVCC I2 – 3mV 0.86V 4(VFB) – – + R5 RUN SOFT START CC CC2 RC RUN/SS CSS 1628 FD/F02 Figure 2 1628fb 9 LTC1628/LTC1628-PG U OPERATIO (Refer to Functional Diagram) Main Control Loop The LTC1628 uses a constant frequency, current mode step-down architecture with the two controller channels operating 180 degrees out of phase. During normal operation, each top MOSFET is turned on when the clock for that channel sets the RS latch, and turned off when the main current comparator, I1, resets the RS latch. The peak inductor current at which I1 resets the RS latch is controlled by the voltage on the ITH pin, which is the output of each error amplifier EA. The VOSENSE pin receives the voltage feedback signal, which is compared to the internal reference voltage by the EA. When the load current increases, it causes a slight decrease in VOSENSE relative to the 0.8V reference, which in turn causes the ITH voltage to increase until the average inductor current matches the new load current. After the top MOSFET has turned off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by current comparator I2, or the beginning of the next cycle. The top MOSFET drivers are biased from floating bootstrap capacitor CB, which normally is recharged during each off cycle through an external diode when the top MOSFET turns off. As VIN decreases to a voltage close to VOUT, the loop may enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector detects this and forces the top MOSFET off for about 500ns every tenth cycle to allow CB to recharge. The main control loop is shut down by pulling the RUN/SS pin low. Releasing RUN/SS allows an internal 1.2µA current source to charge soft-start capacitor CSS. When CSS reaches 1.5V, the main control loop is enabled with the ITH voltage clamped at approximately 30% of its maximum value. As CSS continues to charge, the ITH pin voltage is gradually released allowing normal, full-current operation. When both RUN/SS1 and RUN/SS2 are low, all LTC1628 controller functions are shut down, and the STBYMD pin determines if the standby 5V and 3.3V regulators are kept alive. Low Current Operation The FCB pin is a multifunction pin providing two functions: 1) to provide regulation for a secondary winding by temporarily forcing continuous PWM operation on controller 1 (or both controllers depending upon the FLTCPL pin); and 2) select between two modes of low current operation. When the FCB pin voltage is below 0.800V, the controller forces continuous PWM current mode operation. In this mode, the top and bottom MOSFETs are alternately turned on to maintain the output voltage independent of direction of inductor current. When the FCB pin is below VINTVCC␣ –␣ 2V but greater than 0.80V, the controller enters Burst Mode operation. Burst Mode operation sets a minimum output current level before inhibiting the top switch and turns off the synchronous MOSFET(s) when the inductor current goes negative. This combination of requirements will, at low currents, force the ITH pin below a voltage threshold that will temporarily inhibit turn-on of both output MOSFETs until the output voltage drops. There is 60mV of hysteresis in the burst comparator B tied to the ITH pin. This hysteresis produces output signals to the MOSFETs that turn them on for several cycles, followed by a variable “sleep” interval depending upon the load current. The resultant output voltage ripple is held to a very small value by having the hysteretic comparator after the error amplifier gain block. Constant Frequency Operation When the FCB pin is tied to INTVCC, Burst Mode operation is disabled and the forced minimum output current requirement is removed. This provides constant frequency, discontinuous (preventing reverse inductor current) current operation over the widest possible output current range. This constant frequency operation is not as efficient as Burst Mode operation, but does provide a lower noise, constant frequency operating mode down to approximately 1% of designed maximum output current. Continuous Current (PWM) Operation Tying the FCB pin to ground will force continuous current operation. This is the least efficient operating mode, but may be desirable in certain applications. The output can source or sink current in this mode. When sinking current while in forced continuous operation, current will be forced back into the main power supply potentially boosting the input supply to dangerous voltage levels— BEWARE! 1628fb 10 LTC1628/LTC1628-PG U OPERATIO (Refer to Functional Diagram) Frequency Setting Fault Coupling Pin The FREQSET pin provides frequency adjustment of the internal oscillator from approximately 140kHz to 310kHz. This input is nominally biased through an internal resistor to the 1.19V reference, setting the oscillator frequency to approximately 220kHz. This pin can be driven from an external AC or DC signal source to control the instantaneous frequency of the oscillator. The FLTCPL pin (LTC1628 only) controls two functions that can operate individually (FLTCPL = 0V) or unilaterally (FLTCPL = INTVCC) between the two controllers. When the FLTCPL pin is grounded (internally tied default mode for the LTC1628-PG), 1) the FCB input forces continuous operation only on the first controller when the applied voltage drops below 0.8V and 2) the short-circuit latchoff function only latches off the controller having the shorted output. When the FLTCPL pin is tied to INTVCC, 1) the FCB input forces continuous operation on both controllers when the applied voltage drops below 0.8V and 2) the short-circuit latchoff function latches off both controllers when either has a shorted output. INTVCC/EXTVCC Power Power for the top and bottom MOSFET drivers and most other internal circuitry is derived from the INTVCC pin. When the EXTVCC pin is left open, an internal 5V low dropout linear regulator supplies INTVCC power. If EXTVCC is taken above 4.7V, the 5V regulator is turned off and an internal switch is turned on connecting EXTVCC to INTVCC. This allows the INTVCC power to be derived from a high efficiency external source such as the output of the regulator itself or a secondary winding, as described in the Applications Information. Standby Mode Pin The STBYMD pin is a three-state input that controls common circuitry within the IC as follows: When the STBYMD pin is held at ground, both controller RUN/SS pins are pulled to ground providing a single control pin to shut down both controllers. When the pin is left open, the internal RUN/SS currents are enabled to charge the RUN/SS capacitor(s), allowing the turn-on of either controller and activating necessary common internal biasing. When the STBYMD pin is taken above 2V, both internal linear regulators are turned on independent of the state on the RUN/SS pins of the two switching regulator controllers, providing an output power source for “wake-up” circuitry. Decouple the pin with a small capacitor (0.01µF) to ground if the pin is not connected to a DC potential. Output Overvoltage Protection An overvoltage comparator, OV, guards against transient overshoots (>7.5%) as well as other more serious conditions that may overvoltage the output. In this case, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. Power Good (PGOOD) Pin The PGOOD pin (LTC1628-PG only) is connected to an open drain of an internal MOSFET. The MOSFET turns on and pulls the pin low when both the outputs are not within ±7.5% of their nominal output levels as determined by their resistive feedback dividers. When both outputs meet the ±7.5% requirement, the MOSFET is turned off within 10µs and the pin is allowed to be pulled up by an external resistor to a source of up to 7V. Foldback Current, Short-Circuit Detection and Short-Circuit Latchoff The RUN/SS capacitors are used initially to limit the inrush current of each switching regulator. After the controller has been started and been given adequate time to charge up the output capacitors and provide full load current, the RUN/SS capacitor is used in a short-circuit time-out circuit. If the output voltage falls to less than 70% of its nominal output voltage, the RUN/SS capacitor begins discharging on the assumption that the output is in an overcurrent and/or short-circuit condition. If the condition lasts for a long enough period as determined by the size of the RUN/SS capacitor, the controller (or both controllers as determined by the FLTCPL pin, LTC1628 only) will be shut down until the RUN/SS pin(s) voltage(s) are recycled. This built-in latchoff can be overridden by providing a >5µA pull-up at a compliance of 5V to the RUN/SS pin(s). This current shortens the soft start period but also prevents net discharge of the RUN/SS capacitor(s) during an 1628fb 11 LTC1628/LTC1628-PG U OPERATIO (Refer to Functional Diagram) overcurrent and/or short-circuit condition. Foldback current limiting is also activated when the output voltage falls below 70% of its nominal level whether or not the shortcircuit latchoff circuit is enabled. Even if a short is present and the short-circuit latchoff is not enabled, a safe, low output current is provided due to internal current foldback and actual power wasted is low due to the efficient nature of the current mode switching regulator. With 2-phase operation, the two channels of the dualswitching regulator are operated 180 degrees out of phase. This effectively interleaves the current pulses drawn by the switches, greatly reducing the overlap time where they add together. The result is a significant reduction in total RMS input current, which in turn allows less expensive input capacitors to be used, reduces shielding requirements for EMI and improves real world operating efficiency. THEORY AND BENEFITS OF 2-PHASE OPERATION Figure 3 compares the input waveforms for a representative single-phase dual switching regulator to the new LTC1628 2-phase dual switching regulator. An actual measurement of the RMS input current under these conditions shows that 2-phase operation dropped the input current from 2.53ARMS to 1.55ARMS. While this is an impressive reduction in itself, remember that the power losses are proportional to IRMS2, meaning that the actual power wasted is reduced by a factor of 2.66. The reduced input ripple voltage also means less power is lost in the input power path, which could include batteries, switches, trace/connector resistances and protection circuitry. Improvements in both conducted and radiated EMI also directly accrue as a result of the reduced RMS input current and voltage. The LTC1628 dual high efficiency DC/DC controller brings the considerable benefits of 2-phase operation to portable applications for the first time. Notebook computers, PDAs, handheld terminals and automotive electronics will all benefit from the lower input filtering requirement, reduced electromagnetic interference (EMI) and increased efficiency associated with 2-phase operation. Why the need for 2-phase operation? Up until the LTC1628, constant-frequency dual switching regulators operated both channels in phase (i.e., single-phase operation). This means that both switches turned on at the same time, causing current pulses of up to twice the amplitude of those for one regulator to be drawn from the input capacitor and battery. These large amplitude current pulses increased the total RMS current flowing from the input capacitor, requiring the use of more expensive input capacitors and increasing both EMI and losses in the input capacitor and battery. Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative duty cycles which, in turn, are dependent upon the input voltage VIN (Duty Cycle = VOUT/VIN). Figure 4 shows how 5V SWITCH 20V/DIV 3.3V SWITCH 20V/DIV INPUT CURRENT 5A/DIV INPUT VOLTAGE 500mV/DIV IIN(MEAS) = 2.53ARMS (a) DC236 F03a IIN(MEAS) = 1.55ARMS DC236 F03b (b) Figure 3. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching Regulators Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the LTC1628 2-Phase Regulator Allows Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency 1628fb 12 LTC1628/LTC1628-PG U OPERATIO (Refer to Functional Diagram) 3.0 2.5 INPUT RMS CURRENT (A) the input capacitor requirement to that for just one channel operating at maximum current and 50% duty cycle. SINGLE PHASE DUAL CONTROLLER the RMS input current varies for single-phase and 2-phase operation for 3.3V and 5V regulators over a wide input voltage range. A final question: If 2-phase operation offers such an advantage over single-phase operation for dual switching regulators, why hasn’t it been done before? The answer is that, while simple in concept, it is hard to implement. Constant-frequency current mode switching regulators require an oscillator derived “slope compensation” signal to allow stable operation of each regulator at over 50% duty cycle. This signal is relatively easy to derive in singlephase dual switching regulators, but required the development of a new and proprietary technique to allow 2-phase operation. In addition, isolation between the two channels becomes more critical with 2-phase operation because switch transitions in one channel could potentially disrupt the operation of the other channel. It can readily be seen that the advantages of 2-phase operation are not just limited to a narrow operating range, but in fact extend over a wide region. A good rule of thumb for most applications is that 2-phase operation will reduce The LTC1628 is proof that these hurdles have been surmounted. The new device offers unique advantages for the ever-expanding number of high efficiency power supplies required in portable electronics. 2.0 1.5 2-PHASE DUAL CONTROLLER 1.0 0.5 0 VO1 = 5V/3A VO2 = 3.3V/3A 0 10 20 30 INPUT VOLTAGE (V) 40 1628 F04 Figure 4. RMS Input Current Comparison U W U U APPLICATIO S I FOR ATIO Figure 1 on the first page is a basic LTC1628 application circuit. External component selection is driven by the load requirement, and begins with the selection of RSENSE and the inductor value. Next, the power MOSFETs and D1 are selected. Finally, CIN and COUT are selected. The circuit shown in Figure 1 can be configured for operation up to an input voltage of 28V (limited by the external MOSFETs). RSENSE Selection For Output Current RSENSE is chosen based on the required output current. The LTC1628 current comparator has a maximum threshold of 75mV/RSENSE and an input common mode range of SGND to 1.1(INTVCC). The current comparator threshold sets the peak of the inductor current, yielding a maximum average output current IMAX equal to the peak value less half the peak-to-peak ripple current, ∆IL. Allowing a margin for variations in the LTC1628 and external component values yields: RSENSE = 50mV IMAX When using the controller in very low dropout conditions, the maximum output current level will be reduced due to the internal compensation required to meet stability criterion for buck regulators operating at greater than 50% duty factor. A curve is provided to estimate this reducton in peak output current level depending upon the operating duty factor. Selection of Operating Frequency The LTC1628 uses a constant frequency architecture with the frequency determined by an internal oscillator capacitor. This internal capacitor is charged by a fixed current plus an additional current that is proportional to the voltage applied to the FREQSET pin. A graph for the voltage applied to the FREQSET pin vs frequency is given in Figure 5. As the operating frequency 1628fb 13 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO FREQSET PIN VOLTAGE (V) 2.5 25% of the current limit determined by RSENSE. Lower inductor values (higher ∆IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to decrease. 2.0 1.5 1.0 Inductor Core Selection 0.5 0 120 170 220 270 OPERATING FREQUENCY (kHz) 320 1628 F05 Figure 5. FREQSET Pin Voltage vs Frequency is increased the gate charge losses will be higher, reducing efficiency (see Efficiency Considerations). The maximum switching frequency is approximately 310kHz. Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. So why would anyone ever choose to operate at lower frequencies with larger components? The answer is efficiency. A higher frequency generally results in lower efficiency because of MOSFET gate charge losses. In addition to this basic trade-off, the effect of inductor value on ripple current and low current operation must also be considered. The inductor value has a direct effect on ripple current. The inductor ripple current ∆IL decreases with higher inductance or frequency and increases with higher VIN: ∆IL =  V  1 VOUT  1 – OUT  ( f)(L) VIN   Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy, or Kool Mµ® cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, designs for surface mount are available that do not increase the height significantly. Accepting larger values of ∆IL allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is ∆IL=0.3(IMAX). Remember, the maximum ∆IL occurs at the maximum input voltage. Power MOSFET and D1 Selection The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average inductor current required results in a peak current below The peak-to-peak drive levels are set by the INTVCC voltage. This voltage is typically 5V during start-up (see Two external power MOSFETs must be selected for each controller with the LTC1628: One N-channel MOSFET for the top (main) switch, and one N-channel MOSFET for the bottom (synchronous) switch. Kool Mµ is a registered trademark of Magnetics, Inc. 1628fb 14 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO EXTVCC Pin Connection). Consequently, logic-level threshold MOSFETs must be used in most applications. The only exception is if low input voltage is expected (VIN < 5V); then, sub-logic level threshold MOSFETs (VGS(TH) < 3V) should be used. Pay close attention to the BVDSS specification for the MOSFETs as well; most of the logic level MOSFETs are limited to 30V or less. The term (1+δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs Temperature curve, but δ = 0.005/°C can be used as an approximation for low voltage MOSFETs. CRSS is usually specified in the MOSFET characteristics. The constant k = 1.7 can be used to estimate the contributions of the two terms in the main switch dissipation equation. Selection criteria for the power MOSFETs include the “ON” resistance RDS(ON), reverse transfer capacitance CRSS, input voltage and maximum output current. When the LTC1628 is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: The Schottky diode D1 shown in Figure 1 conducts during the dead-time between the conduction of the two power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on, storing charge during the deadtime and requiring a reverse recovery period that could cost as much as 3% in efficiency at high VIN. A 1A to 3A Schottky is generally a good compromise for both regions of operation due to the relatively small average current. Larger diodes result in additional transition losses due to their larger junction capacitance. V Main Switch Duty Cycle = OUT VIN V –V Synchronous Switch Duty Cycle = IN OUT VIN The MOSFET power dissipations at maximum output current are given by: ( )( ) 2 V PMAIN = OUT IMAX 1 + δ RDS(ON) + VIN ( ) (IMAX )(CRSS )(f) k VIN 2 ( )( ) 2 V –V PSYNC = IN OUT IMAX 1 + δ RDS(ON) VIN where δ is the temperature dependency of RDS(ON) and k is a constant inversely related to the gate drive current. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For VIN < 20V the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CRSS actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during a short-circuit when the synchronous switch is on close to 100% of the period. CIN and COUT Selection The selection of CIN is simplified by the multiphase architecture and its impact on the worst-case RMS current drawn through the input network (battery/fuse/capacitor). It can be shown that the worst case RMS current occurs when only one controller is operating. The controller with the highest (VOUT)(IOUT) product needs to be used in the formula below to determine the maximum RMS current requirement. Increasing the output current, drawn from the other out-of-phase controller, will actually decrease the input RMS ripple current from this maximum value (see Figure 4). The out-of-phase technique typically reduces the input capacitor’s RMS ripple current by a factor of 30% to 70% when compared to a single phase power supply solution. The type of input capacitor, value and ESR rating have efficiency effects that need to be considered in the selection process. The capacitance value chosen should be sufficient to store adequate charge to keep high peak battery currents down. 20µF to 40µF is usually sufficient for a 25W output supply operating at 200kHz. The ESR of the capacitor is important for capacitor power dissipation as well as overall battery efficiency. All of the power (RMS ripple current • ESR) not only heats up the capacitor but wastes power from the battery. 1628fb 15 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO Medium voltage (20V to 35V) ceramic, tantalum, OS-CON and switcher-rated electrolytic capacitors can be used as input capacitors, but each has drawbacks: ceramic voltage coefficients are very high and may have audible piezoelectric effects; tantalums need to be surge-rated; OS-CONs suffer from higher inductance, larger case size and limited surface-mount applicability; electrolytics’ higher ESR and dryout possibility require several to be used. Multiphase systems allow the lowest amount of capacitance overall. As little as one 22µF or two to three 10µF ceramic capacitors are an ideal choice in a 20W to 35W power supply due to their extremely low ESR. Even though the capacitance at 20V is substantially below their rating at zero-bias, very low ESR loss makes ceramics an ideal candidate for highest efficiency battery operated systems. Also consider parallel ceramic and high quality electrolytic capacitors as an effective means of achieving ESR and bulk capacitance goals. In continuous mode, the source current of the top N-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current of one channel must be used. The maximum RMS capacitor current is given by: CIN Re quiredIRMS ≈ IMAX [V (V OUT IN − VOUT )] 1/ 2 VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. The benefit of the LTC1628 multiphase can be calculated by using the equation above for the higher power controller and then calculating the loss that would have resulted if both controller channels switch on at the same time. The total RMS power lost is lower when both controllers are operating due to the interleaving of current pulses through the input capacitor’s ESR. This is why the input capacitor’s requirement calculated above for the worst-case controller is adequate for the dual controller design. Remember that input protection fuse resistance, battery resistance and PC board trace resistance losses are also reduced due to the reduced peak currents in a multiphase system. The overall benefit of a multiphase design will only be fully realized when the source impedance of the power supply/ battery is included in the efficiency testing. The drains of the two top MOSFETS should be placed within 1cm of each other and share a common CIN(s). Separating the drains and CIN may produce undesirable voltage and current resonances at VIN. The selection of COUT is driven by the required effective series resistance (ESR). Typically once the ESR requirement is satisfied the capacitance is adequate for filtering. The output ripple (∆VOUT) is determined by:  1  ∆VOUT ≈ ∆IL  ESR +  8 fCOUT   Where f = operating frequency, COUT = output capacitance, and ∆IL= ripple current in the inductor. The output ripple is highest at maximum input voltage since ∆IL increases with input voltage. With ∆IL = 0.3IOUT(MAX) the output ripple will typically be less than 50mV at max VIN assuming: COUT Recommended ESR < 2 RSENSE and COUT > 1/(8fRSENSE) The first condition relates to the ripple current into the ESR of the output capacitance while the second term guarantees that the output capacitance does not significantly discharge during the operating frequency period due to ripple current. The choice of using smaller output capacitance increases the ripple voltage due to the discharging term but can be compensated for by using capacitors of very low ESR to maintain the ripple voltage at or below 50mV. The ITH pin OPTI-LOOP compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected. 1628fb 16 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO Manufacturers such as Nichicon, United Chemicon and Sanyo can be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest (ESR)(size) product of any aluminum electrolytic at a somewhat higher price. An additional ceramic capacitor in parallel with OS-CON capacitors is recommended to reduce the inductance effects. recommended. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers and to prevent interaction between channels. Higher input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the LTC1628 to be exceeded. The system supply current is normally dominated by the gate charge current. Additional external loading of the INTVCC and 3.3V linear regulators also needs to be taken into account for the power dissipation calculations. The total INTVCC current can be supplied by either the 5V internal linear regulator or by the EXTVCC input pin. When the voltage applied to the EXTVCC pin is less than 4.7V, all of the INTVCC current is supplied by the internal 5V linear regulator. Power dissipation for the IC in this case is highest: (VIN)(IINTVCC), and overall efficiency is lowered. The gate charge current is dependent on operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 2 of the Electrical Characteristics. For example, the LTC1628 VIN current is limited to less than 24mA from a 24V supply when not using the EXTVCC pin as follows: In surface mount applications multiple capacitors may need to be used in parallel to meet the ESR, RMS current handling and load step requirements of the application. Aluminum electrolytic, dry tantalum and special polymer capacitors are available in surface mount packages. Special polymer surface mount capacitors offer very low ESR but have lower storage capacity per unit volume than other capacitor types. These capacitors offer a very cost-effective output capacitor solution and are an ideal choice when combined with a controller having high loop bandwidth. Tantalum capacitors offer the highest capacitance density and are often used as output capacitors for switching regulators having controlled soft-start. Several excellent surge-tested choices are the AVX TPS, AVX TPSV or the KEMET T510 series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Aluminum electrolytic capacitors can be used in cost-driven applications providing that consideration is given to ripple current ratings, temperature and long term reliability. A typical application will require several to many aluminum electrolytic capacitors in parallel. A combination of the above mentioned capacitors will often result in maximizing performance and minimizing overall cost. Other capacitor types include Nichicon PL series, NEC Neocap, Pansonic SP and Sprague 595D series. Consult manufacturers for other specific recommendations. Dissipation should be calculated to also include any added current drawn from the internal 3.3V linear regulator. To prevent maximum junction temperature from being exceeded, the input supply current must be checked operating in continuous mode at maximum VIN. INTVCC Regulator EXTVCC Connection An internal P-channel low dropout regulator produces 5V at the INTVCC pin from the VIN supply pin. INTVCC powers the drivers and internal circuitry within the LTC1628. The INTVCC pin regulator can supply a peak current of 50mA and must be bypassed to ground with a minimum of 4.7µF tantalum, 10µF special polymer, or low ESR type electrolytic capacitor. A 1µF ceramic capacitor placed directly adjacent to the INTVCC and PGND IC pins is highly The LTC1628 contains an internal P-channel MOSFET switch connected between the EXTVCC and INTVCC pins. When the voltage applied to EXTVCC rises above 4.7V, the internal regulator is turned off and the switch closes, connecting the EXTVCC pin to the INTVCC pin thereby supplying internal power. The switch remains closed as long as the voltage applied to EXTVCC remains above 4.5V. This allows the MOSFET driver and control power to be TJ = 70°C + (24mA)(24V)(95°C/W) = 125°C Use of the EXTVCC input pin reduces the junction temperature to: TJ = 70°C + (24mA)(5V)(95°C/W) = 81°C 1628fb 17 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO derived from the output during normal operation (4.7V < VOUT < 7V) and from the internal regulator when the output is out of regulation (start-up, short-circuit). If more current is required through the EXTVCC switch than is specified, an external Schottky diode can be added between the EXTVCC and INTVCC pins. Do not apply greater than 7V to the EXTVCC pin and ensure that EXTVCC␣ (COUT )(VOUT) (10 – 4) (RSENSE) The minimum recommended soft-start capacitor of CSS = 0.1µF will be sufficient for most applications. Fault Conditions: Current Limit and Current Foldback The LTC1628 current comparator has a maximum sense voltage of 75mV resulting in a maximum MOSFET current of 75mV/RSENSE. The maximum value of current limit generally occurs with the largest VIN at the highest ambient temperature, conditions that cause the highest power dissipation in the top MOSFET. The LTC1628 includes current foldback to help further limit load current when the output is shorted to ground. The foldback circuit is active even when the overload shutdown latch described above is overridden. If the output falls below 70% of its nominal output level, then the maximum sense voltage is progressively lowered from 75mV to 25mV. Under short-circuit conditions with very low duty cycles, the LTC1628 will begin cycle skipping in order to limit the short-circuit current. In this situation the bottom MOSFET will be dissipating most of the power but less than in normal operation. The short-circuit ripple current is determined by the minimum on-time tON(MIN) of the LTC1628 (less than 200ns), the input voltage and inductor value: ∆IL(SC) = tON(MIN) (VIN/L) The resulting short-circuit current is: ISC = 25mV 1 + ∆IL(SC) RSENSE 2 1628fb 20 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO Fault Conditions: Overvoltage Protection (Crowbar) The overvoltage crowbar is designed to blow a system input fuse when the output voltage of the regulator rises much higher than nominal levels. The crowbar causes huge currents to flow, that blow the fuse to protect against a shorted top MOSFET if the short occurs while the controller is operating. A comparator monitors the output for overvoltage conditions. The comparator (OV) detects overvoltage faults greater than 7.5% above the nominal output voltage. When this condition is sensed, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. The output of this comparator is only latched by the overvoltage condition itself and will therefore allow a switching regulator system having a poor PC layout to function while the design is being debugged. The bottom MOSFET remains on continuously for as long as the OV condition persists; if VOUT returns to a safe level, normal operation automatically resumes. A shorted top MOSFET will result in a high current condition which will open the system fuse. The switching regulator will regulate properly with a leaky top MOSFET by altering the duty cycle to accommodate the leakage. The Standby Mode (STBYMD) Pin Function The Standby Mode (STBYMD) pin provides several choices for start-up and standby operational modes. If the pin is pulled to ground, the RUN/SS pins for both controllers are internally pulled to ground, preventing start-up and thereby providing a single control pin for turning off both controllers at once. If the pin is left open or decoupled with a capacitor to ground, the RUN/SS pins are each internally provided with a starting current enabling external control for turning on each controller independently. If the pin is provided with a current of >3µA at a voltage greater than 2V, both internal linear regulators (INTVCC and 3.3V) will be on even when both controllers are shut down. In this mode, the onboard 3.3V and 5V linear regulators can provide power to keep-alive functions such as a keyboard controller. This pin can also be used as a latching “on” and/ or latching “off” power switch if so designed. Frequency of Operation The LTC1628 has an internal voltage controlled oscillator. The frequency of this oscillator can be varied over a 2 to 1 range. The pin is internally self-biased at 1.19V, resulting in a free-running frequency of approximately 220kHz. The FREQSET pin can be grounded to lower this frequency to approximately 140kHz or tied to the INTVCC pin to yield approximately 310kHz. The FREQSET pin may be driven with a voltage from 0 to INTVCC to fix or modulate the oscillator frequency as shown in Figure 5. Minimum On-Time Considerations Minimum on-time tON(MIN) is the smallest time duration that the LTC1628 is capable of turning on the top MOSFET. It is determined by internal timing delays and the gate charge required to turn on the top MOSFET. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that tON(MIN) < VOUT VIN( f) If the duty cycle falls below what can be accommodated by the minimum on-time, the LTC1628 will begin to skip cycles. The output voltage will continue to be regulated, but the ripple voltage and current will increase. The minimum on-time for the LTC1628 is generally less than 200ns. However, as the peak sense voltage decreases the minimum on-time gradually increases up to about 300ns. This is of particular concern in forced continuous applications with low ripple current at light loads. If the duty cycle drops below the minimum on-time limit in this situation, a significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple. 1628fb 21 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO FCB Pin Operation The FCB pin can be used to regulate a secondary winding or as a logic level input. Continuous operation is forced when the FCB pin drops below 0.8V. During continuous mode, current flows continuously in the transformer primary. The secondary winding(s) draw current only when the bottom, synchronous switch is on. When primary load currents are low and/or the VIN/VOUT ratio is low, the synchronous switch may not be on for a sufficient amount of time to transfer power from the output capacitor to the secondary load. Forced continuous operation will support secondary windings providing there is sufficient synchronous switch duty factor. Thus, the FCB input pin removes the requirement that power must be drawn from the inductor primary in order to extract power from the auxiliary windings. With the loop in continuous mode, the auxiliary outputs may nominally be loaded without regard to the primary output load. The secondary output voltage VSEC is normally set as shown in Figure 6a by the turns ratio N of the transformer: VSEC ≅ (N + 1) VOUT However, if the controller goes into Burst Mode operation and halts switching due to a light primary load current, then VSEC will droop. An external resistive divider from VSEC to the FCB pin sets a minimum voltage VSEC(MIN):  R6  VSEC(MIN) ≈ 0.8 V 1 +   R5  If VSEC drops below this level, the FCB voltage forces temporary continuous switching operation until VSEC is again above its minimum. In order to prevent erratic operation if no external connections are made to the FCB pin, the FCB pin has a 0.18µA internal current source pulling the pin high. Include this current when choosing resistor values R5 and R6. The following table summarizes the possible states available on the FCB pin: Table 1 FCB Pin Condition 0V to 0.75V Forced Continuous (Current Reversal Allowed—Burst Inhibited) 0.85V < VFCB < 4.3V Minimum Peak Current Induces Burst Mode Operation No Current Reversal Allowed Feedback Resistors Regulating a Secondary Winding >4.8V Burst Mode Operation Disabled Constant Frequency Mode Enabled No Current Reversal Allowed No Minimum Peak Current The FLTCPL pin determines whether only the first or both controllers are temporarily forced into continuous mode when the FCB pin falls below 0.8V. Tying the FLTCPL pin to ground will send only the first controller into continuous operation while tying the FLTCPL pin to INTVCC will send both controllers into continuous operation. Voltage Positioning Voltage positioning can be used to minimize peak-to-peak output voltage excursions under worst-case transient loading conditions. The open-loop DC gain of the control loop is reduced depending upon the maximum load step specifications. Voltage positioning can easily be added to the LTC1628 by loading the ITH pin with a resistive divider having a Thevenin equivalent voltage source equal to the midpoint operating voltage of the error amplifier, or 1.2V (see Figure 8). The resistive load reduces the DC loop gain while maintaining the linear control range of the error amplifier. The maximum output voltage deviation can theoretically be INTVCC RT2 ITH RT1 RC LTC1628 CC 1628 F08 Figure 8. Active Voltage Positioning Applied to the LTC1628 1628fb 22 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO reduced to half or alternatively the amount of output capacitance can be reduced for a particular application. A complete explanation is included in Design Solutions 10. (See www.linear.com.) Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1628 circuits: 1) LTC1628 VIN current (including loading on the 3.3V internal regulator), 2) INTVCC regulator current, 3) I2R losses, 4) Topside MOSFET transition losses. 1. The VIN current has two components: the first is the DC supply current given in the Electrical Characteristics table, which excludes MOSFET driver and control currents; the second is the current drawn from the 3.3V linear regulator output. VIN current typically results in a small (1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can alter its delivery of current quickly enough to prevent this sudden step change in output voltage if the load switch resistance is low and it is driven quickly. If the ratio of CLOAD to COUT is greater than1:50, the switch rise time should be controlled so that the load rise time is limited to approximately 25 • CLOAD. Thus a 10µF capacitor would require a 250µs rise time, limiting the charging current to about 200mA. 1628fb 24 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO Automotive Considerations: Plugging into the Cigarette Lighter As battery-powered devices go mobile, there is a natural interest in plugging into the cigarette lighter in order to conserve or even recharge battery packs during operation. But before you connect, be advised: you are plugging into the supply from hell. The main power line in an automobile is the source of a number of nasty potential transients, including load-dump, reverse-battery, and double-battery. Load-dump is the result of a loose battery cable. When the cable breaks connection, the field collapse in the alternator can cause a positive spike as high as 60V which takes several hundred milliseconds to decay. Reverse-battery is 50A IPK RATING 12V just what it says, while double-battery is a consequence of tow-truck operators finding that a 24V jump start cranks cold engines faster than 12V. The network shown in Figure 9 is the most straight forward approach to protect a DC/DC converter from the ravages of an automotive power line. The series diode prevents current from flowing during reverse-battery, while the transient suppressor clamps the input voltage during load-dump. Note that the transient suppressor should not conduct during double-battery operation, but must still clamp the input voltage below breakdown of the converter. Although the LTC1628 has a maximum input voltage of 36V, most applications will be limited to 30V by the MOSFET BVDSS. VIN LTC1628 TRANSIENT VOLTAGE SUPPRESSOR GENERAL INSTRUMENT 1.5KA24A 1628 F09 Figure 9. Automotive Application Protection 1628fb 25 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO Design Example As a design example for one channel, assume VIN = 12V(nominal), VIN = 22V(max), VOUT = 1.8V, IMAX = 5A, and f = 300kHz. The inductance value is chosen first based on a 30% ripple current assumption. The highest value of ripple current occurs at the maximum input voltage. Tie the FREQSET pin to the INTVCC pin for 300kHz operation. The minimum inductance for 30% ripple current is: Choosing 1% resistors; R1 = 25.5k and R2 = 32.4k yields an output voltage of 1.816V. The power dissipation on the top side MOSFET can be easily estimated. Choosing a Siliconix Si4412DY results in; RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input voltage with T(estimated) = 50°C: ()[ ] 2 (0.042Ω) + 1.7(22V) (5A)(100pF)(300kHz) PMAIN =  V  V ∆IL = OUT  1 – OUT  ( f)(L)  VIN  A 4.7µH inductor will produce 23% ripple current and a 3.3µH will result in 33%. The peak inductor current will be the maximum DC value plus one half the ripple current, or 5.84A, for the 3.3µH value. Increasing the ripple current will also help ensure that the minimum on-time of 200ns is not violated. The minimum on-time occurs at maximum VIN: tON(MIN) = VOUT VIN(MAX)f = 1.8 V = 273ns 22V(300kHz) The RSENSE resistor value can be calculated by using the maximum current sense voltage specification with some accommodation for tolerances: 50mV RSENSE ≤ ≈ 0.01Ω 5.84A Since the output voltage is below 2.4V the output resistive divider will need to be sized to not only set the output voltage but also to absorb the SENSE pins specified input current. 1.8 V 2 5 1 + (0.005)(50°C – 25°C) 22V = 220mW A short-circuit to ground will result in a folded back current of: ISC = 25mV 1  200ns(22V) +   = 3.2A 0.01Ω 2  3.3µH  with a typical value of RDS(ON) and δ = (0.005/°C)(20) = 0.1. The resulting power dissipated in the bottom MOSFET is: ( ) (1.1)(0.042Ω) 22V – 1.8 V 3.2A 22V = 434mW PSYNC = 2 which is less than under full-load conditions. CIN is chosen for an RMS current rating of at least 3A at temperature assuming only this channel is on. COUT is chosen with an ESR of 0.02Ω for low output ripple. The output ripple in continuous mode will be highest at the maximum input voltage. The output voltage ripple due to ESR is approximately: VORIPPLE = RESR(∆IL) = 0.02Ω(1.67A) = 33mVP–P   0.8 V R1(MAX ) = 24k    2.4V – VOUT   0.8 V  = 24K  = 32k  2.4V – 1.8 V  1628fb 26 LTC1628/LTC1628-PG U W U U APPLICATIO S I FOR ATIO PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1628. These items are also illustrated graphically in the layout diagram of Figure 10. The Figure 11 illustrates the current waveforms present in the various branches of the 2-phase synchronous regulators operating in the continuous mode. Check the following in your layout: 1. Are the top N-channel MOSFETs M1 and M3 located within 1cm of each other with a common drain connection at CIN? Do not attempt to split the input decoupling for the two channels as it can cause a large resonant loop. FLTCPL 3 R2 4 R1 5 6 8 3.3V 10 11 12 R3 R4 13 14 SW1 VOSENSE1 BOOST1 FREQSET VIN STBYMD BG1 FCB EXTVCC LTC1628 ITH1 SGND 3.3VOUT ITH2 INTVCC PGND BG2 BOOST2 VOSENSE2 SW2 SENSE2 – TG2 SENSE2 + RUN/SS2 PGOOD L1 RSENSE VOUT1 26 25 CB1 M1 M2 D1 24 23 COUT1 RIN 22 CIN CVIN 21 20 CINTVCC GND + 9 SENSE1 – VPULL-UP (