LTC3405
1.5MHz, 300mA
Synchronous Step-Down
Regulator in ThinSOT
U
FEATURES
DESCRIPTIO
■
The LTC ®3405 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current
mode architecture. Supply current during operation is
only 20µA and drops to
1.5V). In this mode, the efficiency is lower at light loads,
but becomes comparable to Burst Mode operation when
the output load exceeds 25mA. The advantage of pulse
skipping mode is lower output ripple and less interference
to audio circuitry.
500
VOUT = 1.3V
400
VOUT = 2.5V
300
200
100
0
2.5
3.0
3.5
4.0
4.5
SUPPLY VOLTAGE (V)
5.0
5.5
3405 G23
Short-Circuit Protection
When the output is shorted to ground, the frequency of the
oscillator is reduced to about 210kHz, 1/7 the nominal
frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing
runaway. The oscillator’s frequency will progressively
increase to 1.5MHz when VFB rises above 0V.
Dropout Operation
As the input supply voltage decreases to a value approaching the output voltage, the duty cycle increases toward the
maximum on-time. Further reduction of the supply voltage
forces the main switch to remain on for more than one cycle
Figure 2. Maximum Output Current vs Input Voltage
Slope Compensation and Inductor Peak Current
Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by
adding a compensating ramp to the inductor current
signal at duty cycles in excess of 40%. Normally, this
results in a reduction of maximum inductor peak current
for duty cycles > 40%. However, the LTC3405 uses a
patent-pending scheme that counteracts this compensating ramp, which allows the maximum inductor peak
current to remain unaffected throughout all duty cycles.
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The basic LTC3405 application circuit is shown in Figure 1.
External component selection is driven by the load requirement and begins with the selection of L followed by CIN and
COUT.
Table 1. Representative Surface Mount Inductors
MANUFACTURER PART NUMBER
MAX DC
VALUE CURRENT DCR HEIGHT
Taiyo Yuden
LB2016T3R3M
3.3µH
280mA
0.2Ω 1.6mm
Panasonic
ELT5KT4R7M
4.7µH
950mA
0.2Ω 1.2mm
Inductor Selection
Murata
LQH3C4R7M34
4.7µH
450mA
0.2Ω
For most applications, the value of the inductor will fall in
the range of 3.3µH to 10µH. Its value is chosen based on
the desired ripple current. Large value inductors lower
ripple current and small value inductors result in higher
ripple currents. Higher VIN or VOUT also increases the ripple
current as shown in equation 1. A reasonable starting point
for setting ripple current is ∆IL = 120mA (40% of 300mA).
Taiyo Yuden
LB2016T4R7M
4.7µH
210mA
0.25Ω 1.6mm
Panasonic
ELT5KT6R8M
6.8µH
760mA
0.3Ω 1.2mm
Panasonic
ELT5KT100M
10µH
680mA
0.36Ω 1.2mm
Sumida
CMD4D116R8MC 6.8µH
620mA
0.23Ω 1.2mm
∆IL =
⎛ V ⎞
1
VOUT ⎜ 1 − OUT ⎟
( f)(L) ⎝ VIN ⎠
(1)
The DC current rating of the inductor should be at least
equal to the maximum load current plus half the ripple
current to prevent core saturation. Thus, a 360mA rated
inductor should be enough for most applications (300mA
+ 60mA). For better efficiency, choose a low DC-resistance
inductor.
The inductor value also has an effect on Burst Mode
operation. The transition to low current operation begins
when the inductor current peaks fall to approximately
100mA. Lower inductor values (higher ∆IL) will cause this
to occur at lower load currents, which can cause a dip in
efficiency in the upper range of low current operation. In
Burst Mode operation, lower inductance values will cause
the burst frequency to increase.
Inductor Core Selection
Different core materials and shapes will change the size/
current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials
are small and don’t radiate much energy, but generally cost
more than powdered iron core inductors with similar
electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on
what the LTC3405 requires to operate. Table 1 shows some
typical surface mount inductors that work well in LTC3405
applications.
8
2mm
CIN and COUT Selection
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle VOUT/VIN. To prevent large
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum
RMS capacitor current is given by:
1/ 2
VOUT (VIN − VOUT )]
[
CIN required IRMS ≅ IOMAX
VIN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations
do not offer much relief. Note that the capacitor
manufacturer’s ripple current ratings are often based on
2000 hours of life. This makes it advisable to further derate
the capacitor, or choose a capacitor rated at a higher
temperature than required. Always consult the manufacturer if there is any question.
The selection of COUT is driven by the required effective
series resistance (ESR). An ESR in the range of 100mΩ to
200mΩ is necessary to provide a stable loop. For the
LTC3405, the general rule for proper operation is:
0.1Ω ≤ COUT required ESR ≤ 0.6Ω
ESR is a direct function of the volume of the capacitor; that
is, physically larger capacitors have lower ESR. Once the
ESR requirement for COUT has been met, the RMS current
rating generally far exceeds the IRIPPLE(P-P) requirement.
The output ripple ∆VOUT is determined by:
⎛
1 ⎞
∆VOUT ≅ ∆IL ⎜ ESR +
⎟
⎝
8fC OUT ⎠
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where f = operating frequency, COUT = output capacitance
and ∆IL = ripple current in the inductor. For a fixed output
voltage, the output ripple is highest at maximum input
voltage since ∆IL increases with input voltage.
Aluminum electrolytic and dry tantalum capacitors are
both available in surface mount configurations. In the case
of tantalum, it is critical that the capacitors are surge tested
for use in switching power supplies. An excellent choice is
the AVX TPS series of surface mount tantalum. These are
specially constructed and tested for low ESR so they give
the lowest ESR for a given volume. Other capacitor types
include Sanyo POSCAP, Kemet T510 and T495 series, and
Sprague 593D and 595D series. Consult the manufacturer
for other specific recommendations.
Another solution is to connect the feedback resistor to the
SW pin as shown in Figure 4. Taking the feedback information at the SW pin removes the phase lag due to the output
capacitor resulting in a very stable loop. This configuration
lowers the load regulation by the DC resistance of the
inductor multiplied by the load current. This slight shift in
load regulation actually helps reduce the overshoot and
undershoot of the output voltage during a load transient.
VIN
2.7V
TO 4.2V
4
CIN
2.2µF
CER
VIN
SW
LTC3405
1
6
4.7µH
3
887k
22pF
RUN
VFB
MODE
5
GND
VOUT
1.5V
COUT
4.7µF
CER
1M
2
3405 F04
Using Ceramic Input and Output Capacitors
Figure 4. Using All Ceramic Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. However, care must
be taken when these capacitors are used at the input and
the output. When a ceramic capacitor is used at the input
and the power is supplied by a wall adapter through long
wires, a load step at the output can induce ringing at the
input, VIN. At best, this ringing can couple to the output and
be mistaken as loop instability. At worst, a sudden inrush
of current through the long wires can potentially cause a
voltage spike at VIN, large enough to damage the part.
When ceramic capacitors are used at the output, their low
ESR cannot provide sufficient phase lag cancellation to
stabilize the loop. One solution is to use a tantalum
capacitor, with its higher ESR, to provide the bulk capacitance and parallel it with a small ceramic capacitor to
reduce the ripple voltage as shown in Figure 3.
VIN
2.7V
TO 4.2V
4
CIN
2.2µF
CER
VIN
SW
3
22pF
LTC3405
1
6
4.7µH
RUN
VFB
MODE
GND
2
5
COUT1 +
1µF
CER
VOUT
1.5V
COUT2
22µF
TANT
887k
1M
A third solution is to use a high value resistor to inject a
feedforward signal at VFB mimicking the ripple voltage of
a high ESR output capacitor. The circuit in Figure 5 shows
how this technique can be easily realized. The feedforward
resistor, R2B, is connected to SW as in the previous
example. However, in this case, the feedback information
is taken from the resistive divider, R2A and R1, at the
output. This eliminates most of the load regulation degradation due to the DC resistance of the inductor while
providing a stable operation similar to that obtained from
a high ESR tantalum type capacitor. Using this technique,
the extra feedforward resistor, R2B, must be accounted
for when calculating the resistive divider as follows:
R2A • R2B
R2A + R2B
⎛ R2⎞
= 0.8V ⎜ 1 + ⎟
⎝ R1⎠
R2 = R2A || R2B =
VOUT
VIN
2.7V
TO 4.2V
4
CIN
2.2µF
CER
VIN
SW
3
R2B 22pF
1M
LTC3405
1
6
4.7µH
RUN
VFB
MODE
GND
2
5
R2A
R1 215k
200k
3405 F03
Figure 3. Paralleling a Ceramic with a Tantalum Capacitor
VOUT
1.5V
COUT1
4.7µF
CER
3405 F05
Figure 5. Feedforward Injection in an
All Ceramic Capacitor Application
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In pulse skipping mode, the LTC3405 is stable with a 4.7µF
ceramic output capacitor with VIN ≤ 4.2V. For single Li-Ion
applications operating in pulse skipping mode, the circuit
shown in Figure 6 can be used
When choosing the input and output ceramic capacitors,
choose the X5R or X7R dielectric formulations. These
dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size.
4
CIN
2.2µF
CER
VIN
SW
3
6
VOUT
1.5V
22pF
LTC3405
1
4.7µH
RUN
VFB
MODE
GND
2
1
COUT1
4.7µF
CER
5
VIN = 3.6V
0.1
887k
1M
3405 F06
Figure 6. Using All Ceramic Capacitors in Pulse Skipping Mode
POWER LOST (W)
VIN
2.7V
TO 4.2V
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of the
losses in LTC3405 circuits: VIN quiescent current and I2R
losses. The VIN quiescent current loss dominates the
efficiency loss at very low load currents whereas the I2R
loss dominates the efficiency loss at medium to high load
currents. In a typical efficiency plot, the efficiency curve at
very low load currents can be misleading since the actual
power lost is of no consequence as illustrated in Figure 8.
VOUT = 1.8V
0.01
0.001
Output Voltage Programming
VOUT = 1.3V
The output voltage is set by a resistive divider according
to the following formula:
VOUT
0.0001
0.1
1
100
10
LOAD CURRENT (mA)
1000
3405 F08
⎛ R2⎞
= 0.8V ⎜ 1 + ⎟
⎝ R1⎠
(2)
The external resistive divider is connected to the output,
allowing remote voltage sensing as shown in Figure 7.
0.8V ≤ VOUT ≤ 5.5V
R2
VFB
LTC3405
VOUT = 3.3V
VOUT = 2.5V
R1
GND
3405 F07
Figure 7. Setting the LTC3405 Output Voltage
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Figure 8. Power Lost vs Load Current
1. The VIN quiescent current is due to two components:
the DC bias current as given in the electrical characteristics and the internal main switch and synchronous
switch gate charge currents. The gate charge current
results from switching the gate capacitance of the
internal power MOSFET switches. Each time the gate is
switched from high to low to high again, a packet of
charge, dQ, moves from VIN to ground. The resulting
dQ/dt is the current out of VIN that is typically larger than
the DC bias current. In continuous mode, IGATECHG =
f(QT + QB) where QT and QB are the gate charges of the
internal top and bottom switches. Both the DC bias and
gate charge losses are proportional to VIN and thus
their effects will be more pronounced at higher supply
voltages.
2. I2R losses are calculated from the resistances of the
internal switches, RSW, and external inductor RL. In
continuous mode, the average output current flowing
through inductor L is “chopped” between the main
switch and the synchronous switch. Thus, the series
resistance looking into the SW pin is a function of both
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top and bottom MOSFET RDS(ON) and the duty cycle
(DC) as follows:
RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Charateristics
curves. Thus, to obtain I2R losses, simply add RSW to
RL and multiply the result by the square of the average
output current.
ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the
P-channel switch at 70°C is approximately 0.94Ω. Therefore, power dissipated by the part is:
PD = ILOAD2 • RDS(ON) = 84.6mW
For the SOT-23 package, the θJA is 250°C/ W. Thus, the
junction temperature of the regulator is:
TJ = 70°C + (0.0846)(250) = 91.15°C
Other losses including CIN and COUT ESR dissipative
losses and inductor core losses generally account for less
than 2% total additional loss.
which is well below the maximum junction temperature of
125°C.
Thermal Considerations
Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)).
In most applications the LTC3405 does not dissipate
much heat due to its high efficiency. But, in applications
where the LTC3405 is running at high ambient temperature with low supply voltage and high duty cycles, such
as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction
temperature reaches approximately 150°C, both power
switches will be turned off and the SW node will become
high impedance.
To avoid the LTC3405 from exceeding the maximum
junction temperature, the user will need to do a thermal
analysis. The goal of the thermal analysis is to determine
whether the operating conditions exceed the maximum
junction temperature of the part. The temperature rise is
given by:
TR = (PD)(θJA)
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to the
ambient temperature.
The junction temperature, TJ, is given by:
T J = TA + TR
where TA is the ambient temperature.
As an example, consider the LTC3405 in dropout at an
input voltage of 2.7V, a load current of 300mA and an
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to (∆ILOAD • ESR), where ESR is the effective series
resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT, which generates a feedback error signal.
The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability
problem. For a detailed explanation of switching control
loop theory, see Application Note 76.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25 • CLOAD).
Thus, a 10µF capacitor charging to 3.3V would require a
250µs rise time, limiting the charging current to about
130mA.
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PC Board Layout Checklist
4. Keep the switching node, SW, away from the sensitive
VFB node.
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3405. These items are also illustrated graphically in
Figures 9 and 10. Check the following in your layout:
Design Example
As a design example, assume the LTC3405 is used in a
single lithium-ion battery-powered cellular phone
application. The VIN will be operating from a maximum of
4.2V down to about 2.7V. The load current requirement
is a maximum of 0.25A but most of the time it will be in
standby mode, requiring only 2mA. Efficiency at both low
and high load currents is important. Output voltage is
2.5V. With this information we can calculate L using
equation (1),
1. The power traces, consisting of the GND trace, the SW
trace and the VIN trace should be kept short, direct and
wide.
2. Does the VFB pin connect directly to the feedback
resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground.
3. Does the (+) plate of CIN connect to VIN as closely as
possible? This capacitor provides the AC current to the
internal power MOSFETs.
1
L=
RUN
MODE
⎛ V ⎞
1
VOUT ⎜ 1 − OUT ⎟
( f)(∆IL ) ⎝ VIN ⎠
(3)
6
LTC3405
2
GND
–
VFB
5
COUT
VOUT
+
R2
3
L1
SW
VIN
R1
4
CFWD
CIN
R3*
+
VIN
–
3405 F09
BOLD LINES INDICATE HIGH CURRENT PATHS
*ADD R3 FOR APPLICATIONS USING A CERAMIC COUT
Figure 9. LTC3405 Layout Diagram
VIA TO SW NODE
R3*
VOUT
VIA TO GND
VFB
R1
VIN
VIA TO VIN
VIA TO VOUT
R2
PIN 1
L1
CFWD
LTC3405
SW
COUT
CIN
GND
*ADD R3 WHEN USING CERAMIC COUT
3405 F10
Figure 10. LTC3405 Suggested Layout
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Substituting VOUT = 2.5V, VIN = 4.2V, ∆IL = 100mA and
f = 1.5MHz in equation (3) gives:
L=
For the feedback resistors, choose R1 = 412k. R2 can
then be calculated from equation (2) to be:
⎛V
⎞
R2 = ⎜ OUT − 1⎟ R1 = 875.5k; use 887k
⎝ 0.8
⎠
2.5V
⎛ 2.5V ⎞
⎜1 −
⎟ ≅ 6.8µH
1.5MHz(100mA) ⎝ 4.2V ⎠
Figure 11 shows the complete circuit along with its
efficiency curve.
For best efficiency choose a 300mA or greater inductor
with less than 0.3Ω series resistance.
100
CIN will require an RMS current rating of at least 0.125A ≅
ILOAD(MAX)/2 at temperature and COUT will require an ESR
of less than 0.6Ω and greater than 0.1Ω. In most cases,
a tantalum capacitor will satisfy this requirement.
4
†
CIN
2.2µF
CER
VIN
SW
6.8µH*
3
6
VOUT
2.5V
22pF
LTC3405
1
+
VFB
GND
2
5
3405 F11a
60
40
887k
412k
VIN = 4.2V
70
50
COUT**
33µF
TANT
RUN
MODE
VIN = 3.6V
80
EFFICIENCY (%)
VIN
2.7V
TO 4.2V
VIN = 2.7V
90
30
0.1
*SUMIDA CMD4D11-6R8MC
** AVX TPSB336K006R0600
†
TAIYO YUDEN LMK212BJ225MG
1
100
10
OUTPUT CURRENT (mA)
1000
3405 F11b
Figure 11a
Figure 11b
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TYPICAL APPLICATIO S
Single Li-Ion to 1.8V/300mA Regulator
Optimized for Small Footprint and High Efficiency
VIN
2.7V
TO 4.2V
4
CIN**
1µF
CER
VIN
SW
LTC3405
1
6
4.7µH*
3
1M
RUN
VFB
MODE
GND
2
5
22pF
VOUT
1.8V
COUT†
4.7µF
CER
332k
*MURATA LQH3C4R7M34
**TAIYO YUDEN CERAMIC JMK107BJ105MA
†
3405 TA01a
TAIYO YUDEN CERAMIC JMK212BJ475MG
200k
100
VIN = 2.7V
90
VIN = 3.6V
80
EFFICIENCY (%)
VOUT
100mV/DIV
AC COUPLED
70
60
IL
200mA/DIV
VIN = 4.2V
50
ILOAD
200mA/DIV
40
30
0.1
1
100
10
OUTPUT CURRENT (mA)
1000
3405 TA01b
VIN = 3.6V
40µs/DIV
ILOAD = 100mA TO 250mA
3405 TA01c
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LTC3405
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TYPICAL APPLICATIO S
Single Li-Ion to 1.8V/300mA Regulator
Using Ceramic and Tantalum Output Capacitors
VIN
2.7V
TO 4.2V
4
CIN**
2.2µF
CER
VIN
SW
4.7µH*
3
22pF
LTC3405
1
RUN
6
MODE
VFB
5
COUT1*** +
1µF
CER
VOUT
1.8V
COUT2†
22µF
TANT
887k
*MURATA LQH3C4R7M34
**TAIYO YUDEN CERAMIC LMK212BJ225MG
***TAIYO YUDEN CERAMIC JMK107BJ105MA
†
AVX TAJA226M006R
3405 TA02a
GND
698k
2
100
VIN = 2.7V
VOUT
100mV/DIV
AC COUPLED
90
VIN = 4.2V
EFFICIENCY (%)
80
VIN = 3.6V
70
IL
200mA/DIV
60
50
ILOAD
200mA/DIV
40
30
0.1
VIN = 3.6V
40µs/DIV
VOUT = 1.8V
ILOAD = 100mA TO 250mA
1000
1
100
10
OUTPUT CURRENT (mA)
3405 TA02b
3405 TA02c
Single Li-Ion to 1.8V/200mA Regulator
Using All Ceramic Capacitors Optimized for Smallest Footprint
VIN
2.7V
TO 4.2V
4
CIN**
1µF
CER
VIN
SW
LTC3405
1
6
1M
RUN
VFB
MODE
GND
2
3.3µH*
3
5
22pF
VOUT
1.8V
COUT†
4.7µF
CER
332k
200k *TAIYO YUDEN LB2016T3R3M
**TAIYO YUDEN CERAMIC JMK107BJ105MA
†
3405 TA03a TAIYO YUDEN CERAMIC JMK212BJ475MG
100
VOUT
100mV/DIV
AC COUPLED
VIN = 2.7V
90
EFFICIENCY (%)
80
VIN = 3.6V
70
60
IL
200mA/DIV
VIN = 4.2V
50
ILOAD
200mA/DIV
40
30
0.1
1
100
10
OUTPUT CURRENT (mA)
1000
3405 TA03b
VIN = 3.6V
40µs/DIV
ILOAD = 100mA TO 250mA
3405 TA03c
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LTC3405
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TYPICAL APPLICATIO S
Single Li-Ion to 1.8V/300mA Regulator
Using All Ceramic Capacitors Optimized for Lowest Profile, ≤ 1.2mm High
VIN
2.7V
TO 4.2V
4
CIN**
1µF
CER
VIN
SW
LTC3405
1
6
4.7µH*
3
22pF
1M
COUT**
1µF
CER
RUN
VFB
MODE
5
GND
200k
2
VOUT
1.8V
COUT**
1µF
CER
332k
*PANASONIC ELT5KT4R7M
**TAIYO YUDEN CERAMIC JMK107BJ105MA
3405 TA04a
100
VIN = 2.7V
90
VOUT
100mV/DIV
AC COUPLED
EFFICIENCY (%)
80
VIN = 3.6V
70
IL
200mA/DIV
VIN = 4.2V
60
50
ILOAD
200mA/DIV
40
30
0.1
VIN = 3.6V
40µs/DIV
ILOAD = 100mA TO 250mA
1000
1
100
10
OUTPUT CURRENT (mA)
3405 TA04c
3405 TA04b
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PACKAGE DESCRIPTIO
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
0.62
MAX
2.90 BSC
(NOTE 4)
0.95
REF
1.22 REF
3.85 MAX 2.62 REF
1.4 MIN
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45
6 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
1.90 BSC
S6 TSOT-23 0302
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
3405fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LTC3405
U
TYPICAL APPLICATIO S
Single Li-Ion to 1.8V/300mA Regulator
All Ceramic Capacitors with Lowest Parts Count
VIN
2.7V
TO 4.2V
4
CIN**
2.2µF
CER
VIN
SW
LTC3405
1
6
4.7µH*
3
887k
22pF
RUN
VFB
MODE
5
GND
2
698k
3405 TA05a
VOUT
1.8V
COUT†
4.7µF
CER
*MURATA LQH3C4R7M34
**TAIYO YUDEN CERAMIC LMK212BJ225MG
†
TAIYO YUDEN CERAMIC JMK212BJ475MG
100
VIN = 2.7V
VOUT
100mV/DIV
AC COUPLED
90
VIN = 4.2V
EFFICIENCY (%)
80
70
VIN = 3.6V
IL
200mA/DIV
60
50
ILOAD
200mA/DIV
40
30
0.1
1000
1
100
10
OUTPUT CURRENT (mA)
VIN = 3.6V
40µs/DIV
ILOAD = 100mA TO 250mA
3405 TA05b
3405 TA05c
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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LTC1174-5
High Efficiency Step-Down and Inverting DC/DC Converters
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Burst Mode Operation
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Monolithic, IOUT to 250mA, IQ = 10µA, 8-Pin MSOP
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LT1616
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Regulation, VIN from 2.65V to 8.5V
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Constant Off-Time, IOUT to 500mA, 1MHz Operation,
VIN from 2.5V to 5.5V
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Monolithic Synchronous Step-Down Switching Regulator
1.19V VREF Pin, Constant Frequency, IOUT to 600mA,
VIN from 2.65V to 8.5V
LTC1767
1.5A, 1.25MHz Step-Down Switching Regulator
3V to 25V Input, 8-Lead MSOP Package
LTC1779
Monolithic Current Mode Step-Down Switching Regulator
550kHz, 6-Lead ThinSOT, V IN from 2.5V to 9.8V
LTC1877
High Efficiency Monolithic Step-Down Regulator
550kHz, MS8, VIN Up to 10V, IQ = 10µA, IOUT to 600mA at VIN = 5V
LTC1878
High Efficiency Monolithic Step-Down Regulator
550kHz, MS8, VIN Up to 6V, IQ = 10µA, IOUT to 600mA at VIN = 3.3V
LTC3404
1.4MHz High Efficiency Monolithic Step-Down Regulator
1.4MHz, MS8, VIN Up to 6V, IQ = 10µA, IOUT to 600mA at VIN = 3.3V
LTC3405A
1.5MHz High Efficiency Monolithic Step-Down Regulator
Stable with Ceramic Output Capacitor
LTC3405A-1.5/
LTC3405A-1.8
1.5MHz High Efficiency Monolithic Step-Down Regulator
Fixed Output Version of LTC3405A
3405fa
16
Linear Technology Corporation
LT/TP 0604 1K REV A • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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