LTC3406B-2ES5#TRMPBF

LTC3406B-2 2.25MHz, 600mA Synchronous Step-Down Regulator in ThinSOTTM U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LTC ®3406B-2 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current with no load is 350µA, dropping to 40%. However, the LTC3406B-2 uses a patent-pending scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles. 1200 VOUT = 1.8V TA = 25°C 1000 80 EFFICIENCY (%) VOUT 1.8V 600mA Low Supply Operation MAXIMUM OUTPUT CURRENT (mA) 100 † An important detail to remember is that at low input supply voltages, the RDS(ON) of the P-channel switch increases (see Typical Performance Characteristics). Therefore, the user should calculate the power dissipation when the LTC3406B-2 is used at 100% duty cycle with low input voltage (See Thermal Considerations in the Applications Information section). 70 60 VIN = 3.6V VIN = 2.7V 50 40 30 20 10 0.1 VIN = 4.2V 800 600 1000 3406B F01b Figure 1b. Efficiency vs Load Current VOUT = 2.5V VOUT = 1.5V 400 200 0 1 100 10 OUTPUT CURRENT (mA) VOUT = 1.8V 2.5 3.0 3.5 4.0 4.5 SUPPLY VOLTAGE (V) 5.0 5.5 3406B F02 Figure 2. Maximum Output Current vs Input Voltage sn3406b2 3406b2fs 7 LTC3406B-2 U W U U APPLICATIO S I FOR ATIO The basic LTC3406B-2 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Table 1. Representative Surface Mount Inductors Part Number Value (µH) DCR (ΩMAX) MAX DC Current (A) Size WxLxH (mm3) Sumida CDRH2D11 1.5 2.2 3.3 0.068 0.098 0.123 0.90 0.78 0.60 3.2 x 3.2 x 1.2 Sumida CDRH2D18/LD 2.2 3.3 4.7 0.041 0.054 0.078 0.85 0.75 0.63 3.2 x 3.2 x 2.0 Sumida CMD4D06 2.2 3.3 4.7 0.116 0.174 0.216 0.95 0.77 0.75 3.5 x 4.1 x 0.8 Murata LQH32C 1.0 2.2 4.7 0.060 0.097 0.150 1.00 0.79 0.65 2.5 x 3.2 x 2.0 Taiyo Yuden LQLBC2518 1.0 1.5 2.2 0.080 0.110 0.130 0.78 0.66 0.60 1.8 x 2.5 x 1.8 Toko D412F 2.2 3.3 4.7 0.14 0.20 0.22 1.14 0.90 0.80 4.6 x 4.6 x 1.2 Inductor Selection For most applications, the value of the inductor will fall in the range of 1µH to 4.7µH. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is ∆IL = 240mA (40% of 600mA). ∆IL = ⎛ V ⎞ VOUT ⎜ 1 − OUT ⎟ VIN ⎠ ⎝ f L 1 ( )( ) (1) The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 720mA rated inductor should be enough for most applications (600mA + 120mA). For better efficiency, choose a low DC-resistance inductor. Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3406B-2 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3406B-2 applications. CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN required IRMS ≅ IOMAX 8 [ ( VOUT VIN − VOUT VIN 1/ 2 )] This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ∆VOUT is determined by: ⎛ 1 ⎞ ∆VOUT ≅ ∆IL ⎜ ESR + ⎟ ⎝ 8 fCOUT ⎠ where f = operating frequency, COUT = output capacitance and ∆IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ∆IL increases with input voltage. sn3406b2 3406b2fs LTC3406B-2 U W U U APPLICATIO S I FOR ATIO Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3406B-2’s control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Output Voltage Programming The output voltage is set by a resistive divider according to the following formula: ⎛ R2 ⎞ VOUT = 0.6⎜ 1 + ⎟ ⎝ R1⎠ (2) The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 3. 0.6V ≤ VOUT ≤ 5.5V R2 VFB LTC3406B-2 R1 GND 3406B F03 Figure 3. Setting the LTC3406B-2 Output Voltage Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3406B-2 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 4. 1 VIN = 3.6V 0.1 POWER LOSS (W) Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. 0.01 VOUT = 2.5V VOUT = 1.8V VOUT = 1.2V 0.001 VOUT = 1.5V 0.0001 0.1 1 10 100 LOAD CURRENT (mA) 1000 3406B F04 Figure 4. Power Lost vs Load Current sn3406b2 3406b2fs 9 LTC3406B-2 U W U U APPLICATIO S I FOR ATIO 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Charateristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. Thermal Considerations In most applications the LTC3406B-2 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3406B-2 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3406B-2 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(θJA) where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TA + TR where TA is the ambient temperature. As an example, consider the LTC3406B-2 in dropout at an input voltage of 2.7V, a load current of 600mA and an ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the P-channel switch at 70°C is approximately 0.52Ω. Therefore, power dissipated by the part is: PD = ILOAD2 • RDS(ON) = 187.2mW For the SOT-23 package, the θJA is 250°C/ W. Thus, the junction temperature of the regulator is: TJ = 70°C + (0.1872)(250) = 116.8°C which is below the maximum junction temperature of 125°C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (∆ILOAD • ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. sn3406b2 3406b2fs 10 LTC3406B-2 U W U U APPLICATIO S I FOR ATIO A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 • CLOAD). Thus, a 10µF capacitor charging to 3.3V would require a 250µs rise time, limiting the charging current to about 130mA. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the 1 RUN LTC3406B-2. These items are also illustrated graphically in Figures 5 and 6. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. 2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground. 3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 4. Keep the switching node, SW, away from the sensitive VFB node. 5. Keep the (–) plates of CIN and COUT as close as possible. VFB 5 LTC3406B-2 2 – R2 R1 GND COUT VOUT 3 + L1 SW VIN CFWD 4 CIN + VIN – 3406B F05 BOLD LINES INDICATE HIGH CURRENT PATHS Figure 5. LTC3406B-2 Layout Diagram VIA TO GND R1 VIN VIA TO VIN CFWD LTC3406B-2 VOUT L1 VIA TO VOUT R2 PIN 1 SW COUT CIN GND 3406B F06 Figure 6. LTC3406B-2 Suggested Layout sn3406b2 3406b2fs 11 LTC3406B-2 U W U U APPLICATIO S I FOR ATIO Design Example A 2.2µH inductor works well for this application. For best efficiency choose a 720mA or greater inductor with less than 0.2Ω series resistance. CIN will require an RMS current rating of at least 0.3A ≅ ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.25Ω. In most cases, a ceramic capacitor will satisfy this requirement. As a design example, assume the LTC3406B-2 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.6A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 2.5V. With this information we can calculate L using equation (1), ⎛ V ⎞ L= VOUT ⎜ 1 − OUT ⎟ VIN ⎠ ⎝ f ∆IL For the feedback resistors, choose R1 = 316k. R2 can then be calculated from equation (2) to be: ⎛V ⎞ R2 = ⎜ OUT − 1⎟ R1 = 1000k ⎝ 0.6 ⎠ 1 ( )( ) (3) Figure 7 shows the complete circuit along with its efficiency curve. Substituting VOUT = 2.5V, VIN = 4.2V, ∆IL = 240mA and f = 2.25MHz in equation (3) gives: L= ⎛ 2.5V ⎞ 2.5V ⎜1 − ⎟ = 1.87µH 1.5MHz(240mA) ⎝ 4.2V ⎠ VIN 2.7V TO 4.2V 4 CIN† 4.7µF CER VIN SW 2.2µH* 3 22pF COUT** 10µF CER LTC3406B-2 1 VFB RUN VOUT 2.5V 5 1M GND 316k 2 3406B F07a *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JHK316BJ106ML † TAIYO YUDEN JMK212BJ475MG Figure 7a 100 90 VOUT = 2.5V TA = 25°C EFFICIENCY (%) 80 VIN = 2.7V VIN = 4.2V 70 60 50 VIN = 3.6V 40 30 20 10 0.1 1 100 10 OUTPUT CURRENT (mA) 1000 3406B G04 Figure 7b sn3406b2 3406b2fs 12 LTC3406B-2 U TYPICAL APPLICATIO S Single Li-Ion 1.2V/600mA Regulator for High Efficiency and Small Footprint VIN 2.7V TO 4.2V 4 CIN† 4.7µF CER VIN SW 3 2.2µH* 22pF VFB RUN GND 2 VOUT 1.2V COUT** 10µF CER LTC3406B-2 1 Load Step 5 301k VOUT 100mV/DIV AC COUPLED IL 500mA/DIV 301k 3406B TA09 *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JHK316BJ106ML † TAIYO YUDEN JMK212BJ475MG ILOAD 500mA/DIV VIN = 3.6V 20µs/DIV VOUT = 1.2V ILOAD = 0mA TO 600mA Efficiency vs Output Current 100 90 Load Step VOUT = 1.2V TA = 25°C VOUT 100mV/DIV AC COUPLED EFFICIENCY (%) 80 70 IL 500mA/DIV VIN = 3.6V 60 50 40 3406B TA11 VIN = 2.7V ILOAD 500mA/DIV VIN = 4.2V 30 VIN = 3.6V 20µs/DIV VOUT = 1.2V ILOAD = 100mA TO 600mA 20 10 0.1 1 100 10 OUTPUT CURRENT (mA) 3406B TA12 1000 3406B G02 sn3406b2 3406b2fs 13 LTC3406B-2 U TYPICAL APPLICATIO S 5V Input to 3.3V/0.6A Regulator 4 VIN 5V † CIN 4.7µF CER VIN SW 3 2.2µH* VOUT 3.3V 22pF COUT** 10µF CER LTC3406B-2 1 RUN VFB GND 2 Load Step 5 VOUT 100mV/DIV AC COUPLED IL 500mA/DIV 1M 221k ILOAD 500mA/DIV 3406B TA13 *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JHK316BJ106ML † TAIYO YUDEN JMK212BJ475MG VIN = 3.6V 20µs/DIV VOUT = 3.3V ILOAD = 0mA TO 600mA Efficiency vs Output Current 100 90 Load Step VOUT = 3.3V VIN = 5V VOUT 100mV/DIV AC COUPLED 80 EFFICIENCY (%) 3406B TA15 70 IL 500mA/DIV 60 50 ILOAD 500mA/DIV 40 30 VIN = 3.6V 20µs/DIV VOUT = 3.3V ILOAD = 100mA TO 600mA 20 10 0.1 1 100 10 OUTPUT CURRENT (mA) 3406B TA16 1000 3406B TA14 sn3406b2 3406b2fs 14 LTC3406B-2 U PACKAGE DESCRIPTIO S5 Package 5-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1635) 0.62 MAX 0.95 REF 2.90 BSC (NOTE 4) 1.22 REF 1.4 MIN 3.85 MAX 2.62 REF 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.30 – 0.45 TYP 5 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.20 BSC 0.01 – 0.10 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF 0.09 – 0.20 (NOTE 3) NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 1.90 BSC S5 TSOT-23 0302 sn3406b2 3406b2fs Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LTC3406B-2 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1616 500mA (IOUT), 1.4MHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN: 3.6V to 25V, VOUT(MIN) = 1.25V, IQ = 1.9mA, ISD < 1µA, ThinSOT Package LT1676 450mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN: 7.4V to 60V, VOUT(MIN) = 1.24V, IQ = 3.2mA, ISD = 2.5µA, S8 Package LTC1877 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.7V to 10V, VOUT(MIN) = 0.8V, IQ = 10µA, ISD < 1µA, MS8 Package LTC1878 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.7V to 6V, VOUT(MIN) = 0.8V, IQ = 10µA, ISD < 1µA, MS8 Package LTC1879 1.2A (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.7V to 10V, VOUT(MIN) = 0.8V, IQ = 15µA, ISD < 1µA, TSSOP-16 Package LTC3403 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter with Bypass Transistor 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = Dynamically Adjustable, IQ = 20µA, ISD < 1µA, DFN Package LTC3404 600mA (IOUT), 1.4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.7V to 6V, VOUT(MIN) = 0.8V, IQ = 10µA, ISD < 1µA, MS8 Package LTC3405/LTC3405A 300mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 20µA, ISD < 1µA, ThinSOT Package LTC3406/LTC3406B 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20µA, ISD < 1µA, ThinSOT Package LTC3407 Dual Output (600mA × 2) 1.5MHz Synchronous Step-Down DC/DC Converter 95% Efficiency VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA, MS10E Package LTC3408 600mA (IOUT), 1.5MHz Synchronous Step-Down DC/DC Converter with 0.08Ω Bypass Transistor 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = Dynamically Adjustable, IQ = 1.5mA, ISD < 1µA, DFN Package LTC3411 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA, ISD < 1µA, MS Package LTC3412 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA, ISD < 1µA, TSSOP-16E Package LTC3414 4A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.25V to 5.5V, VOUT(MIN) = 0.8V, IQ = 64µA, TSSOP-20E Package LTC3440 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT: 2.5V to 5.5V, IQ = 25µA, ISD < 1µA, MS Package LTC3441 1A (IOUT), 1MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT: 2.4V to 5.25V, IQ = 25µA, DFN Package sn3406b2 3406b2fs 16 Linear Technology Corporation LT/TP 0204 1K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2003