LTC3609
32V, 6A Monolithic
Synchronous Step-Down
DC/DC Converter
Description
Features
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6A Output Current
Wide VIN Range = 4V to 32V (36V Maximum)
Internal N-Channel MOSFETs
True Current Mode Control
Optimized for High Step-Down Ratios
tON(MIN) ≤ 100ns
Extremely Fast Transient Response
Stable with Ceramic COUT
±1% 0.6V Voltage Reference
Power Good Output Voltage Monitor
Adjustable On-Time/Switching Frequency
Adjustable Current Limit
Programmable Soft-Start
Output Overvoltage Protection
Optional Short-Circuit Shutdown Timer
Low Shutdown IQ: 15µA
Available in a 7mm × 8mm 52-Pin QFN Package
Applications
Point of Load Regulation
Distributed Power Systems
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L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.
Protected by U.S. Patents, including 5481178, 6100678, 6580258, 5847554, 6304066.
The LTC®3609 is a high efficiency, monolithic synchronous
step-down DC/DC converter that can deliver up to 6A output
current from a 4V to 32V (36V maximum) input supply. It
uses a valley current control architecture to deliver very
low duty cycle operation at high frequency with excellent
transient response. The operating frequency is selected
by an external resistor and is compensated for variations
in VIN and VOUT.
The LTC3609 can be configured for discontinuous or
forced continuous operation at light load. Forced continuous operation reduces noise and RF interference while
discontinuous mode provides high efficiency by reducing
switching losses at light loads.
Fault protection is provided by internal foldback current
limiting, an output overvoltage comparator and an optional
short-circuit shutdown timer. Soft-start capability for supply sequencing is accomplished using an external timing
capacitor. The regulator current limit is user programmable.
A power good output voltage monitor indicates when
the output is in regulation. The LTC3609 is available in a
compact 7mm × 8mm QFN package.
Typical Application
Efficiency and Power Loss
vs Load Current
High Efficiency Step-Down Converter
VON
ION
RUN/SS
VIN
100pF
187k
100
10µF
x3
1.2µH
SW
1000pF
0.22µF
15.8k
ITH
BOOST
SGND
INTVCC
30.1k
FCB
VRNG
PGOOD
EXTVCC
100µF
x2
10000
EFFICIENCY
80
1000
70
60
50
40
VFB
9.53k
100
POWER LOSS
30
10
20
10 VOUT = 2.5V
EXTVCC = 5V
0
0.01
0.1
1
LOAD CURRENT (A)
4.7µF
PGND
3609 TA01a
VOUT
2.5V
6A
90
POWER LOSS (mW)
LTC3609
VIN
4V TO 32V
EFFICIENCY (%)
0.1µF VOUT
VIN = 12V
VIN = 25V
1
10
3609 TA01b
3609fb
LTC3609
Absolute Maximum Ratings
Pin Configuration
(Note 1)
41 SW
42 SW
43 SW
44 SW
45 SW
46 SW
47 SW
48 PVIN
49 PVIN
50 PVIN
51 PVIN
TOP VIEW
52 PVIN
Input Supply Voltage (SVIN, PVIN, ION)........ 36V to –0.3V
Boosted Topside Driver Supply Voltage
(BOOST)................................................. 42V to –0.3V
SW Voltage............................................. 36V to –0.3V
INTVCC, EXTVCC, (BOOST – SW), RUN/SS,
PGOOD Voltages....................................... 7V to –0.3V
FCB, VON, VRNG Voltages............. INTVCC + 0.3V to –0.3V
ITH, VFB Voltages........................................ 2.7V to –0.3V
Operating Junction Temperature Range
(Notes 2, 4)......................................... –40°C to 125°C
Storage Temperature Range.................... –55°C to 125°C
PVIN 1
40 PGND
PVIN 2
39 PGND
PVIN 3
38 PGND
53
PVIN
PVIN 4
PVIN 5
37 PGND
55
SW
36 PGND
PVIN 6
35 PGND
PVIN 7
34 PGND
SW 8
33 SW
NC 9
32 INTVCC
SGND 10
31 INTVCC
BOOST 11
30 SVIN
54
SGND
RUN/SS 12
29 EXTVCC
VON 13
28 NC
SGND 26
NC 25
NC 24
VFB 23
ION 22
NC 21
SGND 20
FCB 19
ITH 18
VRNG 17
SGND 15
27 SGND
PGOOD 16
SGND 14
WKG PACKAGE
52-LEAD (7mm s 8mm) QFN MULTIPAD
TJMAX = 125°C, θJA = 29°C/W
order information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3609EWKG#PBF
LTC3609EWKG#TRPBF
LTC3609WKG
52-Lead (7mm × 8mm) Plastic QFN
–40°C to 125°C
LTC3609IWKG#PBF
LTC3609IWKG#TRPBF
LTC3609WKG
52-Lead (7mm × 8mm) Plastic QFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3609fb
LTC3609
Electrical
Characteristics
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loop
SVIN
Operating Input Voltage Range
IQ
Input DC Supply Current
Normal
Shutdown Supply Current
VFB
Feedback Reference Voltage
4
ITH = 1.2V (Note 3)
–40°C to 85°C
–40°C to 125°C
l
0.594
0.590
32
V
900
15
2000
30
µA
µA
0.600
0.600
0.606
0.610
V
V
∆VFB(LINEREG)
Feedback Voltage Line Regulation
VIN = 4V to 30V, ITH = 1.2V (Note 3)
0.002
∆VFB(LOADREG)
Feedback Voltage Load Regulation
ITH = 0.5V to 1.9V (Note 3)
–0.05
–0.3
%
IFB
Feedback Input Current
VFB = 0.6V
–5
±50
nA
gm(EA)
Error Amplifier Transconductance
ITH = 1.2V (Note 3)
mS
VFCB
Forced Continuous Threshold
IFCB
Forced Continuous Pin Current
VFCB = 0.6V
tON
On-Time
ION = 60µA, VON = 1.5V
ION = 60µA, VON = 0V
tON(MIN)
Minimum On-Time
tOFF(MIN)
%/V
l
1.4
1.7
2
l
0.54
0.6
0.66
V
–1
–2
µA
220
280
110
340
ns
ns
ION = 180µA, VON = 0V
60
100
ns
Minimum Off-Time
ION = 30µA, VON = 1.5V
320
500
ns
IVALLEY(MAX)
Maximum Valley Current
VRNG = 0V, VFB = 0.56V, FCB = 0V
VRNG = 1.2V, VFB = 0.56V, FCB = 0V
IVALLEY(MIN)
Maximum Reverse Valley Current
VRNG = 0V, VFB = 0.64V, FCB = 0V
VRNG = 1.2V, VFB = 0.64V, FCB = 0V
∆VFB(OV)
Output Overvoltage Fault Threshold
VRUN/SS(ON)
RUN Pin Start Threshold
VRUN/SS(LE)
RUN Pin Latchoff Enable Threshold
RUN/SS Pin Rising
4
4.5
V
VRUN/SS(LT)
RUN Pin Latchoff Threshold
RUN/SS Pin Falling
3.5
4.2
V
IRUN/SS(C)
Soft-Start Charge Current
VRUN/SS = 0V
–0.5
–1.2
–3
µA
IRUN/SS(D)
Soft-Start Discharge Current
VRUN/SS = 4.5V, VFB = 0V
0.8
1.8
3
µA
VIN(UVLO)
Undervoltage Lockout
VIN Falling
l
3.4
3.9
V
VIN(UVLOR)
Undervoltage Lockout Release
VIN Rising
l
3.5
4
V
RDS(ON)
Top Switch On-Resistance
Bottom Switch On-Resistance
18
13
27
22
mΩ
mΩ
l
l
l
4
6
9
14
A
A
4
7
A
A
7
10
13
%
0.8
1.5
2
V
3609fb
LTC3609
Electrical
Characteristics
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
4.7
5
5.5
V
–0.1
±2
%
300
mV
Internal VCC Regulator
VINTVCC
Internal VCC Voltage
6V < VIN < 30V, VEXTVCC = 4V
∆VLDO(LOADREG)
Internal VCC Load Regulation
ICC = 0mA to 20mA, VEXTVCC = 4V
VEXTVCC
EXTVCC Switchover Voltage
ICC = 20mA, VEXTVCC Rising
∆VEXTVCC
EXTVCC Switch Drop Voltage
ICC = 20mA, VEXTVCC = 5V
∆VEXTVCC(HYS)
EXTVCC Switchover Hysteresis
l
l
4.5
4.7
150
V
500
mV
PGOOD Output
∆VFBH
PGOOD Upper Threshold
VFB Rising
7
10
13
%
∆VFBL
PGOOD Lower Threshold
VFB Falling
–7
–10
–13
%
∆VFB(HYS)
PGOOD Hysteresis
VFB Returning
1
2.5
%
VPGL
PGOOD Low Voltage
IPGOOD = 5mA
0.15
0.4
V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD as follows:
TJ = TA + (PD • 29°C/W) (θJA is simulated per JESD51-7 high effective
thermal conductivity test board).
θJC = 1°C/W (θJC is simulated when heat sink is applied at the bottom
of the package).
Note 3: The LTC3609 is tested in a feedback loop that adjusts VFB to
achieve a specified error amplifier output voltage (ITH). The specification at
85°C is not tested in production. This specification is assured by design,
characterization, and correlation to testing at 125°C.
Note 4: The LTC3609 is tested under pulsed load conditions such that
TJ ≈ TA. The LTC3609E is guaranteed to meet specifications from
0°C to 125°C junction temperature. Specifications over the –40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3609I is guaranteed over the full –40°C to 125°C operating junction
temperature range. Note that the maximum ambient temperature
consistent with these specifications is determined by specific operating
conditions in conjunction with board layout, the rated package thermal
impedance and other environmental factors.
Typical Performance Characteristics
Transient Response
Transient Response
(Discontinuous Mode)
VOUT
200mV/DIV
Start-Up
VOUT
200mV/DIV
RUN/SS
2V/DIV
IL
5A/DIV
IL
5A/DIV
VOUT
1V/DIV
ILOAD
5A/DIV
ILOAD
5A/DIV
20µs/DIV
LOAD STEP 0A TO 5A
VIN = 25V
VOUT = 2.5V
FCB = 0V
FIGURE 6 CIRCUIT
3609 G01
IL
5A/DIV
20µs/DIV
LOAD STEP 1A TO 6A
VIN = 25V
VOUT = 2.5V
FCB = INTVCC
FIGURE 6 CIRCUIT
3609 G02
40ms/DIV
VIN = 12V
VOUT = 2.5V
RLOAD = 0.5Ω
FIGURE 6 CIRCUIT
3609 G03
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LTC3609
Typical Performance Characteristics
Efficiency vs Load Current
Efficiency vs Input Voltage
100
100
40
30
ILOAD = 6A
90
ILOAD = 1A
85
20
10 VIN = 12V
FREQUENCY = 550kHz
0
0.01
0.1
1
LOAD CURRENT (A)
80
10
FREQUENCY (kHz)
50
EFFICIENCY (%)
EFFICIENCY (%)
60
600
95
VOUT = 5V
VOUT = 3.3V
VOUT = 2.5V
VOUT = 1.8V
VOUT = 1.5V
VOUT = 1.2V
VOUT = 1V
70
650
FCB = 5V
FIGURE 6 CIRCUIT
90
80
Frequency vs Load Current
0.80
5
8
11
14 17 20 23 26
INPUT VOLTAGE (V)
32
FCB = 0V
FIGURE 6 CIRCUIT
5
300
200
100
2.5
FIGURE 6 CIRCUIT
17
20
23
26
29
32
3609 G06
FIGURE 6 CIRCUIT
2.0
0.20
0
1.5
CONTINUOUS
MODE
1.0
0.5
DISCONTINUOUS
MODE
–0.60
4
6
LOAD CURRENT (A)
14
ITH Voltage vs Load Current
–0.40
2
11
3609 G05
–0.20
DISCONTINUOUS MODE
0
8
INPUT VOLTAGE (V)
ITH VOLTAGE (V)
∆VOUT (%)
FREQUENCY (kHz)
29
400
0.40
400
–0.80
8
0
3609 G07
Load Current vs ITH Voltage at
Different VRNG
10000
15
2
4
6
LOAD CURRENT (A)
0
8
0
2
3609 G08
On-Time vs ION Current
4
6
LOAD CURRENT (A)
8
3609 G09
On-Time vs VON Voltage
1000
VVON = 0V
VRNG = 1.2V
ION = 30µA
800
10
VRNG = 1V
0
1000
ON-TIME (ns)
VRNG = 0.7V
5
ON-TIME (ns)
LOAD CURRENT (A)
ILOAD = 1A
0.60
CONTINUOUS MODE
500
100
600
400
200
–5
–10
500
Load Regulation
700
0
ILOAD = 6A
550
450
3609 G04
600
Frequency vs Input Voltage
0
0.5
1.0
1.5
ITH VOLTAGE (V)
2.0
2.5
3609 G10
10
1
10
ION CURRENT (µA)
100
3609 G11
0
0
1
2
VON VOLTAGE (V)
3
3609 G12
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LTC3609
Typical Performance Characteristics
On-Time vs Temperature
MAXIMUM VALLEY CURRENT LIMIT (A)
250
ON-TIME (ns)
15
IION = 30µA
VVON = 0V
200
150
100
50
0
–50 –25
25
75
0
50
TEMPERATURE (°C)
100
Maximum Valley Current Limit
vs RUN/SS Voltage
15
FIGURE 6 CIRCUIT
MAXIMUM VALLEY CURRENT LIMIT (A)
300
Maximum Valley Current Limit
vs VRNG Voltage
12
9
6
3
0
125
0.5
0.6
0.7 0.8 0.9 1.0
VRNG VOLTAGE (V)
1.1
FIGURE 6 CIRCUIT
12
9
6
3
0
1.65 1.90 2.15 2.40 2.65 2.90 3.15 3.40
RUN/SS VOLTAGE (V)
1.2
3609 G14
3609 G15
3609 G13
Maximum Valley Current Limit
vs Temperature
9
6
3
–25
0
25
50
75
TEMPERATURE (°C)
100
6
4
2
4
3609 G16
12
20
28
INPUT VOLTAGE (V)
75
0
25
50
TEMPERATURE (°C)
100
125
1.6
1.4
3609 G19
1.0
–50
0.2
0.3
VFB (V)
0.4
0.5
0.6
3609 G18
40
EXTVCC OPEN
1200
1.2
–25
0.1
0
1400
INPUT CURRENT (µA)
gm (mS)
0.59
2
30
1000
25
800
50
0
75
25
TEMPERATURE (°C)
100
125
3609 G20
20
SHUTDOWN
600
15
400
0
10
EXTVCC = 5V
200
–25
35
SHUTDOWN CURRENT (µA)
0.60
4
Input and Shutdown Currents
vs Input Voltage
1.8
0.61
6
3609 G17
2.0
0.62
8
0
36
Error Amplifier gm vs Temperature
Feedback Reference Voltage
vs Temperature
FEEDBACK REFERENCE VOLTAGE (V)
8
0
125
MAXIMUM VALLEY CURRENT LIMIT (A)
12
0.58
–50
10
10
MAXIMUM VALLEY CURRENT (A)
MAXIMUM VALLEY CURRENT LIMIT (A)
15
0
–50
Maximum Valley Current Limit
in Foldback
Maximum Valley Current vs Input
Voltage
5
0
5
10
20
15
INPUT VOLTAGE (V)
25
30
0
3609 G21
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LTC3609
Typical Performance Characteristics
INTVCC Load Regulation
EXTVCC Switch Resistance
vs Temperature
EXTVCC SWITCH RESISTANCE (Ω)
0.20
∆INTVCC (%)
0.10
0
–0.10
–0.20
–0.30
–0.40
0
40
10
20
30
INTVCC LOAD CURRENT (mA)
10
25
8
20
IEXTVCC (mA)
0.30
6
4
2
VIN = 24V
15
10
5
0
–50
50
IEXTVCC vs Frequency
–25
0
50
75
25
TEMPERATURE (°C)
100
0
400
125
500
600
700
800
FREQUENCY (kHz)
3609 G22
900
1000
3609 G28
3609 G23
RUN/SS Pin Current
vs Temperature
FCB Pin Current vs Temperature
3
RUN/SS PIN CURRENT (µA)
–0.25
–0.50
–0.75
–1.00
–1.25
–1.50
–50 –25
25
75
0
50
TEMPERATURE (°C)
100
125
5.0
2
RUN/SS PIN CURRENT (µA)
0
FCB PIN CURRENT (µA)
RUN/SS Pin Current
vs Temperature
PULL-DOWN CURRENT
1
0
–1
–2
–50
PULL-UP CURRENT
–25
0
50
75
25
TEMPERATURE (°C)
100
4.5
LATCHOFF ENABLE
4.0
LATCHOFF THRESHOLD
3.5
3.0
–50
125
–25
3609 G25
75
0
25
50
TEMPERATURE (°C)
100
125
3609 G26
3609 G24
Load Step
4.0
Efficiency vs Load Current
100
f = 500kHz
95
IL
5A/DIV
3.5
3.0
90
EFFICIENCY (%)
UNDERVOLTAGE LOCKOUT THRESHOLD (V)
Undervoltage Lockout Threshold
vs Temperature
VOUT
200mV/DIV
2.5
2.0
–50
–25
75
0
25
50
TEMPERATURE (°C)
100
125
3609 G27
40µs/DIV
LOAD STEP 1A TO 4A
VIN = 24V
VOUT = 12V
FCB = 0V
FIGURE 8 CIRCUIT
3609 G29
85
DCM
80
75
70
CCM
65
60
55
50
0.01
VIN = 24V
FREQUENCY = 500kHz
0.1
1
LOAD CURRENT (A)
FIGURE 8 CIRCUIT
10
3609 G30
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LTC3609
Pin Functions
PVIN (Pins 1, 2, 3, 4, 5, 6, 7, 48, 49, 50, 51, 52, 53):
Main Input Supply. Decouple this pin to power PGND with
the input capacitance, CIN.
SW (Pins 8, 33, 41, 42, 43, 44, 45, 46, 47, 55): Switch
Node Connection to the Inductor. The (–) terminal of the
bootstrap capacitor, CB, also connects here. This pin swings
from a diode voltage drop below ground up to VIN.
NC (Pins 9, 21, 24, 25, 28): No Connection.
SGND (Pins 10, 14, 15, 20, 26, 27, 54): Signal Ground. All
small-signal components and compensation components
should connect to this ground, which in turn connects to
PGND at one point.
BOOST (Pin 11): Boosted Floating Driver Supply. The (+)
terminal of the bootstrap capacitor, CB, connects here.
This pin swings from a diode voltage drop below INTVCC
up to VIN + INTVCC.
RUN/SS (Pin 12): Run Control and Soft-Start Input. A
capacitor to ground at this pin sets the ramp time to full
output current (approximately 3s/µF) and the time delay
for overcurrent latchoff (see Applications Information).
Forcing this pin below 0.8V shuts down the device.
VON (Pin 13): On-Time Voltage Input. Voltage trip point for
the on-time comparator. Tying this pin to the output voltage or an external resistive divider from the output makes
the on-time proportional to VOUT. The comparator input
defaults to 0.7V when the pin is grounded and defaults to
2.4V when the pin is tied to INTVCC. Tie this pin to INTVCC
in high VOUT applications to use a lower RON value.
PGOOD (Pin 16): Power Good Output. Open-drain logic
output that is pulled to ground when the output voltage
is not within ± 10% of the regulation point.
ITH (Pin 18): Current Control Threshold and Error Amplifier
Compensation Point. The current comparator threshold
increases with this control voltage. The voltage ranges
from 0V to 2.4V with 0.8V corresponding to zero sense
voltage (zero current).
FCB (Pin 19): Forced Continuous Input. Tie this pin to
ground to force continuous synchronous operation at low
load, to INTVCC to enable discontinuous mode operation at
low load or to a resistive divider from a secondary output
when using a secondary winding.
ION (Pin 22): On-Time Current Input. Tie a resistor from VIN
to this pin to set the one-shot timer current and thereby
set the switching frequency.
VFB (Pin 23): Error Amplifier Feedback Input. This pin
connects the error amplifier input to an external resistive
divider from VOUT.
EXTVCC (Pin 29): External VCC Input. When EXTVCC exceeds
4.7V, an internal switch connects this pin to INTVCC and
shuts down the internal regulator so that controller and
gate drive power is drawn from EXTVCC. Do not exceed
7V at this pin and ensure that EXTVCC < VIN.
SVIN (Pin 30): Supply Pin for Internal PWM Controller.
INTVCC (Pins 31, 32): Internal 5V Regulator Output. The
driver and control circuits are powered from this voltage.
Decouple this pin to power ground with a minimum of
4.7µF low ESR tantalum or ceramic capacitor.
PGND (Pins 34, 35, 36, 37, 38, 39, 40): Power Ground.
Connect this pin closely to the (–) terminal of CVCC and
the (–) terminal of CIN.
VRNG (Pin 17): Current Limit Range Input. The voltage at
this pin adjusts maximum valley current and can be set
from 0.7V to 1.2V by a resistive divider from INTVCC. It
defaults to 0.7V if the VRNG pin is tied to ground which
results in a typical 9A current limit.
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LTC3609
Functional Diagram
RON
VON
ION
13
22
FCB
EXTVCC
19
29
SVIN
30
4.7V
0.7V
2.4V
+
1µA
PVIN
–
1, 2, 3, 4, 5,
6, 7, 48, 49,
50, 51, 52, 53
0.6V
REF
0.6V
CIN
5V
REG
INTVCC
+
–
31, 32
F
11
VVON
tON =
(10pF)
IION
R
S
Q
FCNT
SW
+
ICMP
–
L1
DB
VOUT
8, 33, 41, 42,
43, 44, 45,
46, 47, 55
SWITCH
LOGIC
IREV
–
SHDN
1.4V
COUT
OV
M2
+
CVCC
17
PGND
s
34, 35, 36, 37,
38, 39, 40
(0.5 TO 2)
0.7V
16 PGOOD
1
240k
+
1V
Q2 Q4
–
Q6
ITHB
R2
0.54V
UV
23
Q3 Q1
VFB
+
R1
OV
+
–
–
0.8V
–
SS
+
SGND
10, 14, 15, 20,
26, 27, 54
0.66V
RUN
SHDN
1.2µA
EA
s3.3
NC
9, 21, 24, 25, 28
+
–
–
+
VRNG
CB
M1
ON
20k
+
BOOST
6V
0.6V
18
ITH
0.4V
12
3609 FD
RUN/SS
CSS
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LTC3609
Operation
Main Control Loop
The LTC3609 is a high efficiency monolithic synchronous,
step-down DC/DC converter utilizing a constant on-time,
current mode architecture. It operates from an input
voltage range of 4V to 32V/36V maximum and provides
a regulated output voltage at up to 6A of output current.
The internal synchronous power switch increases efficiency
and eliminates the need for an external Schottky diode. In
normal operation, the top MOSFET is turned on for a fixed
interval determined by a one-shot timer OST. When the
top MOSFET is turned off, the bottom MOSFET is turned
on until the current comparator ICMP trips, restarting the
one-shot timer and initiating the next cycle. Inductor current
is determined by sensing the voltage between the PGND
and SW pins using the bottom MOSFET on-resistance.
The voltage on the ITH pin sets the comparator threshold
corresponding to inductor valley current. The error amplifier, EA, adjusts this voltage by comparing the feedback
signal VFB from the output voltage with an internal 0.6V
reference. If the load current increases, it causes a drop
in the feedback voltage relative to the reference. The ITH
voltage then rises until the average inductor current again
matches the load current.
At light load, the inductor current can drop to zero and
become negative. This is detected by current reversal
comparator IREV which then shuts off M2 (see Functional Diagram), resulting in discontinuous operation. Both
switches will remain off with the output capacitor supplying
the load current until the ITH voltage rises above the zero
current level (0.8V) to initiate another cycle. Discontinuous mode operation is disabled by comparator F when
the FCB pin is brought below 0.6V, forcing continuous
synchronous operation.
The operating frequency is determined implicitly by the
top MOSFET on-time and the duty cycle required to maintain regulation. The one-shot timer generates an on-time
that is proportional to the ideal duty cycle, thus holding
frequency approximately constant with changes in VIN.
The nominal frequency can be adjusted with an external
resistor, RON.
Overvoltage and undervoltage comparators OV and UV
pull the PGOOD output low if the output feedback voltage exits a ±10% window around the regulation point.
Furthermore, in an overvoltage condition, M1 is turned
off and M2 is turned on and held on until the overvoltage
condition clears.
Foldback current limiting is provided if the output is
shorted to ground. As VFB drops, the buffered current
threshold voltage ITHB is pulled down by clamp Q3 to
a 1V level set by Q4 and Q6. This reduces the inductor
valley current level to one sixth of its maximum value as
VFB approaches 0V.
Pulling the RUN/SS pin low forces the controller into its
shutdown state, turning off both M1 and M2. Releasing
the pin allows an internal 1.2µA current source to charge
up an external soft-start capacitor, CSS. When this voltage
reaches 1.5V, the controller turns on and begins switching,
but with the ITH voltage clamped at approximately 0.6V
below the RUN/SS voltage. As CSS continues to charge,
the soft-start current limit is removed.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most of
the internal controller circuitry is derived from the INTVCC
pin. The top MOSFET driver is powered from a floating
bootstrap capacitor, CB. This capacitor is recharged from
INTVCC through an external Schottky diode, DB, when
the top MOSFET is turned off. When the EXTVCC pin is
grounded, an internal 5V low dropout regulator supplies
the INTVCC power from VIN. If EXTVCC rises above 4.7V,
the internal regulator is turned off, and an internal switch
connects EXTVCC to INTVCC. This allows a high efficiency
source connected to EXTVCC, such as an external 5V supply or a secondary output from the converter, to provide
the INTVCC power. Voltages up to 7V can be applied to
EXTVCC for additional gate drive. If the input voltage is
low and INTVCC drops below 3.5V, undervoltage lockout
circuitry prevents the power switches from turning on.
3609fb
10
LTC3609
Applications Information
The basic LTC3609 application circuit is shown on the
front page of this data sheet. External component selection
is primarily determined by the maximum load current.
The LTC3609 uses the on-resistance of the synchronous
power MOSFET for determining the inductor current. The
desired amount of ripple current and operating frequency
also determines the inductor value. Finally, CIN is selected
for its ability to handle the large RMS current into the
converter and COUT is chosen with low enough ESR to meet
the output voltage ripple and transient specification.
VON and PGOOD
The LTC3609 has an open-drain PGOOD output that
indicates when the output voltage is within ±10% of the
regulation point. The LTC3609 also has a VON pin that
allows the on-time to be adjusted. Tying the VON pin high
results in lower values for RON which is useful in high VOUT
applications. The VON pin also provides a means to adjust
the on-time to maintain constant frequency operation in
applications where VOUT changes and to correct minor
frequency shifts with changes in load current.
VRNG Pin and ILIMIT Adjust
The VRNG pin is used to adjust the maximum inductor
valley current, which in turn determines the maximum
average output current that the LTC3609 can deliver. The
maximum output current is given by:
IOUT(MAX) = IVALLEY(MAX) + 1/2 ∆IL
The IVALLEY(MAX) is shown in the figure “Maximum Valley
Current Limit vs VRNG Voltage” in the Typical Performance
Characteristics.
An external resistor divider from INTVCC can be used to
set the voltage on the VRNG pin from 0.7V to 1.2V, or it can
be simply tied to ground force a default value equivalent
to 0.7V. When setting current limit, ensure that the junction temperature does not exceed the maximum rating of
125°C. Do not float the VRNG pin.
Operating Frequency
The choice of operating frequency is a tradeoff between
efficiency and component size. Low frequency operation
improves efficiency by reducing MOSFET switching losses
but requires larger inductance and/or capacitance in order
to maintain low output ripple voltage.
The operating frequency of LTC3609 applications is determined implicitly by the one-shot timer that controls the
on-time, tON, of the top MOSFET switch. The on-time is
set by the current into the ION pin and the voltage at the
VON pin according to:
tON =
VVON
(10pF )
IION
Tying a resistor RON from VIN to the ION pin yields an
on-time inversely proportional to VIN. The current out of
the ION pin is:
IION =
VIN
RON
For a step-down converter, this results in approximately
constant frequency operation as the input supply varies:
f=
VOUT
[ HZ ]
VVON RON(10pF )
To hold frequency constant during output voltage changes,
tie the VON pin to VOUT or to a resistive divider from VOUT
when VOUT > 2.4V. The VON pin has internal clamps that
limit its input to the one-shot timer. If the pin is tied below
0.7V, the input to the one-shot is clamped at 0.7V. Similarly,
if the pin is tied above 2.4V, the input is clamped at 2.4V.
In high VOUT applications, tying VON to INTVCC so that the
comparator input is 2.4V results in a lower value for RON.
Figures 1a and 1b show how RON relates to switching
frequency for several common output voltages.
3609fb
11
LTC3609
Applications Information
Because the voltage at the ION pin is about 0.7V, the current into this pin is not exactly inversely proportional to
VIN, especially in applications with lower input voltages.
To correct for this error, an additional resistor RON2 connected from the ION pin to the 5V INTVCC supply will further
stabilize the frequency.
RON2 =
5V
RON
0.7 V
Changes in the load current magnitude will also cause
frequency shift. Parasitic resistance in the MOSFET
switches and inductor reduce the effective voltage across
the inductance, resulting in increased duty cycle as the
load current increases. By lengthening the on-time slightly
as current increases, constant frequency operation can be
maintained. This is accomplished with a resistive divider
from the ITH pin to the VON pin and VOUT. The values
required will depend on the parasitic resistances in the
specific application. A good starting point is to feed about
25% of the voltage change at the ITH pin to the VON pin
as shown in Figure 2a. Place capacitance on the VON pin
to filter out the ITH variations at the switching frequency.
The resistor load on ITH reduces the DC gain of the error
amp and degrades load regulation, which can be avoided
by using the PNP emitter follower of Figure 2b.
SWITCHING FREQUENCY (kHz)
1000
VOUT = 3.3V
VOUT = 1.5V
100
100
VOUT = 2.5V
1000
RON (kΩ)
10000
3609 F01a
Figure 1a. Switching Frequency vs RON (VON = 0V)
SWITCHING FREQUENCY (kHz)
1000
VOUT = 12V
VOUT = 5V
VOUT = 3.3V
100
100
1000
RON (kΩ)
10000
3609 F01b
Figure 1b. Switching Frequency vs RON (VON = INTVCC)
3609fb
12
LTC3609
Applications Information
2.0
CVON
0.01µF
RVON2
100k
RC
VON
LTC3609
ITH
CC
(2a)
VOUT
INTVCC
RVON1
3k
10k
CVON
0.01µF
RVON2
10k
RC
Q1
2N5087
CC
VON
LTC3609
ITH
1.0
0.5
0
0.25
0.50
0.75
DUTY CYCLE (VOUT/VIN)
1.0
3609 F03
Figure 3. Maximum Switching Frequency vs Duty Cycle
Minimum Off-time and Dropout Operation
The minimum off-time, tOFF(MIN), is the smallest amount
of time that the LTC3609 is capable of turning on the bottom MOSFET, tripping the current comparator and turning
the MOSFET back off. This time is generally about 250ns.
The minimum off-time limit imposes a maximum duty
cycle of tON/(tON + tOFF(MIN)). If the maximum duty cycle
is reached, due to a dropping input voltage for example,
then the output will drop out of regulation. The minimum
input voltage to avoid dropout is:
tON + tOFF(MIN)
tON
A plot of maximum duty cycle vs frequency is shown in
Figure 3.
Setting the Output Voltage
The LTC3609 develops a 0.6V reference voltage between
the feedback pin, VFB, and the signal ground as shown in
Figure 6. The output voltage is set by a resistive divider
according to the following formula:
R2
VOUT = 0.6V 1+
R1
DROPOUT
REGION
3609 F02
Figure 2. Correcting Frequency Shift with Load Current Changes
1.5
0
(2b)
VIN(MIN) = VOUT
SWITCHING FREQUENCY (MHz)
VOUT
RVON1
30k
To improve the frequency response, a feed-forward capacitor, C1, may also be used. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
Inductor Selection
Given the desired input and output voltages, the inductor value and operating frequency determine the ripple
current:
V V
ΔIL = OUT 1− OUT
f L
VIN
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors and output voltage
ripple. Highest efficiency operation is obtained at low
frequency with small ripple current. However, achieving
this requires a large inductor. There is a tradeoff between
component size, efficiency and operating frequency.
A reasonable starting point is to choose a ripple current
that is about 40% of IOUT(MAX). The largest ripple current
occurs at the highest VIN. To guarantee that ripple current
does not exceed a specified maximum, the inductance
should be chosen according to:
V
VOUT
OUT
L =
1− V
f
ΔI
IN(MAX)
L(MAX)
3609fb
13
LTC3609
Applications Information
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores.
A variety of inductors designed for high current, low voltage applications are available from manufacturers such as
Sumida, Panasonic, Coiltronics, Coilcraft and Toko.
CIN and COUT Selection
The input capacitance, CIN, is required to filter the square
wave current at the drain of the top MOSFET. Use a low ESR
capacitor sized to handle the maximum RMS current.
IRMS ≅ IOUT(MAX)
VOUT
VIN
VIN
–1
VOUT
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT(MAX)/2. This simple worst-case condition is
commonly used for design because even significant deviations do not offer much relief. Note that ripple current
ratings from capacitor manufacturers are often based on
only 2000 hours of life which makes it advisable to derate
the capacitor.
The selection of COUT is primarily determined by the
ESR required to minimize voltage ripple and load step
transients. The output ripple ∆VOUT is approximately
bounded by:
1
ΔVOUT ≤ ΔIL ESR+
8fCOUT
Since ∆IL increases with input voltage, the output ripple
is highest at maximum input voltage. Typically, once the
ESR requirement is satisfied, the capacitance is adequate
for filtering and has the necessary RMS current rating.
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, special polymer, aluminum electrolytic and
ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but
have lower capacitance density than other types. Tantalum
capacitors have the highest capacitance density but it is
important to only use types that have been surge tested
for use in switching power supplies. Aluminum electrolytic
capacitors have significantly higher ESR, but can be used
in cost-sensitive applications providing that consideration
is given to ripple current ratings and long-term reliability.
Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient and audible
piezoelectric effects. The high Q of ceramic capacitors with
trace inductance can also lead to significant ringing. When
used as input capacitors, care must be taken to ensure
that ringing from inrush currents and switching does not
pose an overvoltage hazard to the power switches and
controller. To dampen input voltage transients, add a small
5µF to 50µF aluminum electrolytic capacitor with an ESR in
the range of 0.5Ω to 2Ω. High performance through-hole
capacitors may also be used, but an additional ceramic
capacitor in parallel is recommended to reduce the effect
of their lead inductance.
Top MOSFET Driver Supply (CB, DB)
An external bootstrap capacitor, CB, connected to the BOOST
pin supplies the gate drive voltage for the topside MOSFET.
This capacitor is charged through diode DB from INTVCC
when the switch node is low. When the top MOSFET turns
on, the switch node rises to VIN and the BOOST pin rises
to approximately VIN + INTVCC. The boost capacitor needs
to store about 100 times the gate charge required by the
top MOSFET. In most applications an 0.1µF to 0.47µF, X5R
or X7R dielectric capacitor is adequate.
Discontinuous Mode Operation and FCB Pin
The FCB pin determines whether the bottom MOSFET
remains on when current reverses in the inductor. Tying
this pin above its 0.6V threshold enables discontinuous
operation where the bottom MOSFET turns off when inductor current reverses. The load current at which current
reverses and discontinuous operation begins depends on
the amplitude of the inductor ripple current and will vary
3609fb
14
LTC3609
Applications Information
Fault Conditions: Current Limit and Foldback
with changes in VIN. Tying the FCB pin below the 0.6V
threshold forces continuous synchronous operation, allowing current to reverse at light loads and maintaining
high frequency operation.
The LTC3609 has a current mode controller which inherently limits the cycle-by-cycle inductor current not only
in steady-state operation but also in transient. To further
limit current in the event of a short circuit to ground, the
LTC3609 includes foldback current limiting. If the output
falls by more than 25%, then the maximum sense voltage is
progressively lowered to about one sixth of its full value.
In addition to providing a logic input to force continuous
operation, the FCB pin provides a means to maintain a
flyback winding output when the primary is operating
in discontinuous mode. The secondary output VOUT2 is
normally set as shown in Figure 4 by the turns ratio N
of the transformer. However, if the controller goes into
discontinuous mode and halts switching due to a light
primary load current, then VOUT2 will droop. An external
resistor divider from VOUT2 to the FCB pin sets a minimum
voltage VOUT2(MIN) below which continuous operation is
forced until VOUT2 has risen above its minimum:
An internal P-channel low dropout regulator produces the
5V supply that powers the drivers and internal circuitry
within the LTC3609. The INTVCC pin can supply up to 50mA
RMS and must be bypassed to ground with a minimum of
4.7µF tantalum or ceramic capacitor. Good bypassing is
necessary to supply the high transient currents required
by the MOSFET gate drivers.
R4
VOUT2(MIN) = 0.6V 1+
R3
SW
GND
IN4148
NC
SGND
SVIN
EXTVCC
INTVCC
SW
PGND
PGND
PGND
PGND
INTVCC
NC
LTC3609
SW
SGND
FCB
PVIN
ITH
PVIN
VRNG
PVIN
PGOOD
PVIN
SGND
26
25
R4
24
23
22
21
20
OPTIONAL EXTVCC
CONNECTION
5V < VOUT2 < 7V
19
18
17
R3
16
15
SGND
PVIN
1
2
3
4
5
6
7
8
9 10 11 12 13 14
VON
52
SW
RUN/SS
51
ION
BOOST
50
SW
SGND
49
VFB
NC
48
NC
SW
SW
CIN
+
VIN
SW
PVIN
47
NC
PVIN
46
SW
PVIN
45
SGND
PVIN
44
PGND
PGND
43
SW
PVIN
+
42
PVIN
COUT
41
T1
1:N
PVIN
VOUT1
•
CSEC
1µF
+
PGND
40 39 38 37 36 35 34 33 32 31 30 29 28 27
VOUT2
•
INTVCC Regulator and EXTVCC Connection
3609 F04
SW
SGND
Figure 4. Secondary Output Loop and EXTVCC Connection
3609fb
15
LTC3609
Applications Information
The EXTVCC pin can be used to provide MOSFET gate drive
and control power from the output or another external
source during normal operation. Whenever the EXTVCC
pin is above 4.7V the internal 5V regulator is shut off and
an internal 50mA P-channel switch connects the EXTVCC
pin to INTVCC. INTVCC power is supplied from EXTVCC
until this pin drops below 4.5V. Do not apply more than
7V to the EXTVCC pin and ensure that EXTVCC ≤ VIN. The
following list summarizes the possible connections for
EXTVCC:
1. EXTVCC grounded. INTVCC is always powered from the
internal 5V regulator.
2. EXTVCC connected to an external supply. A high efficiency
supply compatible with the MOSFET gate drive requirements (typically 5V) can improve overall efficiency.
3. EXTVCC connected to an output derived boost network.
The low voltage output can be boosted using a charge
pump or flyback winding to greater than 4.7V. The system
will start-up using the internal linear regulator until the
boosted output supply is available.
Soft-Start and Latchoff with the RUN/SS Pin
The RUN/SS pin provides a means to shut down the LTC3609
as well as a timer for soft-start and overcurrent latchoff.
Pulling the RUN/SS pin below 0.8V puts the LTC3609 into
a low quiescent current shutdown (IQ COUT VOUT RSENSE (10 –4 [F/V s])
Generally 0.1µF is more than sufficient.
Overcurrent latchoff operation is not always needed or
desired. Load current is already limited during a short
circuit by the current foldback circuitry and latchoff operation can prove annoying during troubleshooting. The
feature can be overridden by adding a pull-up current
greater than 5µA to the RUN/SS pin. The additional current prevents the discharge of CSS during a fault and also
shortens the soft-start period. Using a resistor to VIN as
shown in Figure 5a is simple, but slightly increases shutdown current. Connecting a resistor to INTVCC as shown
in Figure 5b eliminates the additional shutdown current,
but requires a diode to isolate CSS. Any pull-up network
must be able to pull RUN/SS above the 4.2V maximum
threshold of the latchoff circuit and overcome the 4µA
maximum discharge current.
3609fb
16
LTC3609
Applications Information
INTVCC
RSS*
VIN
3.3V OR 5V
D1
RUN/SS
RSS*
CSS
D2*
RUN/SS
2N7002
CSS
3609 F05
*OPTIONAL TO OVERRIDE
OVERCURRENT LATCHOFF
(5a)
(5b)
Figure 5. RUN/SS Pin Interfacing with Latchoff Defeated
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Although all dissipative
elements in the circuit produce losses, four main sources
account for most of the losses in LTC3609 circuits:
I2R
losses. These arise from the resistance of the
1. DC
internal resistance of the MOSFETs, inductor and PC
board traces and cause the efficiency to drop at high
output currents. In continuous mode the average output
current flows through L, but is chopped between the top
and bottom MOSFETs. The DC I2R loss for one MOSFET
can simply be determined by [RDS(ON) + RL] • IO.
2. Transition loss. This loss arises from the brief amount
of time the top MOSFET spends in the saturated region
during switch node transitions. It depends upon the
input voltage, load current, driver strength and MOSFET
capacitance, among other factors. The loss is significant
at input voltages above 20V and can be estimated from:
Transition Loss ≅ (1.7A–1) VIN2 IOUT CRSS f
3. INTVCC current. This is the sum of the MOSFET driver
and control currents. This loss can be reduced by supplying INTVCC current through the EXTVCC pin from a high
efficiency source, such as an output derived boost network
or alternate supply if available.
4. CIN loss. The input capacitor has the difficult job of
filtering the large RMS input current to the regulator. It
must have a very low ESR to minimize the AC I2R loss and
sufficient capacitance to prevent the RMS current from
causing additional upstream losses in fuses or batteries.
Other losses, including COUT ESR loss, Schottky diode D1
conduction loss during dead time and inductor core loss
generally account for less than 2% additional loss.
When making adjustments to improve efficiency, the input
current is the best indicator of changes in efficiency. If you
make a change and the input current decreases, then the
efficiency has increased. If there is no change in input
current, then there is no change in efficiency.
Checking Transient Response
The regulator loop response can be checked by looking
at the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to ∆ILOAD (ESR), where ESR is the effective series
resistance of COUT. ∆ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the
regulator to return VOUT to its steady-state value. During
this recovery time, VOUT can be monitored for overshoot
or ringing that would indicate a stability problem. The ITH
pin external components shown in Figure 6 will provide
adequate compensation for most applications. For a
detailed explanation of switching control loop theory see
Application Note 76.
3609fb
17
LTC3609
Applications Information
Design Example
Next, set up VRNG voltage and check the ILIMIT. Tying VRNG
to GND will set the typical current limit to 9A, and tying
VRNG to 1.2V will result in a typical current around 14A.
CIN is chosen for an RMS current rating of about 5A at
85°C. The ceramic output capacitors are chosen for an
ESR of 0.002Ω to minimize output voltage changes due
to inductor ripple current and load steps. The ripple voltage is:
As a design example, take a supply with the following
specifications: VIN = 5V to 32V (12V nominal), VOUT =
2.5V ± 5%, IOUT(MAX) = 6A, f = 550kHz. First, calculate the
timing resistor with VON = VOUT:
2.5V
RON =
= 187k
2.4V )( 550kHz )(10pF )
(
and choose the inductor for about 40% ripple current at
the maximum VIN:
2.5V
2.5V
L=
=1.8µH
1−
(550kHz ) (0.4) (6A ) 32V
∆VOUT(RIPPLE) = ∆IL(MAX) (ESR)
= (2.4A) (0.002Ω) = 4.8mV
and a 0A to 6A load step will only cause an output
change of:
∆VOUT(STEP) = ∆ILOAD (ESR) = (6A) (0.002Ω) = 12mV
Selecting a standard value of 1.5µH results in a maximum
ripple current of:
2.5V
2.5V
ΔIL =
1–
= 2.4A
550kHz ) (1.5µH) 12V
(
An optional 22µF ceramic output capacitor is included
to minimize the effect of ESL in the output ripple. The
complete circuit is shown in Figure 6.
INTVCC
CF
0.1µF
50V
CVCC
4.7µF
6.3V
VIN
RF1
1Ω
EXTVCC
C4
0.01µF
SW
GND
50
51
52
NC
SGND
SVIN
EXTVCC
INTVCC
SW
PGND
PGND
PGND
PGND
PGND
PGND
INTVCC
NC
LTC3609
SW
SGND
PVIN
FCB
PVIN
ITH
PVIN
VRNG
PVIN
PGOOD
PVIN
SGND
1
2
3
4
5
6
7
8
9 10 11
CIN: MURATA GRM32ER71H475K
COUT: MURATA GRM435R60J107M
LI: CDEP851R2MC-50
KEEP POWER GROUND AND SIGNAL
GROUND SEPARATE. CONNECT AT
ONE POINT.
INTVCC
DB
CMDSH-3
CB1
0.22µF
SW
26
25
R1
9.53k
1%
24
23
22
VOUT
VIN
INTVCC
JP1
20
CC1
R5 1000pF
15.8k
19
18
17
16
15
CC2
100pF
RPG1
100k
SGND
CVON
0.1µF
RON
187k
1%
21
12 13 14
2Ω
SW
R2
30.1k
1%
SGND
49
SW
VON
CIN
4.7µF
50V
x2
48
ION
RUN/SS
VIN
SW
BOOST
VIN
5V TO 32V
VFB
SGND
47
SW
NC
46
NC
SW
45
SW
PVIN
GND
NC
PVIN
44
SW
PVIN
43
SGND
PVIN
+
SW
PVIN
COUT1
100µF
x2
42
PVIN
41
L1
1.2µH
PVIN
VOUT
2.5V AT
6A
PGND
40 39 38 37 36 35 34 33 32 31 30 29 28 27
VOUT
PGOOD
INTVCC
3609 F06
CSS
0.1µF
Figure 6. Design Example: 5V to 32V Input to 2.5V/6A at 550kHz
3609fb
18
LTC3609
Applications Information
How to Reduce SW Ringing
As with any switching regulator, there will be voltage ringing on the SW node, especially for high input voltages.
The ringing amplitude and duration is dependent on the
switching speed (gate drive), layout (parasitic inductance)
and MOSFET output capacitance. This ringing contributes
to the overall EMI, noise and high frequency ripple. One
way to reduce ringing is to optimize layout. A good layout
minimizes parasitic inductance. Adding RC snubbers from
SW to GND is also an effective way to reduce ringing. Finally,
adding a resistor in series with the BOOST pin will slow
down the MOSFET turn-on slew rate to dampen ringing,
but at the cost of reduced efficiency. Note that since the
IC is buffered from the high frequency transients by PCB
and bondwire inductances, the ringing by itself is normally
not a concern for controller reliability.
PC Board Layout Checklist
When laying out a PC board follow one of the two suggested approaches. The simple PC board layout requires
a dedicated ground plane layer. Also, for higher currents, a
multilayer board is recommended to help with heat sinking
of power components.
• The ground plane layer should not have any traces and
it should be as close as possible to the layer with the
LTC3609.
• Place CIN and COUT all in one compact area, close to
the LTC3609. It may help to have some components
on the bottom side of the board.
• Use a compact plane for the switch node (SW) to improve
cooling of the MOSFETs and to keep EMI down.
• Use planes for VIN and VOUT to maintain good voltage
filtering and to keep power losses low.
• Flood all unused areas on all layers with copper. Flooding with copper reduces the temperature rise of power
components. Connect these copper areas to any DC
net (VIN, VOUT, GND or to any other DC rail in your
system).
When laying out a printed circuit board without a ground
plane, use the following checklist to ensure proper operation of the controller. These items are also illustrated in
Figure 7.
• Segregate the signal and power grounds. All smallsignal components should return to the SGND pin at
one point, which is then tied to the PGND pin.
• Connect the input capacitor(s), CIN, close to the IC.
This capacitor carries the MOSFET AC current.
• Keep the high dV/dT SW, BOOST and TG nodes away
from sensitive small-signal nodes.
• Connect the INTVCC decoupling capacitor, CVCC, closely
to the INTVCC and PGND pins.
• Connect the top driver boost capacitor, CB, closely to
the BOOST and SW pins.
• Connect the VIN pin decoupling capacitor, CF , closely
to the VIN and PGND pins.
• Keep small-signal components close to the LTC3609.
• Ground connections (including LTC3609 SGND and
PGND) should be made through immediate vias to
the ground plane. Use several larger vias for power
components.
3609fb
19
LTC3609
Applications Information
CVCC SW
NC
SGND
SVIN
EXTVCC
INTVCC
SW
PGND
PGND
PGND
PGND
INTVCC
NC
LTC3609
SW
SGND
FCB
PVIN
ITH
PVIN
VRNG
PVIN
PGOOD
PVIN
SGND
2
3
4
5
6
7
8
9 10 11 12 13 14
VON
1
DB
CB
26
25
R1
R2
24
23
RON
22
21
20
19
RC
18
CC1
17
16
15
CC2
SGND
PVIN
RUN/SS
52
SW
BOOST
51
ION
SGND
50
SW
NC
49
VFB
SW
CIN
SW
PVIN
48
NC
PVIN
47
SW
PVIN
46
NC
PVIN
45
SW
PVIN
44
PGND
PGND
43
SGND
PVIN
42
SW
PVIN
41
VOUT
PGND
40 39 38 37 36 35 34 33 32 31 30 29 28 27
COUT
CSS
RF
KEEP POWER GROUND AND SIGNAL
GROUND SEPARATE. CONNECT AT
ONE POINT.
3609 F07
Figure 7. LTC3609 Layout Diagram
3609fb
20
LTC3609
Typical Applications
3.6V Input to 1.5V/6A at 750kHz
INTVCC
VBIAS
5V
CF
0.1µF
50V
CVCC
4.7µF
6.3V
EXTVCC
C4
0.01µF
SW
GND
R2
60.4k
1%
52
NC
SGND
SVIN
EXTVCC
INTVCC
SW
INTVCC
PGND
PGND
PGND
PGND
PGND
PGND
SW
SGND
PVIN
FCB
PVIN
ITH
PVIN
VRNG
PVIN
PGOOD
PVIN
SGND
1
2
3
4
5
6
7
8
9 10 11
KEEP POWER GROUND AND SIGNAL
GROUND SEPARATE. CONNECT AT
ONE POINT.
INTVCC
DB
CMDSH-3
24
RON
113k
1%
23
22
INTVCC
VIN
21
JP1
20
19
CC1
1500pF
R5
8.45k
18
17
16
15
SGND
CVON
0.1µF
CC2
100pF
RPG1
100k
12 13 14
CB1
0.22µF
SW
26
25
2Ω
SW
CIN: MURATA GRM32ER71H475K
COUT: MURATA GRM435R60J167M
LI: CDEP850R5MC-125
VOUT
R1
40.2k
1%
SGND
51
NC
LTC3609
VON
50
SW
RUN/SS
49
ION
BOOST
48
SW
SGND
CIN
4.7µF
50V
x2
VFB
NC
VIN
3.6V
VIN
SW
SW
47
NC
PVIN
46
NC
SW
PVIN
45
SW
PVIN
44
SGND
PVIN
43
SW
PVIN
COUT1
100µF
x2
42
PVIN
GND
41
L1
0.5µH
PVIN
VOUT
1.5V AT
6A
PGND
40 39 38 37 36 35 34 33 32 31 30 29 28 27
PGOOD
INTVCC
VOUT
3609 TA02
CSS
0.1µF
Transient Response
Efficiency vs Load Current
100
95
IL
5A/DIV
EFFICIENCY (%)
90
VOUT
200mV/DIV
20µs/DIV
LOAD STEP 1A TO 5A
VIN = 3.6V
VOUT = 1.5V
FCB = 0V
3609 TA02b
85
80
75
70
65
60
55
50
0.01
VIN = 3.6V
FREQUENCY = 750kHz
0.1
1
LOAD CURRENT (A)
10
3609 TA02c
3609fb
21
LTC3609
Typical Applications
5V to 32V Input to 1.2V/6A at 550kHz
INTVCC
CF
0.1µF
25V
CVCC
4.7µF
6.3V
VIN
RF1
1Ω
EXTVCC
C4
0.01µF
SW
GND
50
51
52
NC
SGND
SVIN
EXTVCC
INTVCC
SW
INTVCC
PGND
PGND
PGND
PGND
PGND
PGND
SW
SGND
PVIN
FCB
PVIN
ITH
PVIN
VRNG
PVIN
PGOOD
PVIN
SGND
1
2
3
4
5
6
7
8
9 10 11
C5: TAIYO YUDEN JMK316BJ226ML-T
CIN: MURATA GRM32ER71H475K
COUT: MURATA GRM435R60J167M
LI: CDEP850R8MC-88
INTVCC
DB
CMDSH-3
KEEP POWER GROUND AND SIGNAL
GROUND SEPARATE. CONNECT AT
ONE POINT.
25
R1
60.4k
1%
24
23
22
RON
182k
1%
VOUT
JP1
20
19
R5
8.45k
18
CC1
1500pF
17
16
R3
0Ω
15
CC2
100pF
RPG1
100k
PGOOD
INTVCC
VOUT
CVON
0.1µF
INTVCC
VIN
21
SGND
CB1
0.22µF
SW
26
12 13 14
2Ω
SW
R2
60.4k
1%
SGND
49
NC
LTC3609
VON
48
SW
RUN/SS
CIN
4.7µF
50V
x2
ION
BOOST
VIN
VIN
5V TO 32V
SW
SGND
47
VFB
NC
46
SW
SW
45
NC
PVIN
GND
SW
PVIN
44
NC
PVIN
43
SW
PVIN
42
SGND
PVIN
COUT1
100µF
x2
SW
PVIN
41
L1
0.8µH
PVIN
VOUT
1.2V AT
6A
PGND
40 39 38 37 36 35 34 33 32 31 30 29 28 27
3609 TA03
CSS
0.1µF
Transient Response
Efficiency vs Load Current
90
85
IL
5A/DIV
VIN = 12V
FREQUENCY = 550kHz
EFFICIENCY (%)
80
VOUT
200mV/DIV
75
70
65
60
20µs/DIV
LOAD STEP 1A TO 6A
VIN = 12V
VOUT = 1.2V
FCB = 0V
3609 TA02b
55
50
0.01
0.1
1
LOAD CURRENT (A)
10
3609 TA02c
3609fb
22
LTC3609
Typical Applications
5V to 32V Input to 1.8V/6A All Ceramic 1MHz
INTVCC
CF
0.1µF
50V
CVCC
4.7µF
6.3V
VIN
RF1
1Ω
EXTVCC
C4
0.01µF
SW
GND
51
52
NC
SGND
SVIN
EXTVCC
INTVCC
SW
PGND
PGND
PGND
PGND
PGND
PGND
INTVCC
NC
LTC3609
SW
SGND
PVIN
FCB
PVIN
ITH
PVIN
VRNG
PVIN
PGOOD
PVIN
SGND
1
2
3
4
5
6
7
8
9 10 11
INTVCC
DB
CMDSH-3
KEEP POWER GROUND AND SIGNAL
GROUND SEPARATE. CONNECT AT
ONE POINT.
R1
10k
1%
24
23
22
CVON
0.1µF
VOUT
RON
102k
1%
INTVCC
VIN
21
JP1
20
19
R5
5.76k
18
CC1
1500pF
17
16
15
CC2
100pF
RPG1
100k
SGND
CB1
0.22µF
SW
26
25
12 13 14
2Ω
SW
C5: TAIYO YUDEN JMK316BJ226ML-T
CIN: MURATA GRM32ER71H475K
COUT: MURATA GRM32ER60J107M
LI: 1HLP25CZERR80M01
R2
20k
1%
SGND
50
SW
VON
49
ION
RUN/SS
48
SW
BOOST
CIN
4.7µF
x2
VFB
SGND
VIN
5V TO 32V
VIN
SW
NC
47
NC
SW
46
SW
PVIN
45
NC
PVIN
GND
SW
PVIN
44
SGND
PVIN
43
SW
PVIN
COUT1
100µF
x2
42
PVIN
41
L1
0.47µH
PVIN
VOUT
1.8V AT
6A
PGND
40 39 38 37 36 35 34 33 32 31 30 29 28 27
PGOOD
INTVCC
VOUT
3609 TA04
CSS
0.1µF
Transient Response
Efficiency vs Load Current
100
90
IL
5A/DIV
EFFICIENCY (%)
80
VOUT
200mV/DIV
20µs/DIV
LOAD STEP 500mA TO 4A
VIN = 12V
VOUT = 1.8V
FCB = 0V
3609 TA04b
70
60
50
40
30
20
10
0
10
100
1000
LOAD CURRENT (mA)
10000
3609 TA04c
3609fb
23
LTC3609
Package Description
WKG Package
52-Lead QFN Multipad (7mm × 8mm)
(Reference LTC DWG # 05-08-1768 Rev Ø)
SEATING PLANE
A
7.00
BSC
0.00 – 0.05
2.625 REF
41
B
PAD 1
CORNER
4
2.90 REF
0.50 BSC
1
40
bbb M C A B
7
8.00
BSC
PIN 1 ID
52
3.40 REF
3.90 ± 0.10
3.20 ± 0.10
2.025
± 0.10
2.925 ± 0.10
3.40 REF
33
8
32
9
1.00 REF
10
aaa C 2x
NX b
TOP VIEW
0.90 ± 0.10
NX
0.08 C
// ccc C
8
7.50 ± 0.05
2.90 REF
0.50 BSC
1.35
± 0.10
1.775
REF
15
0.25 ± 0.05
BOTTOM VIEW
(BOTTOM METALLIZATION DETAILS)
MLP52 QFN REV Ø 0807
NOTE:
1. DIMENSIONING AND TOLERANCING CONFORM TO ASME Y14.5M-1994
2. ALL DIMENSIONS ARE IN MILLIMETERS, ANGLES ARE IN DEGREES (°)
3. N IS THE TOTAL NUMBER OF TERMINALS
4
2.025
± 0.10
3.40 REF
2.925 ± 0.10
3.90 ± 0.10
8.50 ± 0.05
4.275 ± 0.10
PACKAGE
OUTLINE
0.40 ± 0.10
1.775
REF
1.35
± 0.10
THE LOCATION OF THE TERMINAL #1 IDENTIFIER AND TERMINAL NUMBERING
CONVENTION CONFORMS TO JEDEC PUBLICATION 95 SPP-002
5. ND AND NE REFER TO THE NUMBER OF TERMINALS ON EACH D AND E SIDE RESPECTIVELY
6. NJR REFER TO NON JEDEC REGISTERED
7
DIMENSION b APPLIES TO METALLIZED TERMINAL AND IS MEASURED
BETWEEN 0.20mm AND 0.30mm FROM THE TERMINAL TIP. IF THE TERMINAL
HAS THE OPTIONAL RADIUS ON THE OTHER END OF THE TERMINAL, THE
DIMENSION b SHOULD NOT BE MEASURED IN THAT RADIUS AREA.
8
COPLANARITY APPLIES TO THE TERMINALS AND ALL OTHER SURFACE
METALLIZATION
9
DRAWING SHOWN ARE FOR ILLUSTRATION ONLY
1.00 REF
0.25 ± 0.05
0.40 ± 0.10
19
2.625 REF
PIN 1
2.25 ± 0.10
14
26
9
3.40 REF
2.25 ± 0.10
27
0.580 ± 0.10
aaa C 2x
3.20 ± 0.10
4.275 ± 0.10
SYMBOL TOLERANCE
aaa
0.15
bbb
0.10
ccc
0.10
RECOMMENDED SOLDER PAD LAYOUT
TOP VIEW
3609fb
24
LTC3609
Revision History
REV
DATE
B
06 /10
(Revision history begins at Rev B)
DESCRIPTION
PAGE NUMBER
Updated SW voltage range in Absolute Maximum Ratings.
2
Note 4 updated.
4
3609fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
25
LTC3609
Typical Application
INTVCC
CF
0.1µF
50V
CVCC
4.7µF
6.3V
VIN
RF1
1Ω
EXTVCC
C4
0.01µF
SW
GND
50
51
52
NC
SGND
SVIN
EXTVCC
INTVCC
SW
PGND
PGND
PGND
PGND
PGND
PGND
INTVCC
NC
LTC3609
SW
SGND
PVIN
FCB
PVIN
ITH
PVIN
VRNG
PVIN
PGOOD
PVIN
SGND
1
2
3
4
5
6
7
8
9 10 11
CIN: MURATA GRM31CR71H475K
COUT: SANYO 16SVP180MX
LI: CDEP4R3MC-88
INTVCC
DB
CMDSH-3
KEEP POWER GROUND AND SIGNAL
GROUND SEPARATE. CONNECT AT
ONE POINT.
R1
3.16k
RON
1%
1M
1%
24
23
22
VOUT
INTVCC
VIN
21
JP1
20
19
R5
24.3k
18
CC1
1000pF
17
16
15
RPG1
100k
12 13 14
CB1
0.22µF
SW
26
25
PGOOD
SGND
2Ω
SW
R2
60.4k
1%
SGND
49
SW
VON
48
ION
RUN/SS
VIN
CIN
4.7µF
x2
VFB
SW
BOOST
VIN
14V TO 32V
SW
SGND
47
NC
NC
46
SW
SW
45
NC
PVIN
GND
SW
PVIN
44
SGND
PVIN
43
SW
PVIN
42
PVIN
COUT1
180µF
16V
L1
4.3µH
PVIN
41
+
PVIN
VOUT
12V AT
4A
PGND
40 39 38 37 36 35 34 33 32 31 30 29 28 27
CVON
0.1µF
INTVCC
CC2
100pF
INTVCC
3609 TA05
CSS
0.1µF
Figure 8. 14V to 32V Input to 12V/4A at 500kHz
Related Parts
PART NUMBER DESCRIPTION
COMMENTS
LTC3602
2.5A (IOUT), 3MHz, Synchronous Step-Down DC/DC
Converter
95% Efficiency, VIN: 4.5V to 10V, VOUT(MIN) = 0.6V, IQ = 75µA, ISD
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