LTC3631
High Efficiency, High Voltage
100mA Synchronous
Step-Down Converter
Description
Features
Wide Input Voltage Range: Operation from 4.5V to 45V
Overvoltage Lockout Provides Protection Up to 60V
Internal High Side and Low Side Power Switches
No Compensation Required
100mA Output Current
Low Dropout Operation: 100% Duty Cycle
Low Quiescent Current: 12µA
0.8V ±1% Feedback Voltage Reference
Adjustable Peak Current Limit
Internal and External Soft-Start
Precise RUN Pin Threshold with Adjustable
Hysteresis
n 3.3V, 5V and Adjustable Output Versions
n Only Three External Components Required for Fixed
Output Versions
n Low Profile (0.75mm) 3mm × 3mm DFN and
Thermally-Enhanced MS8E Packages
n
n
n
n
n
n
n
n
n
n
n
Applications
n
n
n
n
n
n
4mA to 20mA Current Loops
Industrial Control Supplies
Distributed Power Systems
Portable Instruments
Battery-Operated Devices
Automotive Power Systems
The LTC®3631 is a high voltage, high efficiency step-down
DC/DC converter with internal high side and synchronous
power switches that draws only 12μA typical DC supply
current at no load while maintaining output voltage
regulation.
The LTC3631 can supply up to 100mA load current and
features a programmable peak current limit that provides
a simple method for optimizing efficiency in lower current
applications. The LTC3631’s combination of Burst Mode®
operation, integrated power switches, low quiescent current, and programmable peak current limit provides high
efficiency over a broad range of load currents.
With its wide 4.5V to 45V input range and internal overvoltage monitor capable of protecting the part from 60V surges,
the LTC3631 is a robust converter suited for regulating a
wide variety of power sources. Additionally, the LTC3631
includes a precise run threshold and a soft‑start feature
to guarantee that power system start-up is well-controlled
in any environment.
The LTC3631 is available in the thermally enhanced
3mm × 3mm DFN and MS8E packages.
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
Typical Application
Efficiency and Power Loss vs Load Current
100
90
5V, 100mA Step-Down Converter
3631 TA01a
80
VOUT
5V
10µF 100mA
EFFICIENCY (%)
VIN
SW
LTC3631-5
RUN
VOUT
SS
HYST
ISET
GND
1000
70
60
50
100
POWER LOSS
10
POWER LOSS (mW)
VIN
5V TO 45V
2.2µF
100µH
EFFICIENCY
40
30
20
0.1
VIN = 12V
VIN = 36V
1
10
LOAD CURRENT (mA)
1
100
3631 TA01b
3631fe
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1
LTC3631
Absolute Maximum Ratings (Note 1)
VIN Supply Voltage...................................... –0.3V to 60V
SW Voltage (DC)............................ –0.3V to (VIN + 0.3V)
RUN Voltage............................................... –0.3V to 60V
HYST, ISET, SS Voltages................................ –0.3V to 6V
VFB................................................................ –0.3V to 6V
VOUT (Fixed Output Versions)........................ –0.3V to 6V
Operating Junction Temperature Range
(Note 2)................................................... –40°C to 125°C
Storage Temperature Range.................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
MS8E................................................................. 300°C
Pin Configuration
TOP VIEW
TOP VIEW
SW
VIN
ISET
SS
1
2
3
4
9
GND
8
7
6
5
SW 1
GND
HYST
VOUT/VFB
RUN
VIN 2
ISET 3
9
GND
SS 4
MS8E PACKAGE
8-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 40°C/W, θJC = 5°-10°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
8
GND
7
HYST
6
VOUT/VFB
5
RUN
DD PACKAGE
8-LEAD (3mm × 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 43°C/W, θJC = 3°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
order information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3631EMS8E#PBF
LTC3631EMS8E#TRPBF
LTFDT
8-Lead Plastic MSOP
–40°C to 125°C
LTC3631EMS8E-3.3#PBF
LTC3631EMS8E-3.3#TRPBF
LTFFP
8-Lead Plastic MSOP
–40°C to 125°C
LTC3631EMS8E-5#PBF
LTC3631EMS8E-5#TRPBF
LTFFR
8-Lead Plastic MSOP
–40°C to 125°C
LTC3631IMS8E#PBF
LTC3631IMS8E#TRPBF
LTFDT
8-Lead Plastic MSOP
–40°C to 125°C
LTC3631IMS8E-3.3#PBF
LTC3631IMS8E-3.3#TRPBF
LTFFP
8-Lead Plastic MSOP
–40°C to 125°C
LTC3631IMS8E-5#PBF
LTC3631IMS8E-5#TRPBF
LTFFR
8-Lead Plastic MSOP
–40°C to 125°C
LTC3631EDD#PBF
LTC3631EDD#TRPBF
LFDV
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3631EDD-3.3#PBF
LTC3631EDD-3.3#TRPBF
LFFN
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3631EDD-5#PBF
LTC3631EDD-5#TRPBF
LFFQ
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3631IDD#PBF
LTC3631IDD#TRPBF
LFDV
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3631IDD-3.3#PBF
LTC3631IDD-3.3#TRPBF
LFFN
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3631IDD-5#PBF
LTC3631IDD-5#TRPBF
LFFQ
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3631fe
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LTC3631
Electrical
Characteristics
The
l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are for TA = 25°C (Note 2). VIN = 10V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input Supply (VIN)
VIN
Input Voltage Operating Range
4.5
UVLO
VIN Undervoltage Lockout
VIN Rising
VIN Falling
Hysteresis
OVLO
VIN Overvoltage Lockout
VIN Rising
VIN Falling
Hysteresis
IQ
DC Supply Current (Note 3)
Active Mode
Sleep Mode
Shutdown Mode
l
l
45
V
3.80
3.75
4.15
4.00
150
4.50
4.35
47
45
50
48
2
52
50
V
V
V
125
12
3
220
22
6
µA
µA
µA
VRUN = 0V
V
V
mV
Output Supply (VOUT/VFB)
VOUT
Output Voltage Trip Thresholds
LTC3631-3.3V, VOUT Rising
LTC3631-3.3V, VOUT Falling
l
l
3.260
3.240
3.310
3.290
3.360
3.340
V
V
LTC3631-5V, VOUT Rising
LTC3631-5V, VOUT Falling
l
l
4.940
4.910
5.015
4.985
5.090
5.060
V
V
VFB Rising
l
0.792
0.800
0.808
l
3
5
7
mV
–10
0
10
nA
VFB
Feedback Comparator Trip Voltage
VHYST
Feedback Comparator Hysteresis
IFB
Feedback Pin Current
Adjustable Output Version, VFB = 1V
∆VLINEREG
Feedback Voltage Line Regulation
VIN = 4.5V to 45V
LTC3631-5, VIN = 6V to 45V
VRUN
Run Pin Threshold Voltage
RUN Rising
RUN Falling
Hysteresis
1.17
1.06
1.21
1.10
110
1.25
1.14
V
V
mV
IRUN
Run Pin Leakage Current
RUN = 1.3V
–10
0
10
nA
0.07
0.1
V
0
10
nA
5.5
6.5
0.001
V
%/V
Operation
VHYSTL
Hysteresis Pin Voltage Low
RUN < 1V, IHYST = 1mA
IHYST
Hysteresis Pin Leakage Current
VHYST = 1.3V
–10
ISS
Soft-Start Pin Pull-Up Current
VSS < 1.5V
4.5
tINTSS
Internal Soft-Start Time
SS Pin Floating
IPEAK
Peak Current Trip Threshold
ISET Floating
500k Resistor from ISET to GND
ISET Shorted to GND
RON
Power Switch On-Resistance
Top Switch
Bottom Switch
0.75
ISW = –25mA
ISW = 25mA
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3631 is tested under pulsed load conditions such that
TJ ≈ TA. The LTC3631E is guaranteed to meet specifications from
0°C to 85°C junction temperature. Specifications over the –40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3631I is guaranteed over the full –40°C to 125°C operating junction
l
200
40
225
120
50
3.0
1.5
µA
ms
280
65
mA
mA
mA
Ω
Ω
temperature range. Note that the maximum ambient temperature
consistent with these specifications is determined by specific operating
conditions in conjunction with board layout, the rated package thermal
impedance and other environmental factors. The junction temperature
(TJ, in °C) is calculated from the ambient temperature (TA, in °C) and power
dissipation (PD, in Watts) according to the formula:
TJ = TA + (PD • θJA)
where θJA (in °C/W) is the package thermal impedance.
Note 3: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
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LTC3631
Typical Performance Characteristics
Efficiency vs Load Current,
VOUT = 5V
90
VOUT = 5V
FIGURE 10 CIRCUIT
95
VIN = 12V
VOUT = 2.5V
85 FIGURE 10 CIRCUIT
VIN = 12V
VIN = 36V
80
VIN = 24V
75
80
70
65
65
60
0.1
1
10
LOAD CURRENT (mA)
VIN = 36V
75
70
VIN = 24V
ILOAD = 100mA
1
10
LOAD CURRENT (mA)
ILOAD = 1mA
ILOAD = 100mA
FIGURE 10 CIRCUIT
5.03
0
5.02
5.01
5.00
4.99
4.98
4.97
10
15
30
25
35
20
INPUT VOLTAGE (V)
40
45
–0.20
5
10
15
20 25 30 35
INPUT VOLTAGE (V)
3631 G04
0.799
20
50
80
TEMPERATURE (°C)
4.96
110
LTC1144 • TPC06
10 20 30 40 50 60 70 80 90 100
LOAD CURRENT (mA)
0
3631 G06
5.6
250
VIN = 10V
PEAK CURRENT TRIP THRESHOLD (mA)
FEEDBACK COMPARATOR HYSTERESIS (mV)
0.800
–10
45
Feedback Comparator Hysteresis
vs Temperature
VIN = 10V
0.798
–40
40
3631 G05
Feedback Comparator Trip
Voltage vs Temperature
FEEDBACK COMPARATOR TRIP VOLTAGE (V)
VIN = 10
VOUT = 5V
FIGURE 10 CIRCUIT
5.04
70
0.801
100
Load Regulation
OUTPUT VOLTAGE (V)
∆VOUT/VOUT (%)
ILOAD = 10mA
1
10
LOAD CURRENT (mA)
3631 G03
5.05
–0.10
65
50
0.1
100
0.10
85
75
65
Line Regulation
0.20
VOUT = 5V
FIGURE 10 CIRCUIT
80
70
3631 G02
Efficiency vs Input Voltage
90
VIN = 24V
55
3631 G01
95
VIN = 36V
75
60
60
0.1
100
VIN = 12V
80
85
EFFICIENCY (%)
EFFICIENCY (%)
90
VOUT = 3.3V
FIGURE 10 CIRCUIT
90
85
EFFICIENCY (%)
Efficiency vs Load Current,
VOUT = 2.5V
EFFICIENCY (%)
95
Efficiency vs Load Current,
VOUT = 3.3V
5.4
5.2
5.0
4.8
4.6
4.4
–40
–10
20
50
80
TEMPERATURE (°C)
110
3631 G08
225
Peak Current Trip Threshold
vs Temperature and ISET
VIN = 10V
ISET = OPEN
200
175
150
125
RSET = 500k
100
75
ISET = GND
50
25
0
–40
–10
50
80
20
TEMPERATURE (°C)
110
3631 G09
3631fe
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LTC3631
Typical Performance Characteristics
Peak Current Trip Threshold
vs Input Voltage
250
200
180
160
140
120
100
80
60
40
20
200
0
400
600
800
RISET (kΩ)
1000
14
ISET OPEN
225
12
200
175
150
RSET = 500k
125
100
75
ISET = GND
50
0
1200
0
5
SHUTDOWN
4
0
10 15 20 25 30 35 40 45 50
INPUT VOLTAGE (V)
5
15
25
35
45
INPUT VOLTAGE (V)
3631 G12
Switch On-Resistance
vs Temperature
5
4.5
VIN = 10V
VIN = 10V
SLEEP
10
8
6
4
SHUTDOWN
2
SWITCH 0N-RESISTANCE (Ω)
4.0
SWITCH ON-RESISTANCE (Ω)
VIN SUPPLY CURRENT (µA)
6
Switch On-Resistance
vs Input Voltage
12
3.5
TOP
3.0
2.5
2.0
BOTTOM
1.5
1.0
4
TOP
3
BOTTOM
2
1
0.5
0
–40
–10
20
50
80
TEMPERATURE (°C)
0
110
0
10
30
20
INPUT VOLTAGE (V)
50
80
20
TEMPERATURE (°C)
110
Operating Waveforms
1.300
RUN COMPARATOR THRESHOLD (V)
1.250
0.5
0.4
SWITCH
VOLTAGE
20V/DIV
RISING
1.200
OUTPUT
VOLTAGE
50mV/DIV
1.150
0.3
0.2
FALLING
1.100
SW = 0V
50
80
20
TEMPERATURE (°C)
INDUCTOR
CURRENT
100mA/DIV
1.050
SW = 45V
–10
–10
3631 G15
RUN Comparator Threshold
Voltages vs Temperature
VIN = 45V
0.1
0
–40
3631 G14
Switch Leakage Current
vs Temperature
0
–40
50
40
3631 G13
SWITCH LEAKAGE CURRENT (µA)
8
3631 G11
Quiescent Supply Current
vs Temperature
0.6
SLEEP
10
2
25
3631 G10
14
Quiescent Supply Current
vs Input Voltage
VIN SUPPLY CURRENT (µA)
VIN = 10V
220
PEAK CURRENT TRIP THRESHOLD (mA)
PEAK CURRENT TRIP THRESHOLD (mA)
240
Peak Current Trip Threshold
vs RISET
110
3631 G16
1.000
–40
–10
20
50
80
TEMPERATURE (°C)
110
3631 G17
VIN = 36V
VOUT = 5V
ILOAD = 35mA
20µs/DIV
3631 G18
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5
LTC3631
Typical Performance Characteristics
Soft-Start Waveform
Load Step Transient Response
Short-Circuit Response
OUTPUT
VOLTAGE
50mV/DIV
OUTPUT
VOLTAGE
2V/DIV
LOAD
CURRENT
50mA/DIV
INDUCTOR
CURRENT
100mA/DIV
OUTPUT
VOLTAGE
1V/DIV
CSS = 0.047µF
5ms/DIV
3631 G19
VIN = 24V
VOUT = 5V
1ms/DIV
3631 G20
VIN = 10V
VOUT = 5V
200µs/DIV
3631 G21
Pin Functions
SW (Pin 1): Switch Node Connection to Inductor. This
pin connects to the drains of the internal power MOSFET
switches.
VIN (Pin 2): Main Supply Pin. A ceramic bypass capacitor
should be tied between this pin and GND (Pin 8).
ISET (Pin 3): Peak Current Set Input. A resistor from this
pin to ground sets the peak current trip threshold. Leave
floating for the maximum peak current (225mA). Short
this pin to ground for the minimum peak current (50mA).
A 1µA current is sourced out of this pin.
SS (Pin 4): Soft-Start Control Input. A capacitor to ground
at this pin sets the ramp time to full current output during start-up. A 5µA current is sourced out of this pin. If
left floating, the ramp time defaults to an internal 0.75ms
soft-start.
VOUT/VFB (Pin 6): Output Voltage Feedback. For the fixed
output versions, connect this pin to the output supply. For
the adjustable version, an external resistive divider should
be used to divide the output voltage down for comparison
to the 0.8V reference.
HYST (Pin 7): Run Hysteresis Open-Drain Logic Output.
This pin is pulled to ground when RUN (Pin 5) is below
1.2V. This pin can be used to adjust the RUN pin hysteresis.
See Applications Information.
GND (Pin 8, Exposed Pad Pin 9): Ground. The exposed
pad must be soldered to the printed circuit board ground
plane for optimal electrical and thermal performance.
RUN (Pin 5): Run Control Input. A voltage on this pin
above 1.2V enables normal operation. Forcing this pin
below 0.7V shuts down the LTC3631, reducing quiescent
current to approximately 3µA.
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LTC3631
Block Diagram
VIN
1µA
ISET
2
C2
3
PEAK CURRENT
COMPARATOR
–
+
RUN
+
5
1.2V
LOGIC
AND
SHOOTTHROUGH
PREVENTION
–
SW
L1
VOUT
1
C1
HYST
7
+
REVERSE CURRENT
COMPARATOR
FEEDBACK
COMPARATOR
GND
GND
8
9
–
VOLTAGE
REFERENCE
+
+
–
0.800V
5µA
SS
4
R1
R2
PART
NUMBER
R1
R2
LTC3631
LTC3631-3.3
LTC3631-5
0
2.5M
4.2M
∞
800k
800k
VOUT/VFB
6
3631 BD
IMPLEMENT DIVIDER
EXTERNALLY FOR
ADJUSTABLE VERSION
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7
LTC3631
Operation (Refer to Block Diagram)
The LTC3631 is a step-down DC/DC converter with internal
power switches that uses Burst Mode control, combining low quiescent current with high switching frequency,
which results in high efficiency across a wide range of
load currents. Burst Mode operation functions by using
short “burst” cycles to ramp the inductor current through
the internal power switches, followed by a sleep cycle
where the power switches are off and the load current is
supplied by the output capacitor. During the sleep cycle,
the LTC3631 draws only 12µA of supply current. At light
loads, the burst cycles are a small percentage of the total
cycle time which minimizes the average supply current,
greatly improving efficiency.
greater than the average load current. For this architecture,
the maximum average output current is equal to half of
the peak current.
The hysteretic nature of this control architecture results
in a switching frequency that is a function of the input
voltage, output voltage and inductor value. This behavior
provides inherent short‑circuit protection. If the output
is shorted to ground, the inductor current will decay very
slowly during a single switching cycle. Since the high side
switch turns on only when the inductor current is near
zero, the LTC3631 inherently switches at a lower frequency
during start-up or short-circuit conditions.
Start-Up and Shutdown
Main Control Loop
The feedback comparator monitors the voltage on the VFB
pin and compares it to an internal 800mV reference. If
this voltage is greater than the reference, the comparator
activates a sleep mode in which the power switches and
current comparators are disabled, reducing the VIN pin
supply current to only 12µA. As the load current discharges
the output capacitor, the voltage on the VFB pin decreases.
When this voltage falls 5mV below the 800mV reference,
the feedback comparator trips and enables burst cycles.
At the beginning of the burst cycle, the internal high side
power switch (P-channel MOSFET) is turned on and the
inductor current begins to ramp up. The inductor current
increases until either the current exceeds the peak current comparator threshold or the voltage on the VFB pin
exceeds 800mV, at which time the high side power switch
is turned off, and the low side power switch (N-channel
MOSFET) turns on. The inductor current ramps down until
the reverse current comparator trips, signaling that the
current is close to zero. If the voltage on the VFB pin is
still less than the 800mV reference, the high side power
switch is turned on again and another cycle commences.
The average current during a burst cycle will normally be
If the voltage on the RUN pin is less than 0.7V, the LTC3631
enters a shutdown mode in which all internal circuitry is
disabled, reducing the DC supply current to 3µA. When the
voltage on the RUN pin exceeds 1.21V, normal operation of
the main control loop is enabled. The RUN pin comparator
has 110mV of internal hysteresis, and therefore must fall
below 1.1V to disable the main control loop.
The HYST pin provides an added degree of flexibility for
the RUN pin operation. This open-drain output is pulled
to ground whenever the RUN comparator is not tripped,
signaling that the LTC3631 is not in normal operation. In
applications where the RUN pin is used to monitor the
VIN voltage through an external resistive divider, the HYST
pin can be used to increase the effective RUN comparator
hysteresis.
An internal 1ms soft-start function limits the ramp rate of
the output voltage on start-up to prevent excessive input
supply droop. If a longer ramp time and consequently less
supply droop is desired, a capacitor can be placed from the
SS pin to ground. The 5µA current that is sourced out of
this pin will create a smooth voltage ramp on the capacitor.
If this ramp rate is slower than the internal 1ms soft-start,
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LTC3631
Operation (Refer to Block Diagram)
then the output voltage will be limited by the ramp rate
on the SS pin instead. The internal and external soft-start
functions are reset on start-up and after an undervoltage
or overvoltage event on the input supply.
from the ISET pin to ground. The 1µA current sourced out
of this pin through the resistor generates a voltage that is
translated into an offset in the peak current comparator,
which limits the peak inductor current.
In order to ensure a smooth start-up transition in any
application, the internal soft‑start also ramps the peak
inductor current from 50mA during its 1ms ramp time to
the set peak current threshold. The external ramp on the
SS pin does not limit the peak inductor current during
start-up; however, placing a capacitor from the ISET pin
to ground does provide this capability.
Input Undervoltage and Overvoltage Lockout
Peak Inductor Current Programming
The offset of the peak current comparator nominally
provides a peak inductor current of 225mA. This peak
inductor current can be adjusted by placing a resistor
The LTC3631 implements a protection feature which disables switching when the input voltage is not within the
4.5V to 45V operating range. If VIN falls below 4V typical
(4.35V maximum), an undervoltage detector disables
switching. Similarly, if VIN rises above 50V typical (47V
minimum), an overvoltage detector disables switching.
When switching is disabled, the LTC3631 can safely sustain
input voltages up to the absolute maximum rating of 60V.
Switching is enabled when the input voltage returns to the
4.5V to 45V operating range.
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9
LTC3631
Applications Information
The basic LTC3631 application circuit is shown on the front
page of this data sheet. External component selection is
determined by the maximum load current requirement and
begins with the selection of the peak current programming
resistor, RISET. The inductor value L can then be determined,
followed by capacitors CIN and COUT.
Peak Current Resistor Selection
The peak current comparator has a maximum current limit
of 225mA nominally, which results in a maximum average current of 112mA. For applications that demand less
current, the peak current threshold can be reduced to as
little as 50mA. This lower peak current allows the use of
lower value, smaller components (input capacitor, output
capacitor and inductor), resulting in lower input supply
ripple and a smaller overall DC/DC converter.
The threshold can be easily programmed with an appropriately chosen resistor (RISET) between the ISET pin
and ground. The value of resistor for a particular peak
current can be computed by using Figure 1 or the following equation:
RISET = IPEAK • 4.5 • 106
where 50mA < IPEAK < 225mA.
The peak current is internally limited to be within the
range of 50mA to 225mA. Shorting the ISET pin to ground
programs the current limit to 50mA, and leaving it floating
sets the current limit to the maximum value of 225mA.
When selecting this resistor value, be aware that the
1100
1000
900
RISET (kΩ)
800
700
600
500
400
300
200
100
0
20
30 40 50 60 70 80 90
MAXIMUM LOAD CURRENT (mA)
100
3631 F01
Figure 1. RISET Selection
10
maximum average output current for this architecture is
limited to half of the peak current. Therefore, be sure to
select a value that sets the peak current with enough margin to provide adequate load current under all foreseeable
operating conditions.
Inductor Selection
The inductor, input voltage, output voltage and peak current determine the switching frequency of the LTC3631.
For a given input voltage, output voltage and peak current,
the inductor value sets the switching frequency when the
output is in regulation. A good first choice for the inductor
value can be determined by the following equation:
V
V
L = OUT • 1– OUT
VIN
f • IPEAK
The variation in switching frequency with input voltage
and inductance is shown in the following two figures for
typical values of VOUT. For lower values of IPEAK, multiply
the frequency in Figure 2 and Figure 3 by 225mA/IPEAK.
An additional constraint on the inductor value is the
LTC3631’s 100ns minimum on-time of the high side switch.
Therefore, in order to keep the current in the inductor well
controlled, the inductor value must be chosen so that it
is larger than LMIN, which can be computed as follows:
L MIN =
VIN(MAX) • tON(MIN)
IPEAK(MAX)
where VIN(MAX) is the maximum input supply voltage for
the application, tON(MIN) is 100ns, and IPEAK(MAX) is the
maximum allowed peak inductor current. Although the
above equation provides the minimum inductor value,
higher efficiency is generally achieved with a larger inductor
value, which produces a lower switching frequency. For a
given inductor type, however, as inductance is increased
DC resistance (DCR) also increases. Higher DCR translates into higher copper losses and lower current rating,
both of which place an upper limit on the inductance. The
recommended range of inductor values for small surface
mount inductors as a function of peak current is shown in
Figure 4. The values in this range are a good compromise
between the trade-offs discussed above. For applications
3631fe
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LTC3631
Applications Information
SWITCHING FREQUENCY (kHz)
700
VOUT = 5V
ISET OPEN
600
L = 22µH
where board area is not a limiting factor, inductors with
larger cores can be used, which extends the recommended
range of Figure 4 to larger values.
L = 47µH
Inductor Core Selection
500
400
300
L = 100µH
200
L = 220µH
100
0
5
10
15
20 25 30 35
INPUT VOLTAGE (V)
40
45
3631 F02
Figure 2. Switching Frequency for VOUT = 5V
500
SWITCHING FREQUENCY (kHz)
450
L = 22µH
400
VOUT = 3.3V
ISET OPEN
350
300
L = 47µH
250
200
150
L = 100µH
100
L = 220µH
50
0
5
10
15
20 25 30 35
INPUT VOLTAGE (V)
40
45
3631 F03
Figure 3. Switching Frequency for VOUT = 3.3V
INDUCTOR VALUE (µH)
10000
1000
Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means
that inductance collapses abruptly when the peak design
current is exceeded. This results in an abrupt increase in
inductor ripple current and consequently output voltage
ripple. Do not allow the core to saturate!
Different core materials and shapes will change the size/
current and price/current relationship of an inductor. Toroid
or shielded pot cores in ferrite or permalloy materials are
small and do not radiate energy but generally cost more
than powdered iron core inductors with similar characteristics. The choice of which style inductor to use mainly
depends on the price vs size requirements and any radiated
field/EMI requirements. New designs for surface mount
inductors are available from Coiltronics, Coilcraft, TDK,
Toko, Sumida and Vishay.
CIN and COUT Selection
100
10
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of the more expensive ferrite cores. Actual
core loss is independent of core size for a fixed inductor
value but is very dependent of the inductance selected.
As the inductance increases, core losses decrease. Unfortunately, increased inductance requires more turns of
wire and therefore copper losses will increase.
10
100
PEAK INDUCTOR CURRENT (mA)
1000
The input capacitor, CIN, is needed to filter the trapezoidal
current at the source of the top high side MOSFET. To
prevent large ripple voltage, a low ESR input capacitor
sized for the maximum RMS current should be used.
Approximate RMS current is given by:
IRMS = IOUT(MAX) •
3631 F04
Figure 4. Recommended Inductor Values for Maximum Efficiency
VOUT
VIN
•
−1
VIN
VOUT
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11
LTC3631
Applications Information
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations
do not offer much relief. Note that ripple current ratings
from capacitor manufacturers are often based only on
2000 hours of life which makes it advisable to further
derate the capacitor, or choose a capacitor rated at a
higher temperature than required. Several capacitors may
also be paralleled to meet size or height requirements in
the design.
The output capacitor, COUT, filters the inductor’s ripple
current and stores energy to satisfy the load current when
the LTC3631 is in sleep. The output ripple has a lower limit
of VOUT/160 due to the 5mV typical hysteresis of the feedback comparator. The time delay of the comparator adds
an additional ripple voltage that is a function of the load
current. During this delay time, the LTC3631 continues to
switch and supply current to the output. The output ripple
can be approximated by:
I
4 • 10 – 6 VOUT
ΔVOUT ≈ PEAK – ILOAD
+
2
C
160
OUT
The output ripple is a maximum at no load and approaches
lower limit of VOUT/160 at full load. Choose the output
capacitor COUT to limit the output voltage ripple at minimum load current.
The value of the output capacitor must be large enough
to accept the energy stored in the inductor without a large
change in output voltage. Setting this voltage step equal
to 1% of the output voltage, the output capacitor must be:
COUT
I
2
PEAK
> 50 • L •
VOUT
Typically, a capacitor that satisfies the voltage ripple requirement is adequate to filter the inductor ripple. To avoid
overheating, the output capacitor must also be sized to
handle the ripple current generated by the inductor. The
worst-case ripple current in the output capacitor is given
by IRMS = IPEAK/2. Multiple capacitors placed in parallel
may be needed to meet the ESR and RMS current handling
requirements.
Tantalum, special polymer, aluminum electrolytic, and
ceramic capacitors are all available in surface mount
packages. Special polymer capacitors offer very low ESR
but have lower capacitance density than other types.
Tantalum capacitors have the highest capacitance density
but it is important only to use types that have been surge
tested for use in switching power supplies. Aluminum
electrolytic capacitors have significantly higher ESR but
can be used in cost-sensitive applications provided that
consideration is given to ripple current ratings and longterm reliability. Ceramic capacitors have excellent low ESR
characteristics but can have high voltage coefficient and
audible piezoelectric effects. The high quality factor (Q)
of ceramic capacitors in series with trace inductance can
also lead to significant ringing.
Using Ceramic Input and Output Capacitors
Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. However, care must
be taken when these capacitors are used at the input and
output. When a ceramic capacitor is used at the input and
the power is supplied by a wall adapter through long wires,
a load step at the output can induce ringing at the input,
VIN. At best, this ringing can couple to the output and be
mistaken as loop instability. At worst, a sudden inrush
of current through the long wires can potentially cause a
voltage spike at VIN large enough to damage the LTC3631.
For applications with inductive source impedance, such
as a long wire, a series RC network may be required in
parallel with CIN to dampen the ringing of the input supply.
Figure 5 shows this circuit and the typical values required
to dampen the ringing.
LTC3631
LIN
VIN
R=
LIN
CIN
4 • CIN
3631 F05
CIN
Figure 5. Series RC to Reduce VIN Ringing
3631fe
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LTC3631
Applications Information
Output Voltage Programming
For the adjustable version, the output voltage is set by
an external resistive divider according to the following
equation:
controller is enabled. Figure 7 shows examples of configurations for driving the RUN pin from logic.
VSUPPLY
VIN
4.7M
LTC3631
R1
VOUT = 0.8V • 1+
R2
RUN
LTC3631
RUN
3631 F07
The resistive divider allows the VFB pin to sense a fraction
of the output voltage as shown in Figure 6. Output voltage
can range from 0.8V to VIN.
VOUT
R1
VFB
LTC3631
R2
GND
3631 F06
Figure 6. Setting the Output Voltage
To minimize the no-load supply current, resistor values in
the megohm range should be used; however, large resistor
values should be used with caution. The feedback divider
is the only load current when in shutdown. If PCB leakage current to the output node or switch node exceeds
the load current, the output voltage will be pulled up. In
normal operation, this is generally a minor concern since
the load current is much greater than the leakage. The
increase in supply current due to the feedback resistors
can be calculated from:
V
V
ΔIVIN = OUT • OUT
R1+ R2 VIN
Figure 7. RUN Pin Interface to Logic
The RUN pin can alternatively be configured as a precise
undervoltage lockout (UVLO) on the VIN supply with a
resistive divider from VIN to ground. The RUN pin comparator nominally provides 10% hysteresis when used in
this method; however, additional hysteresis may be added
with the use of the HYST pin. The HYST pin is an opendrain output that is pulled to ground whenever the RUN
comparator is not tripped. A simple resistive divider can
be used as shown in Figure 8 to meet specific VIN voltage
requirements.
VIN
R1
RUN
R2
LTC3631
HYST
R3
3631 F08
Figure 8. Adjustable Undervoltage Lockout
Specific values for these UVLO thresholds can be computed
from the following equations:
Run Pin with Programmable Hysteresis
The LTC3631 has a low power shutdown mode controlled
by the RUN pin. Pulling the RUN pin below 0.7V puts the
LTC3631 into a low quiescent current shutdown mode
(IQ ~ 3µA). When the RUN pin is greater than 1.2V, the
R1
Rising VIN UVLO Threshold = 1.21V • 1+
R2
R1
Falling VIN UVLO Threshold = 1.10V • 1+
R2 + R3
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13
LTC3631
Applications Information
The minimum value of these thresholds is limited to
the internal VIN UVLO thresholds that are shown in the
Electrical Characteristics table. The current that flows
through this divider will directly add to the shutdown,
sleep and active current of the LTC3631, and care should
be taken to minimize the impact of this current on the
overall efficiency of the application circuit. Resistor values
in the megohm range may be required to keep the impact
on quiescent shutdown and sleep currents low. Be aware
that the HYST pin cannot be allowed to exceed its absolute
maximum rating of 6V. To keep the voltage on the HYST
pin from exceeding 6V, the following relation should be
satisfied:
R3
VIN(MAX) •
< 6V
R1+ R2 + R3
The RUN pin may also be directly tied to the VIN supply
for applications that do not require the programmable undervoltage lockout feature. In this configuration, switching
is enabled when VIN surpasses the internal undervoltage
lockout threshold.
Soft-Start
The internal 0.75ms soft-start is implemented by ramping
both the effective reference voltage from 0V to 0.8V and
the peak current limit set by the ISET pin (50mA to 225mA).
To increase the duration of the reference voltage soft-start,
place a capacitor from the SS pin to ground. An internal
5µA pull-up current will charge this capacitor, resulting in
a soft-start ramp time given by:
tSS = CSS •
0.8V
5µA
from ISET to ground. A 1µA current is sourced out of the
ISET pin. With only a capacitor connected between ISET
and ground, the peak current ramps linearly from 50mA
to 225mA, and the peak current soft-start time can be
expressed as:
tSS(ISET) = CISET •
0.8V
1µA
A linear ramp of peak current appears as a quadratic
waveform on the output voltage. For the case where the
peak current is reduced by placing a resistor from ISET
to ground, the peak current offset ramps as a decaying
exponential with a time constant of RISET • CISET. For this
case, the peak current soft-start time is approximately
3 • RISET • CISET.
Unlike the SS pin, the ISET pin does not get pulled to
ground during an abnormal event; however, if the ISET
pin is floating (programmed to 225mA peak current),
the SS and ISET pins may be tied together and connected
to a capacitor to ground. For this special case, both the
peak current and the reference voltage will soft-start on
power‑up and after fault conditions. The ramp time for
this combination is CSS(ISET) • (0.8V/6µA).
Efficiency Considerations
The efficiency of a switching regulator is equal to the output
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
When the LTC3631 detects a fault condition (input supply
undervoltage or overvoltage) or when the RUN pin falls
below 1.1V, the SS pin is quickly pulled to ground and the
internal soft-start timer is reset. This ensures an orderly
restart when using an external soft-start capacitor.
The duration of the 0.75ms internal peak current soft-start
may be increased by placing a capacitor from the ISET pin
to ground. The peak current soft-start will ramp from 50mA
to the final peak current value determined by a resistor
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of
the losses: VIN operating current and I2R losses. The VIN
operating current dominates the efficiency loss at very
low load currents whereas the I2R loss dominates the
efficiency loss at medium to high load currents.
1. The VIN operating current comprises two components:
The DC supply current as given in the electrical characteristics and the internal MOSFET gate charge currents.
3631fe
14
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LTC3631
Applications Information
The gate charge current results from switching the gate
capacitance of the internal power MOSFET switches.
Each time the gate is switched from high to low to
high again, a packet of charge, dQ, moves from VIN to
ground. The resulting dQ/dt is the current out of VIN
that is typically larger than the DC bias current.
2. I2R losses are calculated from the resistances of the
internal switches, RSW, and external inductor RL. When
switching, the average output current flowing through
the inductor is “chopped” between the high side PMOS
switch and the low side NMOS switch. Thus, the series
resistance looking back into the switch pin is a function
of the top and bottom switch RDS(ON) values and the
duty cycle (DC = VOUT/VIN) as follows:
RSW = (RDS(ON)TOP)DC + (RDS(ON)BOT)(1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Characteristics curves. Thus, to obtain the I2R losses, simply add
RSW to RL and multiply the result by the square of the
average output current:
First, calculate the inductor value that gives the required
switching frequency:
3.3V
Ê
ˆ Ê 3.3V ˆ
L=Á
• 1–
˜ @ 47µH
Ë 250kHz • 225mA ˜¯ ÁË
24V ¯
Next, verify that this value meets the LMIN requirement.
For this input voltage and peak current, the minimum
inductor value is:
L MIN =
24V • 100ns
≅ 10µH
225mA
Therefore, the minimum inductor requirement is satisfied,
and the 47μH inductor value may be used.
Next, CIN and COUT are selected. For this design, CIN should
be sized for a current rating of at least:
IRMS = 100mA •
3.3V
24V
•
– 1 ≅ 35mA RMS
24V
3.3V
I2R Loss = IO2(RSW + RL)
Due to the low peak current of the LTC3631, decoupling
the VIN supply with a 1µF capacitor is adequate for most
applications.
Other losses, including CIN and COUT ESR dissipative
losses and inductor core losses, generally account for
less than 2% of the total power loss.
COUT will be selected based on the output voltage ripple
requirement. For a 2% (67mV) output voltage ripple at no
load, COUT can be calculated from:
C OUT =
Thermal Considerations
The LTC3631 does not dissipate much heat due to its high
efficiency and low peak current level. Even in worst-case
conditions (high ambient temperature, maximum peak
current and high duty cycle), the junction temperature
will exceed ambient temperature by only a few degrees.
Design Example
As a design example, consider using the LTC3631 in an
application with the following specifications: VIN = 24V,
VOUT = 3.3V, IOUT = 100mA, f = 250kHz. Furthermore, assume for this example that switching should start when VIN
is greater than 12V and should stop when VIN is less than 8V.
225mA • 4 • 10 –6
3.3V
2 67mV –
160
A 9.7µF capacitor gives this typical output voltage ripple
at no load. Choose a 10µF capacitor as a standard value.
The output voltage can now be programmed by choosing
the values of R1 and R2. Choose R2 = 240k and calculate
R1 as:
V
R1= OUT – 1 • R2 = 750k
0.8V
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15
LTC3631
Applications Information
The undervoltage lockout requirement on VIN can be
satisfied with a resistive divider from VIN to the RUN
and HYST pins. Choose R1 = 2M and calculate R2 and
R3 as follows:
R2 =
R3 =
1.21V
• R1= 224k
VIN(RISING) – 1.21V
1.1V
• R1– R2 = 94.8k
VIN(FALLING) – 1.1V
Choose standard values for R2 = 226k and R3 = 95.3k.
The ISET pin should be left open in this example to select
maximum peak current (225mA). Figure 9 shows a complete schematic for this design example.
VIN
1µF
2M
4. Flood all unused area on all layers with copper. Flooding
with copper will reduce the temperature rise of power
components. You can connect the copper areas to any
DC net (VIN, VOUT, GND or any other DC rail in your
system).
Example Layout
L1
VIN
VIN
VOUT
3.3V
100mA
10µF
SW
LTC3631
RUN
3. Keep the switching node, SW, away from all sensitive
small signal nodes. The rapid transitions on the switching
node can couple to high impedance nodes, in particular
VFB, and create increased output ripple.
VOUT
SW
LTC3631
47µH
VIN
24V
2. Connect the (+) terminal of the input capacitor, CIN, as
close as possible to the VIN pin. This capacitor provides
the AC current into the internal power MOSFETs.
ISET
226k
SS
750k
95.3k
VFB
HYST
GND
240k
R1
RUN
CIN
CSS
HYST
VFB
SS
ISET
COUT
GND
RSET
R2
3631 F09a
3631 F09
Figure 9. 24V to 3.3V, 100mA Regulator at 250kHz
L1
VIN
CIN
VOUT
COUT
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3631. Check the following in your layout:
1. Large switched currents flow in the power switches
and input capacitor. The loop formed by these components should be as small as possible. A ground plane
is recommended to minimize ground impedance.
R1
R2
RSET
CSS
GND
VIAS TO GROUND PLANE
VIAS TO INPUT SUPPLY (VIN)
OUTLINE OF LOCAL GROUND PLANE
3631fe
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LTC3631
TYPICAL APPLICATIONS
L1
100µH
VIN
5V TO 45V
CIN
4.7µF
VIN
SW
LTC3631
RUN
HYST
ISET
R1
1.47M
VFB
SS
GND
VOUT
5V
COUT
100µF
R2
280k
CSS
47nF
3631 F10a
CIN: TDK C5750X7R2A475MT
COUT: AVX 1812D107MAT
L1: TDK SLF7045T-101MR50-PF
Figure 10. High Efficiency 5V Regulator
3.3V, 100mA Regulator with Peak Current Soft-Start, Small Size
VIN
4.5V TO 24V
L1
22µH
CIN
1µF
VIN
SW
RUN
ISET
SS
CSS
22nF
R1
294k
LTC3631
VFB
VOUT
3.3V
COUT 100mA
10µF
R2
93.1k
HYST
GND
Soft-Start Waveforms
OUTPUT
VOLTAGE
1V/DIV
3642 TA03a
INDUCTOR
CURRENT
50mA/DIV
CIN: TDK C3216X7R1E105KT
COUT: AVX 08056D106KAT2A
L1: MURATA LQH43CN220K03
500µs/DIV
Positive-to-Negative Converter
Maximum Load Current vs Input Voltage
L1
100µH
CIN
1µF
VIN
100
SW
LTC3631
RUN
ISET
R1
1M
VFB
HYST
SS
GND
R2
71.5k
CIN: TDK C3225X7R1H105KT
COUT: MURATA GRM32DR71C106KA01
L1: TYCO/COEV DQ6545-101M
COUT
10µF
VOUT
–12V
3631 TA04a
MAXIMUM LOAD CURRENT (mA)
VIN
4.5V TO 33V
3631 TA03b
ISET OPEN
90
VOUT = –3V
VOUT = –5V
80
70
VOUT = –12V
60
50
40
30
20
5
10
15
20 25 30 35
INPUT VOLTAGE (V)
40
45
3631 TA04b
3631fe
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17
LTC3631
TYPICAL APPLICATIONS
Small Size, Limited Peak Current, 20mA Regulator
VIN
7V TO 45V
L1
470µH
CIN
1µF
R3
470k
R4
100k
R5
33k
VIN
SW
R1
470k
LTC3631
RUN
VFB
COUT
10µF
VOUT
5V
20mA
R2
88.7k
HYST
SS
ISET GND
3631 TA05a
CIN: TDK C3225X7R1H105KT
COUT: AVX 08056D106KAT2A
L1: MURATA LQH43CN471K03
High Efficiency 15V, 20mA Regulator
100
L1
1000µH
CIN
1µF
VIN
SW
LTC3631
RUN
ISET
SS
VFB
HYST
GND
CIN: AVX 18125C105KAT2A
COUT: TDK C3216X7R1E475KT
L1: TDK SLF7045T-102MR14
R1
3M
COUT
4.7µF
R2
169k
3631 TA07a
VOUT
15V
20mA
EFFICIENCY (%)
VIN
15V TO 45V
Efficiency vs Load Current
95
VIN = 24V
90
VIN = 36V
85
80
VIN = 45V
75
70
65
60
0.1
1
10
LOAD CURRENT (mA)
100
3631 TA07b
3631fe
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LTC3631
Package Description
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
DD Package
8-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1698 Rev C)
0.70 ±0.05
3.5 ±0.05
1.65 ±0.05
2.10 ±0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50
BSC
2.38 ±0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
PIN 1
TOP MARK
(NOTE 6)
0.200 REF
3.00 ±0.10
(4 SIDES)
R = 0.125
TYP
5
0.40 ± 0.10
8
1.65 ± 0.10
(2 SIDES)
0.75 ±0.05
4
0.25 ± 0.05
1
(DD8) DFN 0509 REV C
0.50 BSC
2.38 ±0.10
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON TOP AND BOTTOM OF PACKAGE
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19
LTC3631
Package Description
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MS8E Package
8-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1662 Rev K)
BOTTOM VIEW OF
EXPOSED PAD OPTION
1.88
(.074)
1
1.88 ±0.102
(.074 ±.004)
0.29
REF
1.68
(.066)
0.889 ±0.127
(.035 ±.005)
0.05 REF
5.10
(.201)
MIN
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
1.68 ±0.102 3.20 – 3.45
(.066 ±.004) (.126 – .136)
8
3.00 ±0.102
(.118 ±.004)
(NOTE 3)
0.65
(.0256)
BSC
0.42 ±0.038
(.0165 ±.0015)
TYP
8
7 6 5
0.52
(.0205)
REF
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
3.00 ±0.102
(.118 ±.004)
(NOTE 4)
4.90 ±0.152
(.193 ±.006)
DETAIL “A”
0° – 6° TYP
GAUGE PLANE
0.53 ±0.152
(.021 ±.006)
DETAIL “A”
1
2 3
4
1.10
(.043)
MAX
0.86
(.034)
REF
0.18
(.007)
SEATING
PLANE
0.22 – 0.38
(.009 – .015)
TYP
0.65
(.0256)
NOTE:
BSC
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD
SHALL NOT EXCEED 0.254mm (.010") PER SIDE.
0.1016 ±0.0508
(.004 ±.002)
MSOP (MS8E) 0213 REV K
3631fe
20
For more information www.linear.com/LTC3631
LTC3631
Revision History
(Revision history begins at Rev B)
REV
DATE
DESCRIPTION
B
05/10
Updated Absolute Maximum Ratings and Order Information Sections
PAGE NUMBER
Updated Note 2
3
Updated Graphs G09, G18 and G19
C
D
E
10/10
08/11
05/13
2
4, 5, 6
Updated GND pin text in Pin Functions
6
Text added to “Output Voltage Programming” section
12
“Example Layout” Art added
16
Updated Typical Application
22
Updated Related Parts
22
Updated CIN and COUT Selection section
12
Updated Efficiency Considerations section
15
Updated equation in Design Example section
16
Updated Figure 9
16
Clarified inductor values in Figures 2 and 3
11
3631fe
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/LTC3631
21
LTC3631
Typical Application
5V, 100mA Regulator for Automotive Applications
VBATT
4.5V TO 45V
AND SURVIVES
TRANSIENTS
UP TO 60V
L1
100µH
CIN
2.2µF
VIN
SW
LTC3631
RUN
ISET
SS
VFB
HYST
GND
CIN: TDK C3225X7R2A225M
COUT: KEMET C1210C106K4RAC
L1: COILTRONICS DRA73-101-R
R1
470k
COUT
10µF
VOUT*
5V
100mA
R2
88.7k
3631 TA06a
*VOUT = VBATT FOR VBATT < 5V
Related Parts
PART NUMBER
LTC3632
DESCRIPTION
50V, 20mA Synchronous Micropower Step-Down DC/DC Converter
COMMENTS
VIN: 4.5V to 50V (60VMAX), VOUT(MIN) = 0.8V, IQ = 12µA,
ISD = 3µA, 3mm × 3mm DFN, MS8E
LTC3642
45V, 50mA Synchronous Micropower Step-Down DC/DC Converter VIN: 4.5V to 45V (60VMAX), VOUT(MIN) = 0.8V, IQ = 12µA,
ISD = 3µA, 3mm × 3mm DFN8, MS8E
VIN: 3V to 18V, VOUT(MIN) = 1.2V, IQ = 10µA, ISD = 6µA, MS8E
LTC1474
18V, 250mA (IOUT), High Efficiency Step-Down DC/DC Converter
VIN: 3.2V to 34V, VOUT(MIN) = 1.25V, IQ = 12µA, ISD < 1µA,
LT1934/LT1934-1 36V, 250mA (IOUT), Micropower Step-Down DC/DC Converter with
Burst Mode Operation
ThinSOT™ Package
LT1939
25V, 2A, 2.5MHz High Efficiency DC/DC Converter and LDO
VIN: 3.6V to 25V, VOUT(MIN) = 0.8V, IQ = 2.5mA, ISD < 10µA,
3mm × 3mm DFN10
Controller
LT1976/LT1977 60V, 1.2A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC VIN: 3.3V to 60V, VOUT(MIN) = 1.2V, IQ = 100µA, ISD < 1µA,
Converter with Burst Mode Operation
TSSOP16E
VIN: 3.3V to 60V, VOUT(MIN) = 1.25V, IQ = 100µA, ISD < 1µA,
LT3437
60V, 400mA (IOUT), Micropower Step-Down DC/DC Converter with
Burst Mode Operation
3mm × 3mm DFN10, TSSOP16E
VIN: 4V to 40V, VOUT(MIN) = 1.2V, IQ = 26µA, ISD < 1µA,
LT3470
40V, 250mA (IOUT), High Efficiency Step-Down DC/DC Converter
with Burst Mode Operation
2mm × 3mm DFN8, ThinSOT
VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70µA, ISD < 1µA,
LT3685
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter
3mm × 3mm DFN10, MSOP10E
3631fe
22 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/LTC3631
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/LTC3631
LT 0513 REV E • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 2009
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