LTC3632
High Efficiency, High Voltage
20mA Synchronous
Step-Down Converter
FEATURES
DESCRIPTION
Wide Input Voltage Range: Operation from
4.5V to 50V
n Overvoltage Lockout Provides Protection Up to 60V
n Internal High Side and Low Side Power Switches
n No Compensation Required
n 20mA Output Current
n Low Dropout Operation: 100% Duty Cycle
n Low Quiescent Current: 12µA
n Wide Output Voltage Range: 0.8V to V
IN
n 0.8V ±1% Feedback Voltage Reference
n Adjustable Peak Current Limit
n Internal and External Soft-Start
n Precise RUN Pin Threshold with Adjustable
Hysteresis
n Few External Components Required
n Low Profile (0.75mm) 3mm × 3mm DFN and
Thermally Enhanced MS8E Packages
The LTC®3632 is a high efficiency, high voltage step-down
DC/DC converter with internal high side and synchronous
power switches that draws only 12μA typical DC supply current at no load while maintaining output voltage
regulation.
n
APPLICATIONS
n
n
n
n
n
n
4mA to 20mA Current Loops
Industrial Control Supplies
Distributed Power Systems
Portable Instruments
Battery-Operated Devices
Automotive Power Systems
The LTC3632 can supply up to 20mA load current and
features a programmable peak current limit that provides
a simple method for optimizing efficiency in lower current
applications. The LTC3632’s combination of Burst Mode®
operation, integrated power switches, low quiescent current, and programmable peak current limit provides high
efficiency over a broad range of load currents.
With its wide 4.5V to 50V input range and internal
overvoltage monitor capable of protecting the part through
60V surges, the LTC3632 is a robust converter suited for
regulating a wide variety of power sources. Additionally,
the LTC3632 includes a precise run threshold and soft‑start
feature to guarantee that the power system start-up is
well-controlled in any environment.
The LTC3632 is available in the thermally enhanced
3mm × 3mm DFN and MS8E packages.
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
TYPICAL APPLICATION
Efficiency and Power Loss vs Load Current
100
5V, 20mA Step-Down Converter
SW
LTC3632
RUN
VFB
HYST
ISET
SS
GND
1.47M
280k
3632 TA01a
VOUT
5V
10µF 20mA
80
EFFICIENCY (%)
1mH
VIN
EFFICIENCY
100
70
60
50
10
POWER LOSS
POWER LOSS (mW)
VIN
5V TO 50V
1µF
1000
90
40
30
20
0.1
VIN = 10V 1
VIN = 48V
1
LOAD CURRENT (mA)
10
3632 TA01b
3632fc
1
LTC3632
ABSOLUTE MAXIMUM RATINGS (Note 1)
VIN Supply Voltage...................................... –0.3V to 60V
SW Voltage (DC).............................–0.3V to (VIN + 0.3V)
RUN Voltage............................................... –0.3V to 60V
VFB, HYST, ISET, SS Voltages......................... –0.3V to 6V
Operating Junction Temperature Range
(Note 2)................................................... –40°C to 125°C
Storage Temperature Range.................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
MS8E................................................................. 300°C
PIN CONFIGURATION
TOP VIEW
TOP VIEW
SW
VIN
ISET
SS
1
2
3
4
9
GND
8
7
6
5
SW 1
GND
HYST
VFB
RUN
VIN 2
ISET 3
9
GND
SS 4
MS8E PACKAGE
8-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 40°C/W, θJC = 5°-10°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
8
GND
7
HYST
6
VFB
5
RUN
DD PACKAGE
8-LEAD (3mm × 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 43°C/W, θJC = 3°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3632EMS8E#PBF
LTC3632EMS8E#TRPBF
LTFFZ
8-Lead Plastic MSOP
–40°C to 125°C
LTC3632IMS8E#PBF
LTC3632IMS8E#TRPBF
LTFFZ
8-Lead Plastic MSOP
–40°C to 125°C
LTC3632EDD#PBF
LTC3632EDD#TRPBF
LFGB
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3632IDD#PBF
LTC3632IDD#TRPBF
LFGB
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3632fc
2
LTC3632
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are for TA = 25°C (Note 2). VIN = 10V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input Supply (VIN)
VIN
Input Voltage Operating Range
4.5
UVLO
VIN Undervoltage Lockout
VIN Rising
VIN Falling
Hysteresis
OVLO
VIN Overvoltage Lockout
VIN Rising
VIN Falling
Hysteresis
IQ
DC Supply Current (Note 3)
Active Mode
Sleep Mode
Shutdown Mode
l
l
50
V
3.80
3.75
4.15
4.00
150
4.50
4.35
54
52
56
54
2
59
57
V
V
V
125
12
3
220
22
6
µA
µA
µA
V
VRUN = 0V
V
V
mV
Output Supply (VFB)
VFB
Feedback Comparator Trip Voltage
VFB Rising
l
0.792
0.800
0.808
VHYST
Feedback Comparator Hysteresis Voltage
l
3
5
7
IFB
Feedback Pin Current
VFB = 1V
∆VLINEREG
Feedback Voltage Line Regulation
VIN = 4.5V to 50V
0
10
VRUN
RUN Pin Threshold Voltage
RUN Rising
RUN Falling
Hysteresis
1.17
1.06
1.21
1.10
110
1.25
1.14
V
V
mV
IRUN
RUN Pin Leakage Current
RUN = 1.3V
–10
0
10
nA
–10
0.001
mV
nA
%/V
Operation
VHYSTL
Hysteresis Pin Voltage Low
RUN < 1V, IHYST = 1mA
0.07
0.1
V
IHYST
Hysteresis Pin Leakage Current
VHYST = 1.3V
–10
0
10
nA
ISS
Soft-Start Pin Pull-Up Current
VSS < 1.5V
4.5
5.5
6.5
µA
tINTSS
Internal Soft-Start Time
SS Pin Floating
IPEAK
Peak Current Trip Threshold
ISET Floating
500k Resistor from ISET to GND
ISET Shorted to GND
60
mA
mA
mA
RON
Power Switch On-Resistance
Top Switch
Bottom Switch
ISW = –10mA
ISW = 10mA
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3632 is tested under pulsed load conditions such that TJ ≈ TA.
LTC3632E is guaranteed to meet specifications from 0°C to 85°C junction
temperature. Specifications over the –40°C to 125°C operating junction
temperature range are assured by design, characterization and correlation
with statistical process controls. The LTC3632I is guaranteed over the
full –40°C to 125°C operating junction temperature range. Note that the
0.75
l
40
8
50
25
10
5.0
2.5
ms
13
Ω
Ω
maximum ambient temperature consistent with these specifications is
determined by specific operating conditions in conjunction with board
layout, the rated package thermal impedance and other environmental
factors. The junction temperature (TJ, in °C) is calculated from the ambient
temperature (TA, in °C) and power dissipation (PD, in Watts) according to
the formula:
TJ = TA + (PD • θJA), where θJA (in °C/W) is the package thermal
impedance.
Note 3: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
3632fc
3
LTC3632
TYPICAL PERFORMANCE CHARACTERISTICS
100
95
VIN = 24V
90
100
95
VIN = 12V
VIN = 36V
70
65
90
80
75
VIN = 48V
70
VIN = 36V
65
60
60
VOUT = 5V
FIGURE 11 CIRCUIT
55
50
0.1
1
LOAD CURRENT (mA)
VOUT = 3.3V
FIGURE 11 CIRCUIT
50
0.1
1
LOAD CURRENT (mA)
3632 G01
ILOAD = 5mA
75
70
10
15
20
25 30 35 40
INPUT VOLTAGE (V)
45
50
0.10
0
–0.10
–0.20
0.798
–40
–10
20
50
80
TEMPERATURE (°C)
5.02
5.00
4.99
4.98
4.97
–0.40
4.96
–0.50
10
5
15
20 25 30 35 40
INPUT VOLTAGE (V)
45
4.95
50
3632 G06
110
3632 G07
60
PEAK CURRENT TRIP THRESHOLD (mA)
VIN = 10V
5.4
5.2
5.0
4.8
4.6
4.4
–40
–10
20
50
80
TEMPERATURE (°C)
110
3632 G08
20
10
5
15
LOAD CURRENT (mA)
0
3632 G05
5.6
10
5.01
–0.30
FEEDBACK COMPARATOR HYSTERESIS (mV)
FEEDBACK COMPARATOR TRIP VOLTAGE (V)
0.799
1
LOAD CURRENT (mA)
VIN = 10V
VOUT = 5V
FIGURE 11 CIRCUIT
5.04
Feedback Comparator Hysteresis
Voltage vs Temperature
0.800
VOUT = 2.5V
FIGURE 11 CIRCUIT
Load Regulation
5.05
0.20
Feedback Comparator Trip
Voltage vs Temperature
VIN = 10V
VIN = 36V
5.03
3632 G04
0.801
VIN = 48V
65
3632 G03
OUTPUT VOLTAGE (V)
ΔVOUT/VOUT (%)
EFFICIENCY (%)
ILOAD = 20mA
ILOAD = 1mA
70
50
0.1
10
ILOAD = 20mA
FIGURE 11 CIRCUIT
0.40
90
80
75
55
0.30
85
80
Line Regulation
0.50
VOUT = 5V
FIGURE 11 CIRCUIT
95
VIN = 24V
3632 G02
Efficiency vs Input Voltage
100
VIN = 12V
85
60
55
10
Efficiency vs Load Current,
VOUT = 2.5V
95
VIN = 24V
85
EFFICIENCY (%)
EFFICIENCY (%)
VIN = 48V
75
100
VIN = 12V
90
85
80
Efficiency vs Load Current,
VOUT = 3.3V
EFFICIENCY (%)
Efficiency vs Load Current,
VOUT = 5V
TA = 25°C, unless otherwise noted.
Peak Current Trip Threshold
vs Temperature and ISET
VIN = 10V
50
ISET OPEN
40
30
RISET = 500k
20
ISET = GND
10
0
–40
–10
20
50
80
TEMPERATURE (°C)
110
3632 G09
3632fc
4
LTC3632
TYPICAL PERFORMANCE CHARACTERISTICS
40
30
20
10
0
200
400
600
800
RISET (kΩ)
1000
50
14
ISET OPEN
12
40
30
RSET = 500k
20
ISET = GND
10
0
1200
VIN = 10V
VIN SUPPLY CURRENT (µA)
60
VIN = 10V
50
0
0
5
SLEEP
8
6
SHUTDOWN
20
50
80
TEMPERATURE (°C)
8
7
7
6
TOP
5
4
3
1
110
BOTTOM
RUN COMPARATOR THRESHOLD (V)
SWITCH LEAKAGE CURRENT (µA)
4
BOTTOM
3
2
1
0
10
20
30
50
40
0
–40
–10
20
50
80
TEMPERATURE (°C)
110
3632 G15
Operating Waveforms
1.30
0.20
0.15
SW = 50V
SW = 0V
–10
TOP
5
3632 G14
VIN = 50V
0
–40
6
RUN Comparator Threshold
Voltage vs Temperature
0.25
0.05
45
VIN = 10V
INPUT VOLTAGE (V)
Switch Leakage Current
vs Temperature
0.10
25
35
INPUT VOLTAGE (V)
3632 G12
8
3632 G13
0.30
15
5
Switch On-Resistance
vs Temperature
2
2
–10
SHUTDOWN
4
0
10 15 20 25 30 35 40 45 50
INPUT VOLTAGE (V)
SWITCH ON-RESISTANCE (Ω)
SWITCH ON-RESISTANCE (Ω)
VIN SUPPLY CURRENT (µA)
12
0
–40
6
Switch On-Resistance
vs Input Voltage
VIN = 10V
4
8
3632 G11
Quiescent Supply Current
vs Temperature
10
SLEEP
10
2
3632 G10
14
Quiescent Supply Current
vs Input Voltage
Peak Current Trip Threshold
vs Input Voltage
PEAK CURRENT TRIP THRESHOLD (mA)
PEAK CURRENT TRIP THRESHOLD (mA)
60
Peak Current Trip Threshold
vs RISET
TA = 25°C, unless otherwise noted.
20
50
80
TEMPERATURE (°C)
110
3632 G16
1.25
SWITCH
VOLTAGE
20V/DIV
RISING
OUTPUT
VOLTAGE
100mV/DIV
INDUCTOR
CURRENT
50mA/DIV
1.20
1.15
FALLING
1.10
VIN = 48V
VOUT = 5V
ILOAD = 10mA
1.05
1.00
–40
–10
20
50
80
TEMPERATURE (°C)
20µs/DIV
3632 G18
110
3632 G17
3632fc
5
LTC3632
TYPICAL PERFORMANCE CHARACTERISTICS
Soft-Start Waveforms
Load Step Transient Response
OUTPUT
VOLTAGE
2V/DIV
OUTPUT
VOLTAGE
100mV/DIV
OUTPUT
VOLTAGE
1V/DIV
LOAD
CURRENT
20mA/DIV
CSS = 0.047µF
5ms/DIV
3632 G19
Short-Circuit Response
INDUCTOR
CURRENT
20mA/DIV
VIN = 10V
VOUT = 5V
1ms/DIV
3632 G20
VIN = 10V
VOUT = 5V
200µs/DIV
3632 G21
PIN FUNCTIONS
SW (Pin 1): Switch Node Connection to Inductor. This
pin connects to the drains of the internal power MOSFET
switches.
VIN (Pin 2): Main Supply Pin. A ceramic bypass capacitor
should be tied between this pin and GND (Pin 8).
ISET (Pin 3): Peak Current Set Input. A resistor from this
pin to ground sets the peak current trip threshold. Leave
floating for the maximum peak current (50mA). Short this
pin to ground for the minimum peak current (10mA). A
1µA current is sourced out of this pin.
SS (Pin 4): Soft-Start Control Input. A capacitor to ground
at this pin sets the ramp time to full current output during start-up. A 5.5µA current is sourced out of this pin. If
left floating, the ramp time defaults to an internal 0.75ms
soft-start.
RUN (Pin 5): Run Control Input. A voltage on this pin
above 1.2V enables normal operation. Forcing this pin
below 0.7V shuts down the LTC3632, reducing quiescent
current to approximately 3µA.
VFB (Pin 6): Output Voltage Feedback. Connect to an
external resistive divider to divide the output voltage down
for comparison to the 0.8V reference.
HYST (Pin 7): Run Hysteresis Open-Drain Logic Output.
This pin is pulled to ground when RUN (Pin 5) is below
1.2V. This pin can be used to adjust the RUN pin hysteresis.
See Applications Information.
GND (Pin 8, Exposed Pad Pin 9): Ground. The exposed
pad must be soldered to the printed circuit board ground
plane for optimal electrical and thermal performance.
3632fc
6
LTC3632
BLOCK DIAGRAM
VIN
1µA
ISET
2
C2
3
PEAK CURRENT
COMPARATOR
–
+
RUN
+
5
1.2V
LOGIC
AND
SHOOTTHROUGH
PREVENTION
–
SW
L1
VOUT
1
C1
HYST
7
+
REVERSE-CURRENT
COMPARATOR
FEEDBACK
COMPARATOR
GND
GND
8
9
–
VOLTAGE
REFERENCE
+
+
–
0.800V
5.5µA
SS
4
VFB
R1
6
3632 BD
R2
3632fc
7
LTC3632
OPERATION (Refer to Block Diagram)
The LTC3632 is a step-down DC/DC converter with internal
power switches that uses Burst Mode control, combining low quiescent current with high switching frequency,
which results in high efficiency across a wide range of
load currents. Burst Mode operation functions by using
short “burst” cycles to ramp the inductor current through
the internal power switches, followed by a sleep cycle
where the power switches are off and the load current is
supplied by the output capacitor. During the sleep cycle,
the LTC3632 draws only 12µA of supply current. At light
loads, the burst cycles are a small percentage of the total
cycle time which minimizes the average supply current,
greatly improving efficiency.
Main Control Loop
The feedback comparator monitors the voltage on the VFB
pin and compares it to an internal 800mV reference. If
this voltage is greater than the reference, the comparator
activates a sleep mode in which the power switches and
current comparators are disabled, reducing the VIN pin
supply current to only 12µA. As the load current discharges
the output capacitor, the voltage on the VFB pin decreases.
When this voltage falls 5mV below the 800mV reference,
the feedback comparator trips and enables burst cycles.
At the beginning of the burst cycle, the internal high side
power switch (P-channel MOSFET) is turned on and the
inductor current begins to ramp up. The inductor current
increases until either the current exceeds the peak current comparator threshold or the voltage on the VFB pin
exceeds 800mV, at which time the high side power switch
is turned off and the low side power switch (N-channel
MOSFET) turns on. The inductor current ramps down until
the reverse-current comparator trips, signaling that the
current is close to zero. If the voltage on the VFB pin is
still less than the 800mV reference, the high side power
switch is turned on again and another cycle commences.
The average current during a burst cycle will normally be
greater than the average load current. For this architecture,
the maximum average output current is equal to half of
the peak current.
The hysteretic nature of this control architecture results
in a switching frequency that is a function of the input
voltage, output voltage and inductor value. This behavior
provides inherent short‑circuit protection. If the output
is shorted to ground, the inductor current will decay very
slowly during a single switching cycle. Since the high side
switch turns on only when the inductor current is near
zero, the LTC3632 inherently switches at a lower frequency
during start-up or short-circuit conditions.
Start-Up and Shutdown
If the voltage on the RUN pin is less than 0.7V, the LTC3632
enters a shutdown mode in which all internal circuitry is
disabled, reducing the DC supply current to 3µA. When the
voltage on the RUN pin exceeds 1.2V, normal operation of
the main control loop is enabled. The RUN pin comparator
has 110mV of internal hysteresis, and therefore must fall
below 1.1V to disable the main control loop.
The HYST pin provides an added degree of flexibility for
the RUN pin operation. This open-drain output is pulled
to ground whenever the RUN comparator is not tripped,
signaling that the LTC3632 is not in normal operation. In
applications where the RUN pin is used to monitor the
VIN voltage through an external resistive divider, the HYST
pin can be used to increase the effective RUN comparator
hysteresis.
An internal 0.75ms soft-start function limits the ramp rate
of the output voltage on start-up to prevent excessive input
supply droop. If a longer ramp time and consequently less
supply droop is desired, a capacitor can be placed from the
SS pin to ground. The 5µA current that is sourced out of this
pin will create a smooth voltage ramp on the capacitor. If
this ramp rate is slower than the internal 0.75ms soft-start,
then the output voltage will be limited by the ramp rate
3632fc
8
LTC3632
OPERATION (Refer to Block Diagram)
on the SS pin instead. The internal and external soft-start
functions are reset on start-up and after an undervoltage
or overvoltage event on the input supply.
through the resistor generates a voltage that is translated
into an offset in the peak current comparator, which limits
the peak inductor current.
In order to ensure a smooth start-up transition in any
application, the internal soft‑start also ramps the peak
inductor current from 10mA during its 0.75ms ramp time
to the set peak current threshold. The external ramp on
the SS pin does not limit the peak inductor current during
start-up; however, placing a capacitor from the ISET pin to
ground does provide this capability.
Input Undervoltage and Overvoltage Lockout
Peak Inductor Current Programming
The offset of the peak current comparator nominally provides a peak inductor current of 50mA. This peak inductor
current can be adjusted by placing a resistor from the ISET
pin to ground. The 1µA current sourced out of this pin
The LTC3632 implements a protection feature which disables switching when the input voltage is not within the
4.5V to 50V operating range. If VIN falls below 4V typical
(4.35V maximum), an undervoltage detector disables
switching. Similarly, if VIN rises above 55V typical (53V
minimum), an overvoltage detector disables switching.
When switching is disabled, the LTC3632 can safely sustain
input voltages up to the absolute maximum rating of 60V.
Switching is enabled when the input voltage returns to the
4.5V to 50V operating range.
3632fc
9
LTC3632
APPLICATIONS INFORMATION
The basic LTC3632 application circuit is shown on the front
page of this data sheet. External component selection is
determined by the maximum load current requirement and
begins with the selection of the peak current programming
resistor, RISET. The inductor value L can then be determined,
followed by capacitors CIN and COUT.
Peak Current Resistor Selection
The peak current comparator has a maximum current limit
of 50mA nominally, which results in a maximum average current of 25mA. For applications that demand less
current, the peak current threshold can be reduced to as
little as 10mA. This lower peak current allows the use of
lower value, smaller components (input capacitor, output
capacitor and inductor), resulting in lower input supply
ripple and a smaller overall DC/DC converter.
The threshold can be easily programmed with an appropriately chosen resistor (RISET) between the ISET pin
and ground. The value of resistor for a particular peak
current can be computed by using Figure 1 or the following equation:
RISET = IPEAK • 21 • 106
where 10mA < IPEAK < 50mA.
The peak current is internally limited to be within the
range of 10mA to 50mA. Shorting the ISET pin to ground
programs the current limit to 10mA, and leaving it floating
sets the current limit to the maximum value of 50mA. When
selecting this resistor value, be aware that the maximum
1100
1000
900
800
RISET (k)
700
600
500
400
300
200
100
0
4
6
10 12 14 16 18
8
MAXIMUM LOAD CURRENT (mA)
20
3632 F01
Figure 1. RISET Selection
10
average output current for this architecture is limited
to half of the peak current. Therefore, be sure to select
a value that sets the peak current with enough margin
to provide adequate load current under all foreseeable
operating conditions.
Inductor Selection
The inductor, input voltage, output voltage and peak current determine the switching frequency of the LTC3632.
For a given input voltage, output voltage and peak current,
the inductor value sets the switching frequency when the
output is in regulation. A good first choice for the inductor
value can be determined by the following equation:
V
V
L = OUT • 1– OUT
VIN
f • IPEAK
The variation in switching frequency with input voltage
and inductance is shown in the following two figures for
typical values of VOUT. For lower values of IPEAK, multiply
the frequency in Figure 2 and Figure 3 by 50mA/IPEAK.
An additional constraint on the inductor value is the
LTC3632’s 100ns minimum on-time of the high side switch.
Therefore, in order to keep the current in the inductor well
controlled, the inductor value must be chosen so that it is
larger than LMIN, which can be computed as follows:
L MIN =
VIN(MAX) • tON(MIN)
IPEAK(MAX)
where VIN(MAX) is the maximum input supply voltage for
the application, tON(MIN) is 100ns, and IPEAK(MAX) is the
maximum allowed peak inductor current. Although the
above equation provides the minimum inductor value,
higher efficiency is generally achieved with a larger inductor
value, which produces a lower switching frequency. For a
given inductor type, however, as inductance is increased
DC resistance (DCR) also increases. Higher DCR translates into higher copper losses and lower current rating,
both of which place an upper limit on the inductance. The
recommended range of inductor values for small surface
mount inductors as a function of peak current is shown in
Figure 4. The values in this range are a good compromise
between the trade-offs discussed above. For applications
3632fc
LTC3632
APPLICATIONS INFORMATION
350
VOUT = 5V
ISET OPEN
SWITCHING FREQUENCY (kHz)
300
where board area is not a limiting factor, inductors with
larger cores can be used, which extends the recommended
range of Figure 4 to larger values.
L = 220µH
250
200
Inductor Core Selection
L = 470µH
150
100
L = 1000µH
50
L = 2200µH
0
5
10
15
20 25 30 35 40
INPUT VOLTAGE (V)
45
50
3632 F02
Figure 2. Switching Frequency for VOUT = 5V
SWITCHING FREQUENCY (kHz)
250
L = 220µH
VOUT = 3.3V
ISET OPEN
200
150
L = 470µH
100
L = 1000µH
50
0
L = 2200µH
5
10
15
20 25 30 35 40
INPUT VOLTAGE (V)
45
50
3632 F03
Figure 3. Switching Frequency for VOUT = 3.3V
INDUCTOR VALUE (µH)
10000
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of the more expensive ferrite cores. Actual
core loss is independent of core size for a fixed inductor
value but is very dependent of the inductance selected.
As the inductance increases, core losses decrease. Unfortunately, increased inductance requires more turns of
wire and therefore copper losses will increase.
Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can
concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard,” which means that
inductance collapses abruptly when the peak design current
is exceeded. This results in an abrupt increase in inductor
ripple current and consequently output voltage ripple. Do
not allow the core to saturate!
Different core materials and shapes will change the
size/current and price/current relationship of an inductor.
Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate energy but generally
cost more than powdered iron core inductors with similar
characteristics. The choice of which style inductor to use
mainly depends on the price vs size requirements and any
radiated field/EMI requirements. New designs for surface
mount inductors are available from Coiltronics, Coilcraft,
TDK, Toko, Sumida and Vishay.
CIN and COUT Selection
1000
100
10
50
The input capacitor, CIN, is needed to filter the trapezoidal
current at the source of the top high side MOSFET. To
prevent large ripple voltage, a low ESR input capacitor
sized for the maximum RMS current should be used.
Approximate RMS current is given by:
PEAK INDUCTOR CURRENT (mA)
IRMS = IOUT(MAX) •
3632 F04
Figure 4. Recommended Inductor Values for Maximum Efficiency
VOUT
VIN
•
−1
VIN
VOUT
3632fc
11
LTC3632
APPLICATIONS INFORMATION
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations
do not offer much relief. Note that ripple current ratings
from capacitor manufacturers are often based only on
2000 hours of life which makes it advisable to further
derate the capacitor, or choose a capacitor rated at a
higher temperature than required. Several capacitors may
also be paralleled to meet size or height requirements in
the design.
The output capacitor, COUT, filters the inductor’s ripple current and stores energy to satisfy the load current when the
LTC3632 is in sleep. The output ripple has a lower limit of
VOUT/160 due to the 5mV typical hysteresis of the feedback comparator. The time delay of the comparator adds
an additional ripple voltage that is a function of the load
current. During this delay time, the LTC3632 continues to
switch and supply current to the output. The output ripple
can be approximated by:
I
4 • 10 – 6 VOUT
ΔVOUT ≈ PEAK – ILOAD
+
2
C
160
OUT
The output ripple is a maximum at no load and approaches
lower limit of VOUT/160 at full load. Choose the output
capacitor COUT to limit the output voltage ripple at minimum load current.
The value of the output capacitor must be large enough
to accept the energy stored in the inductor without a large
change in output voltage. Setting this voltage step equal to
1% of the output voltage, the output capacitor must be:
I
2
PEAK
COUT > 50 • L •
VOUT
Typically, a capacitor that satisfies the voltage ripple requirement is adequate to filter the inductor ripple. To avoid
overheating, the output capacitor must also be sized to
handle the ripple current generated by the inductor. The
worst-case ripple current in the output capacitor is given
by IRMS = IPEAK/2. Multiple capacitors placed in parallel
may be needed to meet the ESR and RMS current handling
requirements.
Dry tantalum, special polymer, aluminum electrolytic,
and ceramic capacitors are all available in surface mount
packages. Special polymer capacitors offer very low ESR
but have lower capacitance density than other types.
Tantalum capacitors have the highest capacitance density
but it is important only to use types that have been surge
tested for use in switching power supplies. Aluminum
electrolytic capacitors have significantly higher ESR but
can be used in cost-sensitive applications provided that
consideration is given to ripple current ratings and longterm reliability. Ceramic capacitors have excellent low ESR
characteristics but can have high voltage coefficient and
audible piezoelectric effects. The high quality factor (Q)
of ceramic capacitors in series with trace inductance can
also lead to significant ringing.
Using Ceramic Input and Output Capacitors
Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. However, care must
be taken when these capacitors are used at the input and
output. When a ceramic capacitor is used at the input and
the power is supplied by a wall adapter through long wires,
a load step at the output can induce ringing at the input,
VIN. At best, this ringing can couple to the output and be
mistaken as loop instability. At worst, a sudden inrush
of current through the long wires can potentially cause a
voltage spike at VIN large enough to damage the LTC3632.
For applications with inductive source impedance, such
as a long wire, a series RC network may be required in
parallel with CIN to dampen the ringing of the input supply.
Figure 5 shows this circuit and the typical values required
to dampen the ringing.
LTC3632
LIN
VIN
R=
LIN
CIN
4 • CIN
3632 F05
CIN
Figure 5. Series RC to Reduce VIN Ringing
3632fc
12
LTC3632
APPLICATIONS INFORMATION
Output Voltage Programming
The output voltage is set by an external resistive divider
according to the following equation:
R1
VOUT = 0.8V • 1+
R2
VSUPPLY
VIN
4.7M
LTC3632
RUN
The resistive divider allows the VFB pin to sense a fraction
of the output voltage as shown in Figure 6. Output voltage
adjustment range is from 0.8V to VIN.
VOUT
R1
VFB
LTC3632
controller is enabled. Figure 7 shows examples of configurations for driving the RUN pin from logic.
R2
GND
3632 F06
Figure 6. Setting the Output Voltage
To minimize the no-load supply current, resistor values in
the megohm range should be used; however, large resistor
values should be used with caution. The feedback divider
is the only load current when in shutdown. If PCB leakage current to the output node or switch node exceeds
the load current, the output voltage will be pulled up. In
normal operation, this is generally a minor concern since
the load current is much greater than the leakage. The
increase in supply current due to the feedback resistors
can be calculated from:
V
V
ΔIVIN = OUT • OUT
R1+R2 VIN
Run Pin with Programmable Hysteresis
The LTC3632 has a low power shutdown mode controlled
by the RUN pin. Pulling the RUN pin below 0.7V puts the
LTC3632 into a low quiescent current shutdown mode
(IQ ~ 3µA). When the RUN pin is greater than 1.2V, the
LTC3632
RUN
3632 F07
Figure 7. RUN Pin Interface to Logic
The RUN pin can alternatively be configured as a precise
undervoltage lockout (UVLO) on the VIN supply with a
resistive divider from VIN to ground. The RUN pin comparator nominally provides 10% hysteresis when used in
this method; however, additional hysteresis may be added
with the use of the HYST pin. The HYST pin is an opendrain output that is pulled to ground whenever the RUN
comparator is not tripped. A simple resistive divider can
be used as shown in Figure 8 to meet specific VIN voltage
requirements.
VIN
R1
RUN
R2
LTC3632
HYST
R3
3632 F08
Figure 8. Adjustable Undervoltage Lockout
Specific values for these UVLO thresholds can be computed
from the following equations:
R1
Rising VIN UVLO Threshold = 1.21V • 1+
R2
R1
Falling VIN UVLO Threshold = 1.10V • 1+
R2 +R3
3632fc
13
LTC3632
APPLICATIONS INFORMATION
The minimum value of these thresholds is limited to the
internal VIN UVLO thresholds that are shown in the Electrical Characteristics table. The current that flows through
this divider will directly add to the shutdown, sleep and
active current of the LTC3632, and care should be taken to
minimize the impact of this current on the overall efficiency
of the application circuit. Resistor values in the megohm
range may be required to keep the impact on quiescent
shutdown and sleep currents low. Be aware that the HYST
pin cannot be allowed to exceed its absolute maximum
rating of 6V. To keep the voltage on the HYST pin from
exceeding 6V, the following relation should be satisfied:
R3
VIN(MAX) •
< 6V
R1+R2 +R3
The RUN pin may also be directly tied to the VIN supply
for applications that do not require the programmable
undervoltage lockout feature. In this configuration, switching is enabled when VIN surpasses the internal undervoltage
lockout threshold.
Soft-Start
The internal 0.75ms soft-start is implemented by ramping
both the effective reference voltage from 0V to 0.8V and the
peak current limit set by the ISET pin (10mA to 50mA).
To increase the duration of the reference voltage soft-start,
place a capacitor from the SS pin to ground. An internal
5µA pull-up current will charge this capacitor, resulting in
a soft-start ramp time given by:
tSS = CSS •
0.8V
5µA
When the LTC3632 detects a fault condition (input supply
undervoltage or overvoltage) or when the RUN pin falls
below 1.1V, the SS pin is quickly pulled to ground and the
internal soft-start timer is reset. This ensures an orderly
restart when using an external soft-start capacitor.
The duration of the 0.75ms internal peak current softstart may be increased by placing a capacitor from the
ISET pin to ground. The peak current soft-start will ramp
from 10mA to the final peak current value determined by a
resistor from ISET to ground. A 1µA current is sourced out
of the ISET pin. With only a capacitor connected between
ISET and ground, the peak current ramps linearly from
10mA to 50mA, and the peak current soft-start time can
be expressed as:
tSS(ISET) = CISET •
0.8V
1µA
A linear ramp of peak current appears as a quadratic
waveform on the output voltage. For the case where the
peak current is reduced by placing a resistor from ISET
to ground, the peak current offset ramps as a decaying
exponential with a time constant of RISET • CISET. For this
case, the peak current soft-start time is approximately
3 • RISET • CISET.
Unlike the SS pin, the ISET pin does not get pulled to ground
during an abnormal event; however, if the ISET pin is floating (programmed to 50mA peak current), the SS and ISET
pins may be tied together and connected to a capacitor
to ground. For this special case, both the peak current
and the reference voltage will soft-start on power‑up and
after fault conditions. The ramp time for this combination
is CSS(ISET) • (0.8V/6µA).
Efficiency Considerations
The efficiency of a switching regulator is equal to the output
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of
the losses: VIN operating current and I2R losses. The VIN
operating current dominates the efficiency loss at very
low load currents whereas the I2R loss dominates the
efficiency loss at medium to high load currents.
1. The VIN operating current comprises two components:
The DC supply current as given in the electrical characteristics and the internal MOSFET gate charge currents.
3632fc
14
LTC3632
APPLICATIONS INFORMATION
The gate charge current results from switching the gate
capacitance of the internal power MOSFET switches.
Each time the gate is switched from high to low to
high again, a packet of charge, dQ, moves from VIN to
ground. The resulting dQ/dt is the current out of VIN
that is typically larger than the DC bias current.
2. I2R losses are calculated from the resistances of the
internal switches, RSW, and external inductor RL. When
switching, the average output current flowing through
the inductor is “chopped” between the high side PMOS
switch and the low side NMOS switch. Thus, the series
resistance looking back into the switch pin is a function
of the top and bottom switch RDS(ON) values and the
duty cycle (DC = VOUT/VIN) as follows:
RSW = (RDS(ON)TOP)DC + (RDS(ON)BOT)(1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Characteristics curves. Thus, to obtain the I2R losses, simply add
RSW to RL and multiply the result by the square of the
average output current:
I2R Loss = IO2(RSW + RL)
Other losses, including CIN and COUT ESR dissipative
losses and inductor core losses, generally account for
less than 2% of the total power loss.
Thermal Considerations
The LTC3632 does not dissipate much heat due to its high
efficiency and low peak current level. Even in worst-case
conditions (high ambient temperature, maximum peak
current and high duty cycle), the junction temperature will
exceed ambient temperature by only a few degrees.
Design Example
As a design example, consider using the LTC3632 in an
application with the following specifications: VIN = 24V,
VOUT = 3.3V, IOUT = 20mA, f = 250kHz. Furthermore, assume for this example that switching should start when
VIN is greater than 12V and should stop when VIN is less
than 8V.
First, calculate the inductor value that gives the required
switching frequency:
3.3V
L=
250kHz • 50mA
3.3V
• 1–
≅ 220µH
24V
Next, verify that this value meets the LMIN requirement.
For this input voltage and peak current, the minimum
inductor value is:
L MIN =
24V • 100ns
≅ 48µH
50mA
Therefore, the minimum inductor requirement is satisfied,
and the 220μH inductor value may be used.
Next, CIN and COUT are selected. For this design, CIN should
be size for a current rating of at least:
IRMS = 20mA •
3.3V
24V
•
– 1 ≅ 7mA RMS
24V
3.3V
Due to the low peak current of the LTC3632, decoupling
the VIN supply with a 1µF capacitor is adequate for most
applications.
COUT will be selected based on the output voltage ripple
requirement. For a 1% (33mV) output voltage ripple at no
load, COUT can be calculated from:
50mA • 4 • 10 –6
C OUT =
3.3V
2 33mV –
160
An 8.1µF capacitor gives this typical output voltage ripple at
no load. Choose a 10µF capacitor as a standard value.
The output voltage can now be programmed by choosing
the values of R1 and R2. Choose R2 = 240k and calculate
R1 as:
V
R1= OUT – 1 • R2 = 750k
0.8V
3632fc
15
LTC3632
APPLICATIONS INFORMATION
The undervoltage lockout requirement on VIN can be
satisfied with a resistive divider from VIN to the RUN and
HYST pins. Choose R1 = 2M and calculate R2 and R3 as
follows:
1.21V
• R1= 224k
R2 =
–
1.21V
V
IN(RISING)
1.1V
• R1– R2 = 90.8k
R3 =
–
1.1V
V
IN(FALLING)
3. Keep the switching node, SW, away from all sensitive small-signal nodes. The rapid transitions on the
switching node can couple to high impedance nodes,
in particular VFB, and create increased output ripple.
4. Flood all unused area on all layers with copper. Flooding
with copper will reduce the temperature rise of power
components. You can connect the copper areas to any
DC net (VIN, VOUT, GND or any other DC rail in your
system).
2
VIN
Choose standard values for R2 = 226k and R3 = 91k.
The ISET pin should be left open in this example to select
maximum peak current (50mA). Figure 9 shows a complete
schematic for this design example.
5
CIN
7
4
CSS
220µH
VIN
24V
VIN
1µF
2M
SW
LTC3632
RUN
91k
ISET
SS
750k
VFB
HYST
GND
240k
226k
VOUT
3.3V
20mA
10µF
VIN
SW
LTC3632
VFB
RUN
HYST
SS
ISET
1
L1
VOUT
R1
6
COUT
3
GND
8, 9
R2
RSET
3632 F10a
L1
3632 F09
VIN
CIN
VOUT
COUT
Figure 9. 24V to 3.3V, 20mA Regulator at 250kHz
R1
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3632. Check the following in your layout:
1. Large switched currents flow in the power switches
and input capacitor. The loop formed by these components should be as small as possible. A ground plane
is recommended to minimize ground impedance.
2. Connect the (+) terminal of the input capacitor, CIN, as
close as possible to the VIN pin. This capacitor provides
the AC current into the internal power MOSFETs.
R2
RSET
CSS
GND
3632 F10b
VIAS TO GROUND PLANE
VIAS TO INPUT SUPPLY (VIN)
OUTLINE OF LOCAL GROUND PLANE
Figure 10. Layout Example
3632fc
16
LTC3632
TYPICAL APPLICATIONS
L1
1mH
VIN
5V TO 50V
VIN
CIN
4.7µF
SW
LTC3632
RUN
HYST
ISET
R1
4.2M
VFB
COUT
100µF
VOUT
5V
20mA
R2
800k
SS
CSS
470nF
GND
3632 F11
CIN: TDK C5750X7R2A475MT
COUT: AVX 1812D107MAT
L1: COILCRAFT LPS6235-105ML
Figure 11. High Efficiency 5V Regulator
3.3V, 20mA Regulator with Peak Current Soft-Start, Small Size
VIN
4.5V TO 24V
L1
470µH
VIN
CIN
1µF
SW
R1
294k
LTC3632
RUN
ISET
SS
CSS
0.1µF
VFB
COUT
10µF
VOUT
3.3V
20mA
R2
93.1k
HYST
GND
Soft-Start Waveforms
OUTPUT
VOLTAGE
1V/DIV
INDUCTOR
CURRENT
20mA/DIV
3632 TA03a
2ms/DIV
CIN: TDK C3216X7R1E105KT
COUT: AVX 08056D106KAT2A
L1: MURATA LQH43CN471K03
Positive-to-Negative Converter
Maximum Load Current vs Input Voltage
L1
1mH
CIN
1µF
VIN
20
SW
LTC3632
RUN
ISET
R1
1M
VFB
HYST
SS
GND
R2
71.5k
CIN: TDK C3225X7R1H105KT
COUT: MURATA GRM32DR71C106KA01
L1: TYCO/COEV DQ6545-102M
COUT
10µF
VOUT
–12V
3632 TA04a
MAXIMUM LOAD CURRENT (mA)
VIN
4.5V TO 38V
3632 TA03b
ISET OPEN
VOUT = –3V
VOUT = –5V
15
VOUT = –12V
10
5
5
10
15
20 25 30 35 40
INPUT VOLTAGE (V)
45
50
3632 TA04b
3632fc
17
LTC3632
TYPICAL APPLICATIONS
Small Size, Limited Peak Current, 4mA Regulator
VIN
7V TO 50V
L1
2.2mH
CIN
1µF
R3
470k
R4
100k
R5
33k
VIN
SW
R1
470k
LTC3632
RUN
VFB
COUT
10µF
VOUT
5V
4mA
R2
88.7k
HYST
SS
ISET GND
3632 TA05a
CIN: TDK C3225X7R1H105KT
COUT: AVX 08056D106KAT2A
L1: MURATA LQH43NN222K03
High Efficiency 15V, 4mA Regulator
95
L1
10mH
CIN
1µF
VIN
SW
LTC3632
RUN
ISET
SS
VFB
HYST
GND
CIN: AVX 18125C105KAT2A
COUT: TDK C3216X7R1E475KT
L1: COILCRAFT LPS6235-106ML
R1
3M
R2
169k
3642 TA07a
COUT
4.7µF
VOUT
15V
4mA
90
VIN = 24V
85
EFFICIENCY (%)
VIN
15V TO 50V
Efficiency vs Load Current
80
75
VIN = 36V
VIN = 48V
70
65
60
55
50
0.1
1
LOAD CURRENT (mA)
4
3632 TA07b
3632fc
18
LTC3632
PACKAGE DESCRIPTION
DD Package
8-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1698 Rev C)
0.70 ±0.05
3.5 ±0.05
1.65 ±0.05
2.10 ±0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50
BSC
2.38 ±0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
PIN 1
TOP MARK
(NOTE 6)
0.200 REF
3.00 ±0.10
(4 SIDES)
R = 0.125
TYP
5
0.40 ± 0.10
8
1.65 ± 0.10
(2 SIDES)
0.75 ±0.05
4
0.25 ± 0.05
1
(DD8) DFN 0509 REV C
0.50 BSC
2.38 ±0.10
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON TOP AND BOTTOM OF PACKAGE
3632fc
19
LTC3632
PACKAGE DESCRIPTION
MS8E Package
8-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1662 Rev I)
BOTTOM VIEW OF
EXPOSED PAD OPTION
1.88
(.074)
1
1.88 ± 0.102
(.074 ± .004)
0.29
REF
1.68
(.066)
0.889 ± 0.127
(.035 ± .005)
0.05 REF
5.23
(.206)
MIN
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
1.68 ± 0.102 3.20 – 3.45
(.066 ± .004) (.126 – .136)
8
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
0.65
(.0256)
BSC
0.42 ± 0.038
(.0165 ± .0015)
TYP
8
7 6 5
0.52
(.0205)
REF
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
DETAIL “A”
0° – 6° TYP
GAUGE PLANE
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
1
2 3
4
1.10
(.043)
MAX
0.86
(.034)
REF
0.18
(.007)
SEATING
PLANE
0.22 – 0.38
(.009 – .015)
TYP
0.65
(.0256)
BSC
0.1016 ± 0.0508
(.004 ± .002)
MSOP (MS8E) 0910 REV I
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD
SHALL NOT EXCEED 0.254mm (.010") PER SIDE.
3632fc
20
LTC3632
REVISION HISTORY
(Revision history begins at Rev B)
REV
DATE
DESCRIPTION
B
6/10
Text updates in Description
PAGE NUMBER
Updates to Electrical Characteristics
Updates to graphs G08, G09, G17, G18, G19
Updated description for Pins 8 and 9 in Pin Functions
C
10/10
1
3
4, 5, 6
6
Text updates in Operation section
8, 9
Text updates in Applications Information section
13
Figure 10 graphic added
16
Asterisk and related text added to Typical Application
22
Related Parts updated
22
Updates to CIN and COUT Selection section
12
Updates to Efficiency Considerations section
15
3632fc
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
21
LTC3632
TYPICAL APPLICATION
5V, 20mA Regulator for Automotive Applications
VBATT
4.5V TO 50V
TRANSIENTS UP TO 60V
L1
1mH
CIN
1µF
VIN
SW
LTC3632
RUN
ISET
SS
VFB
HYST
GND
CIN: TDK C3225X7R2A105M
COUT: KEMET C1210C106K4RAC
L1: COILTRONICS DRA73-102-R
R1
470k
COUT
10µF
VOUT*
5V
20mA
R2
88.7k
3632 TA06a
*VOUT = VBATT FOR VBATT < 5V
RELATED PARTS
PART NUMBER
LTC3631/LTC3631-3.3/
LTC3631-5
LTC3642/LTC3642-3.3/
LTC3642-5
LTC1474
LT1934/LT1934-1
LT1939
LT3437
LT3470
LT3685
DESCRIPTION
COMMENTS
45V, 100mA Synchronous Micropower Step-Down DC/DC Converter VIN: 4.5V to 45V (60VMAX), VOUT(MIN) = 0.8V, IQ = 12µA,
ISD = 3µA, 3mm × 3mm DFN8, MSOP8E
45V, 50mA Synchronous Micropower Step-Down DC/DC Converter VIN: 4.5V to 45V (60VMAX), VOUT(MIN) = 0.8V, IQ = 12µA,
ISD = 3µA, 3mm × 3mm DFN8, MSOP8E
VIN: 3V to 18V, VOUT(MIN) = 1.2V, IQ = 10µA, ISD = 6µA,
18V, 250mA (IOUT), High Efficiency Step-Down DC/DC Converter
MSOP8
VIN: 3.2V to 34V, VOUT(MIN) = 1.25V, IQ = 12µA, ISD < 1µA,
36V, 250mA (IOUT), Micropower Step-Down DC/DC Converter with
Burst Mode Operation
ThinSOT™ Package
25V, 2A, 2.5MHz High Efficiency DC/DC Converter and LDO
Controller
60V, 400mA (IOUT), Micropower Step-Down DC/DC Converter with
Burst Mode Operation
40V, 250mA (IOUT), High Efficiency Step-Down DC/DC Converter
with Burst Mode Operation
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter
VIN: 3.6V to 25V, VOUT(MIN) = 0.8V, IQ = 2.5mA, ISD < 10µA,
3mm × 3mm DFN10
VIN: 3.3V to 60V, VOUT(MIN) = 1.25V, IQ = 100µA, ISD < 1µA,
3mm × 3mm DFN10, TSSOP16E
VIN: 4V to 40V, VOUT(MIN) = 1.2V, IQ = 26µA, ISD < 1µA,
2mm × 3mm DFN8, ThinSOT
VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70µA, ISD < 1µA,
3mm × 3mm DFN10, MSOP10E
3632fc
22 Linear Technology Corporation
LT 1010 REV C • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
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FAX: (408) 434-0507 ● www.linear.com
LINEAR TECHNOLOGY CORPORATION 2009