LTC3703
100V Synchronous
Switching Regulator
Controller
Description
Features
High Voltage Operation: Up to 100V
Large 1Ω Gate Drivers
No Current Sense Resistor Required
Step-Up or Step-Down DC/DC Converter
Dual N-Channel MOSFET Synchronous Drive
Excellent Line and Load Transient Response
Programmable Constant Frequency: 100kHz to
600kHz
n ±1% Reference Accuracy
n Synchronizable up to 600kHz
n Selectable Pulse-Skip Mode Operation
n Low Shutdown Current: 50µA Typ
n Programmable Current Limit
n Undervoltage Lockout
n Programmable Soft-Start
n 16-Pin Narrow SSOP and 28-Pin SSOP Packages
n
n
n
n
n
n
The LTC®3703 is a synchronous step-down switching
regulator controller that can directly step down voltages
from up to 100V, making it ideal for telecom and automotive applications. The LTC3703 drives external N-channel
MOSFETs using a constant frequency (up to 600kHz),
voltage mode architecture.
n
A precise internal reference provides 1% DC accuracy.
A high bandwidth error amplifier and patented line feedforward compensation provide very fast line and load
transient response. Strong 1Ω gate drivers allow the
LTC3703 to drive multiple MOSFETs for higher current applications. The operating frequency is user programmable
from 100kHz to 600kHz and can also be synchronized to
an external clock for noise-sensitive applications. Current limit is programmable with an external resistor and
utilizes the voltage drop across the synchronous MOSFET
to eliminate the need for a current sense resistor. For applications requiring up to 60V operation with logic-level
MOSFETS, refer to the LTC3703-5 data sheet.
Applications
48V Telecom and Base Station Power Supplies
Networking Equipment, Servers
n Automotive and Industrial Control
n
n
PARAMETER
Maximum VIN
MOSFET Gate Drive
VCC UV+
VCC UV–
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and
ThinSOT and No RSENSE are trademarks of Linear Technology Corporation. All other trademarks
are the property of their respective owners. Protected by U.S. Patents, including 5408150,
5055767, 6677210, 5847554, 5481178, 6304066, 6580258.
LTC3703-5
60V
4.5V to 15V
3.7V
3.1V
LTC3703
100V
9.3V to 15V
8.7V
6.2V
Typical Application
VCC
9.3V TO 15V
+
22µF
25V
BAS19
MODE/SYNC VIN
10k
1000pF 470pF
15k
8.06k
1%
0.1µF
113k
1%
100Ω
2200pF
+
fSET
BOOST
LTC3703
TG
COMP
FB
SW
IMAX
VCC
INV
RUN/SS
GND
DRVCC
BG
BGRTN
100
68µF
0.1µF
8µH
270µF
16V
10Ω
VIN = 25V
VIN = 50V
95
Si7456DP
VOUT
12V
5A
+
EFFICIENCY (%)
30k
Efficiency vs Load Current
VIN
15V TO 100V
VIN = 75V
90
85
Si7456DP
MBR1100
10µF
80
1µF
3703 F01
Figure 1. High Efficiency High Voltage Step-Down Converter
0
1
3
2
LOAD (A)
4
5
3703 F01b
3703fc
1
LTC3703
Absolute Maximum Ratings (Note 1)
Supply Voltages
VCC, DRVCC ........................................... –0.3V to 15V
(DRVCC – BGRTN), (BOOST – SW)........ –0.3V to 15V
BOOST.................................................. –0.3V to 115V
BGRTN........................................................ –5V to 0V
VIN Voltage............................................... –0.3V to 100V
SW Voltage (Note 10).................................. –1V to 100V
RUN/SS Voltage........................................... –0.3V to 5V
MODE/SYNC, INV Voltages........................ –0.3V to 15V
fSET, FB, IMAX Voltages................................ –0.3V to 3V
Peak Output Current 125°C
RUN/SS = 0V
MIN
l
TYP
9.3
15
l
l
UNITS
V
100
V
1.7
50
2.5
mA
µA
0
0
5
5
µA
µA
360
360
0
500
800
5
µA
µA
µA
0.800
0.808
0.812
V
V
0.007
0.05
%/V
0.01
0.1
%
0.81
0.87
V
l
l
MAX
Main Control Loop
VFB
Feedback Voltage
(Note 4)
l
∆VFB(LINE)
Feedback Voltage Line Regulation
9V < VCC < 15V (Note 4)
l
∆VFB(LOAD)
VMODE/SYNC
Feedback Voltage Load Regulation
1V < VCOMP < 2V (Note 4)
l
MODE/SYNC Threshold
MODE/SYNC Rising
∆VMODE/SYNC
MODE/SYNC Hysteresis
0.75
20
IMODE/SYNC
MODE/SYNC Current
VINV
Invert Threshold
IINV
Invert Current
0 ≤ VINV ≤ 15V
IVIN
VIN Sense Input Current
VIN = 100V
RUN/SS = 0V, VIN = 10V
IMAX
IMAX Source Current
VIMAX = 0V
VOS(IMAX)
VIMAX Offset Voltage
VRUN/SS
Shutdown Threshold
IRUN/SS
VUV
0.792
0.788
0 ≤ VMODE/SYNC ≤ 15V
1
mV
0
1
µA
1.5
2
V
0
1
µA
100
0
140
1
µA
µA
10.5
12
13.5
µA
|VSW| – VIMAX at IRUN/SS = 0µA
H Grade
–25
–25
10
10
55
65
mV
mV
0.7
0.9
1.2
V
RUN/SS Source Current
RUN/SS = 0V
2.5
4
5.5
µA
Maximum RUN/SS Sink Current
|VSW| – VIMAX ≥ 200mV, VRUN/SS = 3V
Undervoltage Lockout
VCC Rising
VCC Falling
RSET = 25k
l
l
9
17
25
µA
8.0
5.7
8.7
6.2
9.3
6.8
V
V
270
300
330
kHz
600
kHz
Oscillator
fOSC
Oscillator Frequency
fSYNC
External Sync Frequency Range
tON(MIN)
Minimum On-Time
DCMAX
Maximum Duty Cycle
100
200
f < 200kHz
89
93
1.5
2
1.5
2
ns
96
%
Driver
IBG(PEAK)
BG Driver Peak Source Current
RBG(SINK)
BG Driver Pull-Down RDS(ON)
ITG(PEAK)
TG Driver Peak Source Current
RTG(SINK)
TG Driver Pull-Down RDS(ON)
(Note 8)
1
(Note 8)
1
A
1.5
Ω
A
1.5
Ω
Feedback Amplifier
AVOL
Op Amp DC Open Loop Gain
(Note 4)
fU
Op Amp Unity Gain Crossover Frequency
(Note 6)
IFB
FB Input Current
0 ≤ VFB ≤ 3V
ICOMP
COMP Sink/Source Current
74
85
dB
25
MHz
0
±5
±10
1
µA
mA
3703fc
3
LTC3703
Electrical Characteristics
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3703E is guaranteed to meet performance specifications from
0°C to 85°C. Specifications over the –40°C to 85°C operating temperature
range are assured by design, characterization and correlation with statistical
process controls. The LTC3703I is guaranteed over the full –40°C to 125°C
operating junction temperature range. The LTC3703H is guaranteed over the
full –40°C to 150°C operating junction temperature range.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
LTC3703: TJ = TA + (PD • 100 °C/W) G Package
Note 4: The LTC3703 is tested in a feedback loop that servos VFB to the
reference voltage with the COMP pin forced to a voltage between 1V and 2V.
Note 5: The dynamic input supply current is higher due to the power
MOSFET gate charge being delivered at the switching frequency (QG • fOSC).
Note 6: Guaranteed by design. Not subject to test.
Note 7: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
Note 8: RDS(ON) guaranteed by correlation to wafer level measurement.
Note 9: High junction temperatures degrade operating lifetimes. Operating
lifetime at junction temperatures greater than 125°C is derated to 1000 hours.
Note 10: Transient voltages (such as due to inductive ringing) are allowed
beyond this range provided that the voltage does not exceed 10V below
ground and duration does not exceed 20ns per switching cycle.
Typical Performance Characteristics TA = 25°C unless otherwise noted.
Efficiency vs Input Voltage
Efficiency vs Load Current
IOUT = 5A
95
95
EFFICIENCY (%)
EFFICIENCY (%)
90
IOUT = 0.5A
85
80
0
10
20
70
30 40 50 60
INPUT VOLTAGE (V)
VIN = 45V
90
VIN = 75V
85
IOUT
2A/DIV
80
70
80
VOUT
50mV/DIV
VIN = 15V
VOUT = 5V
f = 250kHz
PULSE SKIP ENABLED
75
VOUT = 12V
f = 300kHz
PULSE SKIP DISABLED
75
70
Load Transient Response
100
100
VIN = 50V
50µs/DIV
VOUT = 12V
1A TO 5A LOAD STEP
0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0
LOAD CURRENT (A)
3703 G02
3703 G01
VCC Current vs VCC Voltage
4
3.5
VCC Shutdown Current vs
VCC Voltage
VCC Current vs Temperature
100
90
2.0
VFB = 0V
1.5
1.0
80
COMP = 1.5V
3
VCC CURRENT (µA)
COMP = 1.5V
2.5
VCC CURRENT (mA)
VCC CURRENT (mA)
3.0
2
VFB = 0V
1
70
60
50
40
30
20
0.5
0
3703 G03
10
6
8
10
12
VCC VOLTAGE (V)
14
16
3703 G04
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3703 G05
0
6
8
12
10
VCC VOLTAGE (V)
14
16
3703 G06
3703fc
4
LTC3703
Typical Performance Characteristics
VCC Shutdown Current vs
Temperature
Reference Voltage vs
Temperature
70
Normalized Frequency vs
Temperature
0.803
1.20
1.15
55
50
45
40
0.802
NORMALIZED FREQUENCY
60
REFERENCE VOLTAGE (V)
VCC CURRENT (µA)
65
0.801
0.800
0.799
30
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
0.798
–50 –25
0
0.95
0.90
VCC = 10V
PEAK SOURCE CURRENT (A)
1.4
2.6
Driver Peak Source Current vs
Supply Voltage
3.0
1.6
VCC = 10V
1.2
2.4
RDS(ON) (Ω)
2.2
2.0
1.8
1.6
1.0
0.8
0.6
0.4
1.4
0.2
1.2
1.0
–50 –25
0
25
50
75
0
–50 –25
100 125 150
TEMPERATURE (°C)
0
70
1.0
0.5
0.7
15
3703 G13
8 9 10 11 12 13 14 15
7
DRVCC/BOOST VOLTAGE (V)
8
DRVCC, BOOST = 10V
7
RISE
50
40
30
FALL
20
10
14
6
RUN/SS Pull-Up Current vs
Temperature
RUN/SS CURRENT (µA)
RISE/FALL TIME (ns)
0.8
5
3703 G12
60
1.0
8
9 10 11 12 13
DRVCC/BOOST VOLTAGE (V)
1.5
Rise/Fall Time vs Gate
Capacitance
1.1
7
2.0
3703 G11
Driver Pull-Down RDS(ON) vs
Supply Voltage
6
2.5
0
25 50 75 100 125 150
TEMPERATURE (°C)
3703 G10
0.9
25 50 75 100 125 150
TEMPERATURE (°C)
3703 G09
Driver Pull-Down RDS(ON) vs
Temperature
3.0
2.8
0
3703 G08
Driver Peak Source Current vs
Temperature
PEAK SOURCE CURRENT (A)
1.00
0.80
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3703 G07
RDS(ON) (Ω)
1.05
0.85
35
0.6
1.10
0
6
5
4
3
2
1
0
6000
2000
4000
8000
GATE CAPACITANCE (pF)
10000
3703 G14
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
1573 G15
3703fc
5
LTC3703
Typical Performance Characteristics
RUN/SS Sink Current vs
SW Voltage
6
25
5
20
4
3
2
1
0
Max % DC vs RUN/SS Voltage
100
IMAX = 0.3V
90
80
15
MAX DUTY CYCLE (%)
RUN/SS SINK CURRENT (µA)
RUN/SS PULLUP CURRENT (µA)
RUN/SS Pull-Up Current vs
VCC Voltage
10
5
0
6
8
10
12
VCC VOLTAGE (V)
14
16
0
0.1
0.2 0.3 0.4 0.5
|SW| VOLTAGE (V)
0.6
0.7
12
1.0
2.0
1.5
RUN VOLTAGE (V)
2.5
Max % DC vs Frequency and
Temperature
100
VIN = 10V
95
60
VIN = 75V
40
VIN = 50V
VIN = 25V
20
25 50 75 100 125 150
TEMPERATURE (°C)
0
3.0
3703 G18
MAX DUTY CYCLE (%)
80
–45°C
90
25°C
85
90°C
80
150°C
75
0.5
0.75
1.00
1.25 1.50
COMP (V)
1.75
70
2.00
0
100
200 300 400 500
FREQUENCY (kHz)
3703 G20
3703 G19
Shutdown Threshold vs
Temperature
125°C
600
700
3703 G21
tON(MIN) vs Temperature
1.4
200
1.2
180
160
1.0
tON(MIN) (ns)
SHUTDOWN THRESHOLD (V)
20
–10
0.5
100
DUTY CYCLE (%)
IMAX SOURCE CURRENT (µA)
30
% Duty Cycle vs COMP Voltage
13
0.8
0.6
0.4
140
120
100
80
0.2
0
–50 –25
40
3703 G17
IMAX Current vs Temperature
0
50
0
3703 G16
11
–50 –25
60
10
–5
–10
70
60
0
25 50 75 100 125 150
TEMPERATURE (°C)
3703 G22
40
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3703 G23
3703fc
6
LTC3703
Pin Functions
(GN16/G28)
MODE/SYNC (Pin 1/Pin 6): Pulse-Skip Mode Enable/Sync
Pin. This multifunction pin provides pulse-skip mode
enable/disable control and an external clock input for
synchronization of the internal oscillator. Pulling this pin
below 0.8V or to an external logic-level synchronization
signal disables pulse-skip mode operation and forces
continuous operation. Pulling the pin above 0.8V enables
pulse-skip mode operation. This pin can also be connected
to a feedback resistor divider from a secondary winding
on the inductor to regulate a second output voltage.
fSET (Pin 2/Pin 7): Frequency Set. A resistor connected
to this pin sets the free running frequency of the internal
oscillator. See Applications Information section for resistor
value selection details.
COMP (Pin 3/Pin 8): Loop Compensation. This pin is connected directly to the output of the internal error amplifier.
An RC network is used at the COMP pin to compensate
the feedback loop for optimal transient response.
FB (Pin 4/Pin 9): Feedback Input. Connect FB through a
resistor divider network to VOUT to set the output voltage. Also connect the loop compensation network from
COMP to FB.
IMAX (Pin 5/Pin 10): Current Limit Set. The IMAX pin sets
the current limit comparator threshold. If the voltage drop
across the bottom MOSFET exceeds the magnitude of the
voltage at IMAX, the controller goes into current limit. The
IMAX pin has an internal 12µA current source, allowing the
current threshold to be set with a single external resistor
to ground. See the Current Limit Programming section
for more information on choosing RIMAX.
INV (Pin 6/Pin 11): Top/Bottom Gate Invert. Pulling this
pin above 2V sets the controller to operate in step-up
(boost) mode with the TG output driving the synchronous
MOSFET and the BG output driving the main switch. Below
1V, the controller will operate in step-down (buck) mode.
RUN/SS (Pin 7/Pin 13): Run/Soft-Start. Pulling RUN/SS
below 0.9V will shut down the LTC3703, turn off both of
the external MOSFET switches and reduce the quiescent
supply current to 50µA. A capacitor from RUN/SS to
ground will control the turn-on time and rate of rise of
the output voltage at power-up. An internal 4µA current
source pull-up at the RUN/SS pin sets the turn-on time
at approximately 750ms/µF.
GND (Pin 8/Pin 14): Ground Pin.
BGRTN (Pin 9/Pin 15): Bottom Gate Return. This pin
connects to the source of the pull-down MOSFET in the
BG driver and is normally connected to ground. Connecting a negative supply to this pin allows the synchronous
MOSFET’s gate to be pulled below ground to help prevent
false turn-on during high dV/dt transitions on the SW node.
See the Applications Information section for more details.
BG (Pin 10/Pin 19): Bottom Gate Drive. The BG pin drives
the gate of the bottom N-channel synchronous switch
MOSFET. This pin swings from BGRTN to DRVCC.
DRVCC (Pin 11/Pin 20): Driver Power Supply Pin. DRVCC
provides power to the BG output driver. This pin should
be connected to a voltage high enough to fully turn on
the external MOSFETs, normally 10V to 15V for standard
threshold MOSFETs. DRVCC should be bypassed to BGRTN
with a 10µF, low ESR (X5R or better) ceramic capacitor.
VCC (Pin 12/Pin 21): Main Supply Pin. All internal circuits
except the output drivers are powered from this pin. VCC
should be connected to a low noise power supply voltage
between 9V and 15V and should be bypassed to GND
(Pin 8) with at least a 0.1µF capacitor in close proximity
to the LTC3703.
SW (Pin 13/Pin 26): Switch Node Connection to Inductor
and Bootstrap Capacitor. Voltage swing at this pin is from a
Schottky diode (external) voltage drop below ground to VIN.
TG (Pin 14/Pin 27): Top Gate Drive. The TG pin drives the
gate of the top N-channel synchronous switch MOSFET.
The TG driver draws power from the BOOST pin and
returns to the SW pin, providing true floating drive to the
top MOSFET.
BOOST (Pin 15/Pin 28): Top Gate Driver Supply. The
BOOST pin supplies power to the floating TG driver. The
BOOST pin should be bypassed to SW with a low ESR
(X5R or better) 0.1µF ceramic capacitor. An additional fast
recovery Schottky diode from DRVCC to BOOST will create
a complete floating charge-pumped supply at BOOST.
VIN (Pin 16/Pin 1): Input Voltage Sense Pin. This pin is
connected to the high voltage input of the regulator and is
used by the internal feedforward compensation circuitry
to improve line regulation. This is not a supply pin.
3703fc
7
LTC3703
Functional Diagram
RSET
2
fSET
GN16
OVERCURRENT
12µA
4µA
–
5
IMAX
RMAX
+
50mV
–
±
+
RUN/SS
5
–
CSS
0.9V
3.2V
1
±
CHIP
SD
+
UVSD OTSD
MODE/SYNC
SYNC
DETECT
EXT SYNC
OSC
4
R2
R1
16
12
DB
–
INV
REVERSE
CURRENT
BOOST
TG
COMP
0.8V
FB
VCC
+
FORCED CONTINUOUS
3
INV
+
+ FB
–
÷
% DC
LIMIT
SW
–
PWM
+
DRIVE
LOGIC
VIN
DRVCC
BG
+MIN–
+MAX–
BGRTN
INV
VCC
(> VOUT, the
top MOSFETs’ “on” resistance is normally less important
for overall efficiency than its input capacitance at operating
frequencies above 300kHz. MOSFET manufacturers have
designed special purpose devices that provide reasonably low “on” resistance with significantly reduced input
capacitance for the main switch application in switching
regulators.
Selection criteria for the power MOSFETs include the “on”
resistance RDS(ON), input capacitance, breakdown voltage
and maximum output current.
The most important parameter in high voltage applications
is breakdown voltage BVDSS. Both the top and bottom
MOSFETs will see full input voltage plus any additional
ringing on the switch node across its drain-to-source during its off-time and must be chosen with the appropriate
3703fc
13
LTC3703
Applications Information
breakdown specification. Since many high voltage MOSFETs have higher threshold voltages (typically, VGS(MIN)
≥ 6V), the LTC3703 is designed to be used with a 9V to
15V gate drive supply (DRVCC pin).
For maximum efficiency, on-resistance RDS(ON) and input
capacitance should be minimized. Low RDS(ON) minimizes
conduction losses and low input capacitance minimizes
transition losses. MOSFET input capacitance is a combination of several components but can be taken from the
typical “gate charge” curve included on most data sheets
(Figure 9).
VIN
VGS
MILLER EFFECT
a
V
b
QIN
CMILLER = (QB – QA)/VDS
+
VGS
–
+V
DS
–
3703 F09
Figure 9. Gate Charge Characteristic
The curve is generated by forcing a constant input current into the gate of a common source, current source
loaded stage and then plotting the gate voltage versus
time. The initial slope is the effect of the gate-to-source
and the gate-to-drain capacitance. The flat portion of the
curve is the result of the Miller multiplication effect of the
drain-to-gate capacitance as the drain drops the voltage
across the current source load. The upper sloping line is
due to the drain-to-gate accumulation capacitance and
the gate-to-source capacitance. The Miller charge (the
increase in coulombs on the horizontal axis from a to b
while the curve is flat) is specified for a given VDS drain
voltage, but can be adjusted for different VDS voltages by
multiplying by the ratio of the application VDS to the curve
specified VDS values. A way to estimate the CMILLER term
is to take the change in gate charge from points a and b
on a manufacturers data sheet and divide by the stated
VDS voltage specified. CMILLER is the most important selection criteria for determining the transition loss term in
the top MOSFET but is not directly specified on MOSFET
data sheets. CRSS and COS are specified sometimes but
definitions of these parameters are not included.
When the controller is operating in continuous mode the
duty cycles for the top and bottom MOSFETs are given by:
Main Switch Duty Cycle =
VOUT
VIN
Synchronous Switch Duty Cycle =
VIN – VOUT
VIN
The power dissipation for the main and synchronous
MOSFETs at maximum output current are given by:
PMAIN =
VOUT
2
IMAX ) (1+ δ)RDS(ON) +
(
VIN
2 IMAX
VIN
(RDR )(CMILLER )•
2
1
1
+
(f)
VCC – VTH(IL) VTH(IL)
V −V
PSYNC = IN OUT (IMAX )2 (1+ δ)RDS(0N)
VIN
where δ is the temperature dependency of RDS(ON), RDR
is the effective top driver resistance (approximately 2Ω at
VGS = VMILLER), VIN is the drain potential and the change
in drain potential in the particular application. VTH(IL) is
the data sheet specified typical gate threshold voltage
specified in the power MOSFET data sheet at the specified
drain current. CMILLER is the calculated capacitance using
the gate charge curve from the MOSFET data sheet and
the technique described above.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which peak at the highest input voltage. For VIN < 25V,
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 25V, the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
a short circuit when the synchronous switch is on close
to 100% of the period.
3703fc
14
LTC3703
Applications Information
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs temperature curve, and
typically varies from 0.005/°C to 0.01/°C depending on
the particular MOSFET used.
Multiple MOSFETs can be used in parallel to lower RDS(ON)
and meet the current and thermal requirements if desired.
The LTC3703 contains large low impedance drivers capable
of driving large gate capacitances without significantly
slowing transition times. In fact, when driving MOSFETs
with very low gate charge, it is sometimes helpful to slow
down the drivers by adding small gate resistors (5Ω or less)
to reduce noise and EMI caused by the fast transitions.
Schottky Diode Selection
The Schottky diode D1 shown in Figure 1 conducts during
the dead time between the conduction of the power MOSFETs. This prevents the body diode of the bottom MOSFET
from turning on and storing charge during the dead time
and requiring a reverse recovery period that could cost
as much as 1% to 2% in efficiency. A 1A Schottky diode
is generally a good size for 3A to 5A regulators. Larger
diodes result in additional losses due to their larger junction capacitance. The diode can be omitted if the efficiency
loss can be tolerated.
Input Capacitor Selection
In continuous mode, the drain current of the top MOSFET
is approximately a square wave of duty cycle VOUT/VIN
which must be supplied by the input capacitor. To prevent
large input transients, a low ESR input capacitor sized for
the maximum RMS current is given by:
V
V
ICIN(RMS) ≅IO(MAX) OUT IN – 1
VIN VOUT
1/2
This formula has a maximum at VIN = 2VOUT, where IRMS =
IO(MAX)/2. This simple worst-case condition is commonly
used for design because even significant deviations do not
offer much relief. Note that the ripple current ratings from
capacitor manufacturers are often based on only 2000
hours of life. This makes it advisable to further derate the
capacitor or to choose a capacitor rated at a higher tempera-
ture than required. Several capacitors may also be placed in
parallel to meet size or height requirements in the design.
Because tantalum and OS-CON capacitors are not available
in voltages above 30V, for regulators with input supplies
above 30V, choice of input capacitor type is limited to
ceramics or aluminum electrolytics. Ceramic capacitors
have the advantage of very low ESR and can handle high
RMS current, however ceramics with high voltage ratings
(>50V) are not available with more than a few microfarads
of capacitance. Furthermore, ceramics have high voltage
coefficients which means that the capacitance values
decrease even more when used at the rated voltage. X5R
and X7R type ceramics are recommended for their lower
voltage and temperature coefficients. Another consideration when using ceramics is their high Q which if not
properly damped, may result in excessive voltage stress
on the power MOSFETs. Aluminum electrolytics have much
higher bulk capacitance, however, they have higher ESR
and lower RMS current ratings.
A good approach is to use a combination of aluminum
electrolytics for bulk capacitance and ceramics for low ESR
and RMS current. If the RMS current cannot be handled
by the aluminum capacitors alone, when used together,
the percentage of RMS current that will be supplied by the
aluminum capacitor is reduced to approximately:
% IRMS,ALUM ≈
1
1+(8fCRESR )2
•100%
where RESR is the ESR of the aluminum capacitor and C
is the overall capacitance of the ceramic capacitors. Using
an aluminum electrolytic with a ceramic also helps damp
the high Q of the ceramic, minimizing ringing.
Output Capacitor Selection
The selection of COUT is primarily determined by the ESR
required to minimize voltage ripple. The output ripple
(∆VOUT) is approximately equal to:
1
∆VOUT ≤ ∆IL ESR +
8fCOUT
3703fc
15
LTC3703
Applications Information
Since ∆IL increases with input voltage, the output ripple
is highest at maximum input voltage. ESR also has a significant effect on the load transient response. Fast load
transitions at the output will appear as voltage across the
ESR of COUT until the feedback loop in the LTC3703 can
change the inductor current to match the new load current
value. Typically, once the ESR requirement is satisfied the
capacitance is adequate for filtering and has the required
RMS current rating.
Manufacturers such as Nichicon, Nippon Chemi-Con and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON (organic semiconductor
dielectric) capacitor available from Sanyo has the lowest
product of ESR and size of any aluminum electrolytic at
a somewhat higher price. An additional ceramic capacitor
in parallel with OS-CON capacitors is recommended to
reduce the effect of their lead inductance.
In surface mount applications, multiple capacitors placed
in parallel may be required to meet the ESR, RMS current
handling and load step requirements. Dry tantalum, special
polymer and aluminum electrolytic capacitors are available
in surface mount packages. Special polymer capacitors
offer very low ESR but have lower capacitance density
than other types. Tantalum capacitors have the highest
capacitance density but it is important to only use types
that have been surge tested for use in switching power
supplies. Several excellent surge-tested choices are the
AVX TPS and TPSV or the KEMET T510 series. Aluminum
electrolytic capacitors have significantly higher ESR, but
can be used in cost-driven applications providing that
consideration is given to ripple current ratings and long
term reliability. Other capacitor types include Panasonic
SP and Sanyo POSCAPs.
Output Voltage
The LTC3703 output voltage is set by a resistor divider
according to the following formula:
R1
VOUT = 0.8V 1+
R2
The external resistor divider is connected to the output as
shown in the Functional Diagram, allowing remote voltage
sensing. The resultant feedback signal is compared with
the internal precision 800mV voltage reference by the
error amplifier. The internal reference has a guaranteed
tolerance of ±1%. Tolerance of the feedback resistors will
add additional error to the output voltage. 0.1% to 1%
resistors are recommended.
MOSFET Driver Supplies (DRVCC and BOOST)
The LTC3703 drivers are supplied from the DRVCC and
BOOST pins (see Figure 3), which have an absolute
maximum voltage of 15V. If the main supply voltage,
VIN, is higher than 15V a separate supply with a voltage
between 9V and 15V must be used to power the drivers.
If a separate supply is not available, one can easily be
generated from the main supply using one of the circuits
shown in Figure 10. If the output voltage is between 10V
and 15V, the output can be used to directly power the
drivers as shown in Figure 10a. If the output is below
10V, Figure 10b shows an easy way to boost the supply
voltage to a sufficient level. This boost circuit uses the
LT1613 in a ThinSOT™ package and a chip inductor for
minimal extra area ( VIN.
For hard shorts, the inductor current is limited only by the
input supply capability. Refer to Current Limit Programming for buck mode for further considerations for current
limit programming.
Boost Converter: Feedback Loop/Compensation
Compensating a voltage mode boost converter is unfortunately more difficult than for a buck converter. This is
due to an additional right-half plane (RHP) zero that is
present in the boost converter but not in a buck. The additional phase lag resulting from the RHP zero is difficult
if not impossible to compensate even with a Type 3 loop,
so the best approach is usually to roll off the loop gain at
a lower frequency than what could be achievable in buck
converter.
A typical gain/phase plot of a voltage mode boost converter
is shown in Figure 16. The modulator gain and phase can
be measured as described for a buck converter or can be
estimated as follows:
GAIN (COMP-to-VOUT DC gain) = 20Log(VOUT2/VIN)
Dominant Pole: fP =
VIN
1
•
VOUT 2π LC
Since significant phase shift begins at frequencies above
the dominant LC pole, choose a crossover frequency no
greater than about half this pole frequency. The gain of
the compensation network should equal –GAIN at this
frequency so that the overall loop gain is 0dB here. The
3703 F16
Figure 16. Transfer Function of Boost Modulator
compensation component to achieve this, using a Type 1
amplifier (see Figure 12), is:
G = 10–GAIN/20
C1 = 1/(2π • f • G • R1)
Run/Soft-Start Function
The RUN/SS pin is a multipurpose pin that provide a softstart function and a means to shut down the LTC3703.
Soft-start reduces the input supply’s surge current by
gradually increasing the duty cycle and can also be used
for power supply sequencing.
Pulling RUN/SS below 0.9V puts the LTC3703 into a low
quiescent current shutdown (IQ ≅ 50µA). This pin can be
driven directly from logic as shown in Figure 17. Releasing
the RUN/SS pin allows an internal 4µA current source to
RUN/SS
2V/DIV
VOUT
5V/DIV
IL
2A/DIV
VIN = 50V
ILOAD = 2A
CSS = 0.01µF
2ms/DIV
3703 F17
Figure 17. LTC3703 Start-Up Operation
3703fc
23
LTC3703
Applications Information
charge up the soft-start capacitor CSS. When the voltage
on RUN/SS reaches 0.9V, the LTC3703 begins operating
at its minimum on-time. As the RUN/SS voltage increases
from 1.4V to 3V, the duty cycle is allowed to increase from
0% to 100%. The duty cycle control minimizes input supply
inrush current and eliminates output voltage overshoot at
start-up and ensures current limit protection even with a
hard short. The RUN/SS voltage is internally clamped at 4V.
If RUN/SS starts at 0V, the delay before starting is
approximately:
tDELAY,START =
1V
C = (0.25s/µF)CSS
4µA SS
plus an additional delay, before the output will reach its
regulated value, of:
tDELAY,REG ≥
3V – 1V
C = (0.5s/µF)CSS
4µA SS
The start delay can be reduced by using diode D1 in
Figure 18.
3.3V
OR 5V
RUN/SS
RUN/SS
D1
CSS
CSS
3703 F18
Figure 18. RUN/SS Pin Interfacing
MODE/SYNC Pin (Operating Mode and Secondary
Winding Control)
The MODE/SYNC pin is a dual function pin that can be
used for enabling or disabling pulse-skip mode operation
and also as an external clock input for synchronizing the
internal oscillator (see next section). Pulse-skip mode is
enabled when the MODE/SYNC pin is above 0.8V and is
disabled, i.e., forced continuous, when the pin is below 0.8V.
In addition to providing a logic input to force continuous
operation and external synchronization, the MODE/SYNC
pin provides a means to regulate a flyback winding output
as shown in Figure 10c. The auxiliary output is taken from
a second winding on the core of the inductor, converting
it to a transformer. The auxiliary output voltage is set by
the main output voltage and the turns ratio of the extra
winding to the primary winding as follows:
VSEC ≈ (N + 1)VOUT
Since the secondary winding only draws current when the
synchronous switch is on, load regulation at the auxiliary
output will be relatively good as long as the main output
is running in continuous mode. As the load on the primary
output drops and the LTC3703 switches to pulse-skip
mode operation, the auxiliary output may not be able to
maintain regulation, especially if the load on the auxiliary
output remains heavy. To avoid this, the auxiliary output
voltage can be divided down with a conventional feedback
resistor string with the divided auxiliary output voltage fed
back to the MODE/SYNC pin. The MODE/SYNC threshold
is trimmed to 800mV with 20mV of hysteresis, allowing
precise control of the auxiliary voltage and is set as follows:
R1
VSEC(MIN) ≈ 0.8V 1+
R2
where R1 and R2 are shown in Figure 10c.
If the LTC3703 is operating in pulse-skip mode and the
auxiliary output voltage drops below VSEC(MIN), the MODE/
SYNC pin will trip and the LTC3703 will resume continuous operation regardless of the load on the main output.
Thus, the MODE/SYNC pin removes the requirement that
power must be drawn from the inductor primary in order
to extract power from the auxiliary winding. With the loop
in continuous mode (MODE/SYNC < 0.8V), the auxiliary
outputs may nominally be loaded without regard to the
primary output load.
The following table summarizes the possible states available on the MODE/SYNC pin:
Table 1
MODE/SYNC PIN
CONDITION
DC Voltage: 0V to 0.75V
Forced Continuous
Current Reversal Enabled
DC Voltage: ≥ 0.87V
Pulse-Skip Mode Operation
No Current Reversal
Feedback Resistors
Regulating a Secondary Winding
Ext. Clock: 0V to ≥ 2V
Forced Continuous
Current Reversal Enabled
3703fc
24
LTC3703
Applications Information
MODE/SYNC Pin (External Synchronization)
The internal LTC3703 oscillator can be synchronized to
an external oscillator by applying and clocking the MODE/
SYNC pin with a signal above 2VP-P . The internal oscillator
locks to the external clock after the second clock transition
is received. When external synchronization is detected,
LTC3703 will operate in forced continuous mode. If an
external clock transition is not detected for three successive periods, the internal oscillator will revert to the
frequency programmed by the RSET resistor. The internal
oscillator can synchronize to frequencies between 100kHz
and 600kHz, independent of the frequency programmed by
the RSET resistor. However, it is recommended that an RSET
resistor be chosen such that the frequency programmed
by the RSET resistor is close to the expected frequency of
the external clock. In this way, the best converter operation
(ripple, component stress, etc) is achieved if the external
clock signal is lost.
Fault Conditions: Output Overvoltage Protection
(Crowbar)
The output overvoltage crowbar is designed to blow a
system fuse in the input lead when the output of the regulator rises much higher than nominal levels. This condition
causes huge currents to flow, much greater than in normal
operation. This feature is designed to protect against a
shorted top MOSFET; it does not protect against a failure
of the controller itself.
The comparator (MAX in the Functional Diagram) detects
overvoltage faults greater than 5% above the nominal
output voltage. When this condition is sensed the top
MOSFET is turned off and the bottom MOSFET is forced
on. The bottom MOSFET remains on continuously for as
long as the 0V condition persists; if VOUT returns to a safe
level, normal operation automatically resumes.
Minimum On-Time Considerations (Buck Mode)
Minimum on-time tON(MIN) is the smallest amount of time
that the LTC3703 is capable of turning the top MOSFET on
and off again. It is determined by internal timing delays
and the amount of gate charge required to turn on the
top MOSFET. Low duty cycle applications may approach
this minimum on-time limit and care should be taken to
ensure that:
tON =
VOUT
>t
VIN • f ON(MIN)
where tON(MIN) is typically 200ns.
If the duty cycle falls below what can be accommodated
by the minimum on-time, the LTC3703 will begin to skip
cycles. The output will be regulated, but the ripple current
and ripple voltage will increase. If lower frequency operation is acceptable, the on-time can be increased above
tON(MIN) for the same step-down ratio.
Pin Clearance/Creepage Considerations
The LTC3703 is available in two packages (GN16 and G28)
both with identical functionality. The GN16 package gives
the smallest size solution, however the 0.013" (minimum)
space between pins may not provide sufficient PC board
trace clearance between high and low voltage pins in higher
voltage applications. Where clearance is an issue, the G28
package should be used. The G28 package has four unconnected pins between the all adjacent high voltage and
low voltage pins, providing 5(0.0106") = 0.053" clearance
which will be sufficient for most applications up to 100V.
For more information, refer to the printed circuit board
design standards described in IPC-2221 (www.ipc.org).
Efficiency Considerations
The efficiency of a switching regulator is equal to the output power divided by the input power (x100%). Percent
efficiency can be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power. It is often useful to analyze the individual
losses to determine what is limiting the efficiency and
what change would produce the most improvement.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3703 circuits: 1) LTC3703 VCC current, 2)
MOSFET gate current, 3) I2R losses, 4) Topside MOSFET
transition losses.
3703fc
25
LTC3703
Applications Information
1. VCC supply current. The VCC current is the DC supply
current given in the Electrical Characteristics table which
powers the internal control circuitry of the LTC3703.
Total supply current is typically about 2.5mA and usually
results in a small (
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