LTC3732
3-Phase, 5-Bit VID,
600kHz, Synchronous Buck
Switching Regulator Controller
DESCRIPTIO
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FEATURES
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3-Phase Current Mode Controller with
Onboard MOSFET Drivers
±5% Output Current Matching Optimizes Thermal
Performance and Size of Inductors and MOSFETs
4.5V ≤ VCC ≤ 7V; 4.5V ≤ VIN ≤ 32V
Differential Amplifier Accurately Senses VOUT
Reduced Input and Output Capacitance
Reduced Power Supply Induced Noise
VID DAC Programmable from 1.1V to 1.85V
(VRM9.0/9.1)
±10% Power Good Output Indicator
250kHz to 600kHz Per Phase, PLL, Fixed Frequency
PWM, Stage SheddingTM or Burst Mode® Operation
OPTI-LOOP® Compensation Minimizes COUT
Adjustable Soft-Start Current Ramping
Short-Circuit Shutdown Timer with Defeat Option
Overvoltage Soft Latch
Small 36-Lead Narrow (0.209") SSOP Package
QFN 5mm × 7mm 38-Lead Package
The LTC®3732 is a PolyPhase® synchronous step-down
switching regulator controller that drives all N-channel
external power MOSFET stages in a phase-lockable fixed
frequency architecture. The 3-phase controller drives its
output stages with 120° phase separation at frequencies
of up to 600kHz per phase to minimize the RMS current
losses in both the input and output filter capacitors. The 3phase technique effectively triples the fundamental frequency, improving transient response while operating
each controller at an optimal frequency for high efficiency
and ease of thermal design. Light load efficiency is optimized by using a choice of output Stage Shedding or Burst
Mode technology.
Desktop Computers
High Performance Notebook Computers
High Output Current DC/DC Power Supplies
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode, OPTI-LOOP and PolyPhase are registered trademarks of Linear Technology
Corporation. Stage Shedding is a trademark of Linear Technology Corporation
A differential amplifier provides true remote sensing of both
the high and low side of the output voltage at load points.
Soft-start and a defeatable, timed short-circuit shutdown
protect the MOSFETs and the load. A foldback current
circuit also provides protection for the external MOSFETs
under short-circuit or overload conditions.
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APPLICATIO S
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TYPICAL APPLICATIO
VCC
4.5V TO 7V
VCC
TG1
LTC3732
10µF
BOOST1
BOOST2
BOOST3
0.1µF
SW3 SW2 SW1
PGOOD
PLLIN
POWER GOOD INDICATOR
OPTIONAL SYNC IN
PLLFLTR
1µH
0.003Ω
1µH
0.003Ω
1µH
0.003Ω
+
SW1
VIN
5V TO 28V
22µF
35V
BG1
SENSE1+
SENSE1–
TG2
VIN
VOUT
1.1V TO 1.85V
55A
SW2
BG2
PGND
5 VID BITS
VID0-VID4
680pF
5k
0.1µF
SENSE2+
SENSE2–
ITH
TG3
RUN/SS
SW3
SGND
EAIN
BG3
100pF
IN –
IN +
SENSE3+
SENSE3–
VIN
+
COUT
470µF
4V
3732 F01
Figure 1. High Current Triple Phase Step-Down Converter
3732f
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LTC3732
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AXI U
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ABSOLUTE
RATI GS
(Note 1)
Topside Driver Voltages (BOOSTN) ............ 38V to –0.3V
Switch Voltage (SWN)................................... 32V to –5V
Boosted Driver Voltage (BOOSTN – SWN) .... 7V to –0.3V
Peak Output Current 100ns
RPLL IN
PLLIN Input Resistance
IPLLFLTR
Phase Detector Output Current
Sinking Capability
Sourcing Capability
RRELPHS
fPLLIN < fOSC
fPLLIN > fOSC
Controller 2-Controller 1 Phase
Controller 3-Controller 1 Phase
1
V
50
kΩ
20
20
µA
µA
120
240
Deg
Deg
Differential Amplifier
AV
Differential Gain
VOS
Input Offset Voltage
CM
Common Mode Input Voltage Range
CMRR
Common Mode Rejection Ratio
0.995
IN+ = IN– = 1.2V, IOUT = 1mA,
Input Referred; Gain = 1
1.000
1.005
V/V
0.5
5
mV
0
0V < IN+ = IN– < 5V, I
OUT = 1mA, Input Referred
VCC
50
70
10
V
dB
ICL
Output Current
40
mA
GBP
Gain Bandwidth Product
IOUT = 1mA
2
MHz
SR
Slew Rate
RL = 2k
5
V/µs
VO(MAX)
Maximum High Output Voltage
IOUT = 1mA
RIN
Input Resistance
Measured at IN+ Pin
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired. A maximum current of 200µA is allowed to
pull-up the RUN/SS pin to prevent overcurrent shutdown.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
LTC3732CG: TJ = TA + (PD × 95°C/W)
LTC3732CUHF: TJ = TA + (PD × 34°C/W)
Note 3: The IC is tested in a feedback loop that includes the differential
amplifier in a unity-gain configuration loaded with 100µA to ground driving
the VID DAC into the error amplifier and servoing the resultant voltage to
the midrange point for the error amplifier (VITH = 1.2V).
VCC – 1.2 VCC –␣ 0.8
80
V
kΩ
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 5: The minimum on-time condition corresponds to an inductor peakto-peak ripple current of ≥ 40% of IMAX (see minimum on-time
considerations in the Applications Information Section).
Note 6: ATTENERR specification is in addition to the output voltage
accuracy specified at VID code = 11111.
Note 7: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
3732f
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LTC3732
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TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs IOUT (Figure 14)
95
80 VFCB = 5V
VFCB = 0V
70
90
60
50
40
10
VIN = 5V
IL = 50A
70
60
1
95
75
65
0.1
50
100
0
5
10
15
VIN (V)
20
595
90
Maximum ISENSE Threshold vs
Temperature
85
5.5
5.0
4.5
4.0
–45 –30 –15
110
0 15 30 45 60
TEMPERATURE (°C)
3732 G04
75
VO = 1.8V
75
VO = 0.6V
70
65
–45 –30 –15
90
0 15 30 45 60
TEMPERATURE (°C)
90
Undervoltage Reset Voltage vs
Temperature
700
VPLLFLTR = 2.4V
75
3732 G06
Oscillator Frequency vs VPLLFLTR
700
5
600
FREQUENCY (kHz)
500
400
VPLLFLTR = 1.2V
300
VPLLFLTR = 0V
100
UNDER VOLTAGE RESET (V)
VPLLFLTR = 5V
500
400
300
200
100
0
0
–45 –30 –15
80
3732 G05
Oscillator Frequency vs
Temperature
600
500
3732 G03
MAXIMUM ISENSE THRESHOLD (mV)
ERROR AMPLIFIER gm (mmho)
REFERENCE VOLTAGE (mV)
600
400
FREQUENCY (kHz)
6.0
605
300
3732 G02
610
FREQUENCY (kHz)
75
200
25
Error Amplifier gm vs
Temperature
10 30 50 70
TEMPERATURE (°C)
VIN = 8V
55
Reference Voltage vs
Temperature
200
VIN = 12V
85
VIN = 20V
3732 G01
590
–50 –30 –10
90
80
INDUCTOR CURRENT (A)
600
ILOAD = 20A
VOUT = 1.5V
IL = 20A
80
20
0
VOUT = 1.5V
f = 250kHz
85
30
VIN = 8V
VOUT = 1.5V
100
EFFICIENCY (%)
90 VFCB = OPEN
EFFICIENCY (%)
EFFICIENCY (%)
100
10
Efficiency vs Frequency (Figure 14)
Efficiency vs VIN (Figure 14)
100
0 15 30 45 60
TEMPERATURE (°C)
75
90
0
0.6
1.2
1.8
2.4
VPLLFLTR
3732 G07
3732 G08
4
3
2
1
0
–45 –30 –15
0 15 30 45 60
TEMPERATURE (°C)
75
90
3732 G09
3732f
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LTC3732
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TYPICAL PERFOR A CE CHARACTERISTICS
Short-Circuit Arming and Latchoff
vs Temperature
RUN/SS Pull-Up Current vs
Temperature
Supply Current vs Temperature
100
2.8
2.5
VCC = 5V
ARMING
SUPPLY CURRENT (mA)
RUN/SS PIN VOLTAGE (V)
LATCHOFF
3
2
2.0
60
1.6
40
1.2
0.8
20
1
SHUTDOWN CURRENT (µA)
80
2.4
4
0.4
0
–45 –30 –15
0 15 30 45 60
TEMPERATURE (°C)
75
90
0
–45 –30 –15
0
0 15 30 45 60
TEMPERATURE (°C)
3732 G10
75
0.5
50
40
30
20
0
50
25
0
6
75
90
Peak Current Threshold vs VITH
10
5
0 15 30 45 60
TEMPERATURE (°C)
75
ISENSE VOLTAGE THRESHOLD (mV)
ISENSE VOLTAGE (mV)
MAXIMUM ISENSE (mV)
60
3
4
VRUN/SS VOLTAGE (V)
1.0
3732 G12
70
2
1.5
0
–45 –30 –15
90
75
1
2.0
Maximum Current Sense
Threshold vs Duty Factor
80
0
VRUN/SS = 1.9V
3732 G11
Maximum ISENSE vs VRUN/SS
0
20
40
60
DUTY FACTOR (%)
80
60
45
30
15
0
–15
100
0
0.6
1.2
VITH (V)
1.8
2.4
3732 G14
3732 G13
3732 G13a
Maximum Duty Factor vs
Temperature
Percentage of Nominal Output vs
Peak ISENSE (Foldback)
80
ISENSE Pin Current vs VOUT
100
40
VPLLFLTR = 0V
60
50
40
30
20
30
98
ISENSE PIN CURRENT (µA)
MAXIMUM DUTY FACTOR (%)
70
PEAK ISENSE VOLTAGE (mV)
RUN/SS PULLUP CURRENT (mV)
5
96
94
20
10
0
–10
92
–20
10
0
0 10 20 30 40 50 60 70 80 90 100
PERCENTAGE OF NOMINAL OUTPUT VOLTAGE (%)
3732 G15
90
–45 –30 –15
0 15 30 45 60
TEMPERATURE (°C)
75
90
3732 G16
–30
0
1
3
2
VOUT (V)
4
5
3732 G17
3732f
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LTC3732
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TYPICAL PERFOR A CE CHARACTERISTICS
Differential Amplifier Gain-Phase
35
135
30
90
25
45
20
0
15
–45
10
–90
RL = 10k AC LOAD
5
1k
10k
100k
100
FREQUENCY (Hz)
0
0
–135
1M
GAIN (dB)
180
–3
–45
–6
–90
–9
–135
–12
–180
–15
0001
0.01
0.1
–225
1
10
FREQUENCY (MHz)
3732 G18
3732 G19
Shed Mode at 1Amp, Light Load
Current (Circuit of Figure 14)
Burst Mode at 1Amp, Light Load
Current (Circuit of Figure 14)
VOUT
AC, 20mV/DIV
VOUT
AC, 20mV/DIV
VSW1
10V/DIV
VSW1
10V/DIV
VSW2
10V/DIV
VSW3
10V/DIV
VSW2
10V/DIV
VSW3
10V/DIV
4µs/DIV
VIN = 12V
VOUT = 1.5V
VFCB = VCC
FREQUENCY = 250kHz
PHASE (DEG)
40
PHASE (DEG)
GAIN (dB)
Error Amplifier Gain-Phase
4µs/DIV
VIN = 12V
VOUT = 1.5V
VFCB = OPEN
FREQUENCY = 250kHz
3732 G20
Transient Load Current Response: 0Amp
to 50Amp (Circuit of Figure 14)
Continuous Mode at 1Amp, Light
Load Current (Circuit of Figure 14)
VOUT
AC, 20mV/DIV
3732 G21
VOUT
AC, 20mV/DIV
VSW1
10V/DIV
IL
20A/DIV
VSW2
10V/DIV
VSW3
10V/DIV
VIN = 12V
4µs/DIV
VOUT = 1.5V
VFCB = 0V
FREQUENCY = 250kHz
3732 G22
VIN = 12V
20µs/DIV
VOUT = 1.5V
VFCB = VCC
FREQUENCY = 250kHz
3732 G23
3732f
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LTC3732
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PI FU CTIO S
VID0 to VID4: Output Voltage Programming Input Pins. A
3µA internal pull-up current is provided on each input pin.
See Table 1 for details. Do not apply voltage to these pins
prior to the application of voltage on the VCC pin.
PLLIN: Synchronization Input to Phase Detector. This pin
is internally terminated to SGND with 50kΩ. The phaselocked loop will force the rising top gate signal of controller 1 to be synchronized with the rising edge of the PLLIN
signal.
PLLFLTR: The phase-locked loop’s lowpass filter is tied to
this pin. Alternatively, this pin can be driven with an AC or
DC voltage source to vary the frequency of the internal
oscillator. (Do not apply voltage directly to this pin prior to
the application of voltage on the VCC pin.)
FCB: Forced Continuous Control Input. The voltage applied to this pin sets the operating mode of the controller.
The forced continuous current mode is active when the
applied voltage is less than 0.6V. Burst Mode operation
will be active when the pin is allowed to float and a stage
shedding mode will be active if the pin is tied to the VCC pin.
(Do not apply voltage directly to this pin prior to the
application of voltage on the VCC pin.)
IN+, IN–: Inputs to a precision, unity-gain differential
amplifier with internal precision resistors. This provides
true remote sensing of both the positive and negative load
terminals for precise output voltage control.
DIFFOUT: Output of the Remote Output Voltage Sensing
Differential Amplifier.
EAIN: This is the input to the error amplifier which compares the VID divided, feedback voltage to the internal
0.6V reference voltage.
PADDLE (UHF Package Only): This pin is connected to
the heat spreading metal pad at the center of the package
bottom and is tied to the IC’s substrate. It must be
connected to the SGND pin.
SGND: Signal Ground. This pin must be routed separately
under the IC to the PGND pin and then to the main ground
plane.
SENSE1+, SENSE2+, SENSE3+, SENSE1–, SENSE2–,
SENSE3–: The Inputs to Each Differential Current Comparator. The ITH pin voltage and built-in offsets between
SENSE– and SENSE+ pins, in conjunction with RSENSE, set
the current trip threshold level.
RUN/SS: Combination of Soft-Start, Run Control Input
and Short-Circuit Detection Timer. A capacitor to ground
at this pin sets the ramp time to full current output as well
as the time delay prior to an output voltage short-circuit
shutdown. A minimum value of 0.01µF is recommended
on this pin.
ITH: Error Amplifier Output and Switching Regulator Compensation Point. All three current comparator’s thresholds
increase with this control voltage.
PGND: Driver Power Ground. This pin connects directly to
the sources of the bottom N-channel external MOSFETs
and the (–) terminals of CIN.
BG1 to BG3: High Current Gate Drives for Bottom NChannel MOSFETs. Voltage swing at these pins is from
ground to VCC.
VCC: Main Supply Pin. Because this pin supplies both the
controller circuit power as well as the high power pulses
supplied to drive the external MOSFET gates, this pin
needs to be very carefully and closely decoupled to the IC’s
PGND pin.
DRVCC (UHF Package Only): This pin provides power to
the bottom MOSFET on-chip drivers. Tie this pin to the VCC
pin and carefully decouple this pin to the PGND pin with a
minimum of 5µF of ceramic capacitance immediately
adjacent to the IC package.
SW1 to SW3: Switch Node Connections to Inductors.
Voltage swing at these pins is from a Schottky diode
(external) voltage drop below ground to VIN (where VIN is
the external MOSFET supply rail).
TG1 to TG3: High Current Gate Drives for Top N-channel
MOSFETs. These are the outputs of floating drivers with a
voltage swing equal to the boost voltage source superimposed on the switch node voltage SW.
BOOST1 to BOOST3: Positive Supply Pins to the Topside
Floating Drivers. Bootstrapped capacitors, charged with
external Schottky diodes and a boost voltage source, are
connected between the BOOST and SW pins. Voltage
swing at the BOOST pins is from boost source voltage
(typically VCC) to this boost source voltage + VIN (where
VIN is the external MOSFET supply rail).
PGOOD: This open-drain output is pulled low when the
output voltage has been outside the PGOOD tolerance
window for the VPGDLY delay of approximately 100µs.
3732f
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LTC3732
W
FU CTIO AL DIAGRA
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PLLIN
PHASE DET
FIN
50k
RLP PLLFLTR
CLK1
CLP
OSCILLATOR
CLK2
CLK3
FCB
DUPLICATE FOR SECOND AND THIRD
CONTROLLER CHANNELS
+
FCB
–
0.6V
PGOOD
EAIN
–
PROTECTION
IN–
DROP
OUT
DET
0.66V
+
100µs
DELAY
40k
+
0.54V
RS
LATCH
S
Q
R
Q
BOT
+
B
40k
–
+
–
BG
PGND
SHDN
+ +
–
–
+
I2
L
VCC
36k
3mV
SENSE +
SLOPE
COMP
R1
20k
VFB
+
+
RSENSE
–
36k SENSE
5(VFB)
COUT
+
–
0.660V
SW
BOT
FCB
–
I1
0.600V
CIN
SWITCH
LOGIC
0.55V
DIFFOUT
EAIN
+
FORCE BOT
VCC/DRVCC*
–
A1
+
40k
CB
TG
TOP
40k
IN+
DB
BOOST
–
VIN
VCC
EA
54k
SS
CLAMP
54k
VOUT
2.4V
OV
–
ITH
FCB
SHED
0.600V
VREF
VCC
CC
R2 VARIABLE
RC
1.5µA
SHDN
RST
RUN
SOFTSTART
5(VFB)
5-BIT VID DECODER
VCC
INTERNAL
SUPPLY
SGND
UV/ OVERTEMP
RESET
6V
VCC
+
CCC
RUN/SS
VID0 VID1 VID2 VID3 VID4
CSS
3732 F02
* UHF PACKAGE CONNECTION
Figure 2
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OPERATIO
(Refer to Functional Diagram)
Main Control Loop
The IC uses a constant frequency, current mode stepdown architecture. During normal operation, each top
MOSFET is turned on each cycle when the oscillator sets
the RS latch, and turned off when the main current
comparator, I1, resets each RS latch. The peak inductor
current at which I1 resets the RS latch is controlled by the
voltage on the ITH pin, which is the output of the error
amplifier EA. The EAIN pin receives a portion of this
voltage feedback signal via the DIFFOUT pin through the
internal VID DAC and is compared to the internal reference
3732f
9
LTC3732
U
OPERATIO
(Refer to Functional Diagram)
voltage. When the load current increases, it causes a slight
decrease in the EAIN pin voltage relative to the 0.6V
reference, which in turn causes the ITH voltage to increase
until each inductor’s average current matches one third of
the new load current (assuming all three current sensing
resistors are equal). In Burst Mode operation and stage
shedding mode, after each top MOSFET has turned off, the
bottom MOSFET is turned on until either the inductor
current starts to reverse, as indicated by current comparator I2, or the beginning of the next cycle.
The top MOSFET drivers are biased from floating bootstrap capacitor CB, which is normally recharged during
each off cycle, through an external Schottky diode. When
VIN decreases to a voltage close to VOUT, however, the loop
may enter dropout and attempt to turn on the top MOSFET
continuously. The dropout detector counts the number of
oscillator cycles that the bottom MOSFET remains off and
periodically forces a brief on period to allow CB to recharge.
The main control loop is shut down by pulling the RUN/SS
pin low. Releasing RUN/SS allows an internal 1.5µA
current source to charge soft-start capacitor CSS. When
CSS reaches 1.5V, the main control loop is enabled and the
internally buffered ITH voltage is clamped but allowed to
ramp as the voltage on CSS continues to ramp. This “softstart” clamping prevents abrupt current from being drawn
from the input power source. When the RUN/SS pin is low,
all functions are kept in a controlled state. The RUN/SS pin
is pulled low when the VCC input voltage is below 4V or
when the IC die temperature rises above 150°C.
Low Current Operation
The FCB pin is a multifunction pin: 1) an analog comparator input to provide regulation for a secondary winding by
forcing temporary forced PWM operation and 2) a logic
input to select between three modes of operation.
A) Burst Mode Operation
When the FCB pin voltage is below 0.6V, the controller
performs as a continuous, PWM current mode synchronous switching regulator. The top and bottom MOSFETs
are alternately turned on to maintain the output voltage
independent of direction of inductor current. When the
FCB pin is below VCC –␣ 1.5V but greater than 0.6V, the
controller performs as a Burst Mode switching regulator.
Burst Mode operation sets a minimum output current level
before turning off the top switch and turns off the synchronous MOSFET(s) when the inductor current goes negative. This combination of requirements will, at low current,
force the ITH pin below a voltage threshold that will
temporarily shut off both output MOSFETs until the output
voltage drops slightly. There is a burst comparator having
60mV of hysteresis tied to the ITH pin. This hysteresis
results in output signals to the MOSFETs that turn them on
for several cycles, followed by a variable “sleep” interval
depending upon the load current. The resultant output
voltage ripple is held to a very small value by having the
hysteretic comparator after the error amplifier gain block.
B) Stage Shedding Operation
When the FCB pin is tied to the VCC pin, Burst Mode
operation is disabled and the forced minimum inductor
current requirement is removed. This provides constant
frequency, discontinuous current operation over the widest possible output current range. At approximately 10%
of maximum designed load current, the second and third
output stages are shut off and the phase 1 controller alone
is active in discontinuous current mode. This “stage
shedding” optimizes efficiency by eliminating the gate
charging losses and switching losses of the other two
output stages. Additional cycles will be skipped when the
output load current drops below 1% of maximum designed load current in order to maintain the output voltage.
This stage shedding operation is not as efficient as Burst
Mode operation at very light loads, but does provide lower
noise, constant frequency operating mode down to very
light load conditions.
C) Continuous Current Operation
Tying the FCB pin to ground will force continuous current
operation. This is the least efficient operating mode, but
may be desirable in certain applications. The output can
source or sink current in this mode. When forcing continuous operation and sinking current, this current will be
forced back into the main power supply, potentially
boosting the input supply to dangerous voltage levels—
BEWARE!
3732f
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LTC3732
U
OPERATIO
(Refer to Functional Diagram)
Frequency Synchronization
Short-Circuit Detection
The phase-locked loop allows the internal oscillator to be
synchronized to an external source using the PLLIN pin.
The output of the phase detector at the PLLFLTR pin is also
the DC frequency control input of the oscillator which
operates over a 250kHz to 600kHz range corresponding to
a voltage input from 0V to 2.4V. When locked, the PLL
aligns the turn on of the top MOSFET to the rising edge of
the synchronizing signal. When no frequency information
is supplied to the PLLIN pin, PLLFLTR goes low, forcing
the oscillator to minimum frequency. A DC source can be
applied to the PLLFLTR pin to externally set the desired
operating frequency. An approximate 20µA discharge
current will be present at the pin with no PLLIN signal.
The RUN/SS capacitor is used initially to turn on and limit
the inrush current from the input power source. Once the
controllers have been given time, as determined by the
capacitor on the RUN/SS pin, to charge up the output
capacitors and provide full load current, the RUN/SS
capacitor is then used as a short-circuit timeout circuit. If
the output voltage falls to less than 70% of its nominal
output voltage, the RUN/SS capacitor begins discharging,
assuming that the output is in a severe overcurrent and/or
short-circuit condition. If the condition lasts for a long
enough period, as determined by the size of the RUN/SS
capacitor, the controller will be shut down until the RUN/SS
pin voltage is recycled. This built-in latchoff can be overridden by providing >5µA at a compliance of 4V to the
RUN/SS pin. This additional current shortens the softstart period but prevents net discharge of the RUN/SS
capacitor during a severe overcurrent and/or short-circuit
condition. Foldback current limiting is activated when the
output voltage falls below 70% of its nominal level whether
or not the short-circuit latchoff circuit is enabled. Foldback
current limit can be overridden by clamping the EAIN pin
such that the voltage is held above the (70%)(0.6V) or
0.42V level even when the actual output voltage is low.
Input capacitance ESR requirements and efficiency losses
are reduced substantially in a multiphase architecture
because the peak current drawn from the input capacitor
is effectively divided by the number of phases used and
power loss is proportional to the RMS current squared. A
3-stage, single output voltage implementation can reduce
input path power loss by 90%.
Differential Amplifier
This amplifier provides true differential output voltage
sensing. Sensing both VOUT+ and VOUT– benefits regulation in high current applications and/or applications having electrical interconnection losses. This sensing also
isolates the physical power ground from the physical
signal ground preventing the possibility of troublesome
“ground loops” on the PC layout and prevents voltage
errors caused by board-to-board interconnects, particularly helpful in VRM designs.
Power Good
The PGOOD pin is connected to the drain of an internal
N-channel MOSFET. The MOSFET is turned on once an
internal delay has elapsed and the output voltage has been
away from its nominal value by greater than 10%. If the
output returns to normal prior to the delay timeout, the
timer is reset. There is no delay time for the rising of the
PGOOD output once the output voltage is within the ±10%
“window.”
Input Undervoltage Reset
The RUN/SS capacitor will be reset if the input voltage,
(VCC) is allowed to fall below approximately 3.8V. The
capacitor on the RUN/SS pin will be discharged until the
short-circuit arming latch is disarmed. The RUN/SS capacitor will attempt to cycle through a normal soft-start
ramp up after the VCC supply rises above 3.8V. This circuit
prevents power supply latchoff in the event of input power
switching break-before-make situations. The PGOOD pin
is held low during startup until the RUN/SS capacitor rises
above the short-circuit latch-off arming threshold of approximately 3.8V.
The basic application circuit is shown in Figure 1 on the
first page of this data sheet. External component selection
is driven by the load requirement, and normally begins
with the selection of an inductance value based upon the
desired operating frequency, inductor current and output
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(Refer to Functional Diagram)
voltage ripple requirements. Once the inductors and
operating frequency have been chosen, the current sensing resistors can be calculated. Next, the power MOSFETs
and Schottky diodes are selected. Finally, C IN and COUT
are selected according to the required voltage ripple
requirements. The circuit shown in Figure 1 can be
configured for operation up to a MOSFET supply voltage
of 28V (limited by the external MOSFETs and possibly the
minimum on-time).
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Operating Frequency
The IC uses a constant frequency, phase-lockable architecture with the frequency determined by an internal
capacitor. This capacitor is charged by a fixed current plus
an additional current which is proportional to the voltage
applied to the PLLFLTR pin. Refer to the Phase-Locked
Loop and Frequency Synchronization section for additional information.
A graph for the voltage applied to the PLLFLTR pin versus
frequency is given in Figure 3. As the operating frequency
is increased the gate charge losses will be higher, reducing
efficiency (see Efficiency Considerations). The maximum
switching frequency is approximately 680kHz.
MOSFET gate charge and transition losses. In addition to
this basic tradeoff, the effect of inductor value on ripple
current and low current operation must also be considered. The PolyPhase approach reduces both input and
output ripple currents while optimizing individual output
stages to run at a lower fundamental frequency, enhancing
efficiency.
The inductor value has a direct effect on ripple current. The
inductor ripple current ∆IL per individual section, N,
decreases with higher inductance or frequency and increases with higher VIN or VOUT:
∆IL =
VOUT
fL
VOUT
1−
VIN
700
OPERATING FREQUENCY (kHz)
where f is the individual output stage operating frequency.
600
In a PolyPhase converter, the net ripple current seen by the
output capacitor is much smaller than the individual
inductor ripple currents due to the ripple cancellation. The
details on how to calculate the net output ripple current
can be found in Application Note 77.
500
400
300
200
0
0.5
1
1.5
2
PLLFLTR PIN VOLTAGE (V)
2.5
3731 F03
Figure 3. Operating Frequency vs VPLLFLTR
Inductor Value Calculation and Output Ripple Current
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
Figure 4 shows the net ripple current seen by the output
capacitors for the different phase configurations. The
output ripple current is plotted for a fixed output voltage as
the duty factor is varied between 10% and 90% on the
x-axis. The output ripple current is normalized against the
inductor ripple current at zero duty factor. The graph can
be used in place of tedious calculations. As shown in
Figure 4, the zero output ripple current is obtained when:
VOUT
k
=
where k = 1, 2, ..., N – 1
VIN
N
So the number of phases used can be selected to minimize
the output ripple current and therefore the output ripple
voltage at the given input and output voltages. In applica3732f
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tions having a highly varying input voltage, additional
phases will produce the best results.
inductor ripple current and consequent output voltage
ripple. Do not allow the core to saturate!
Accepting larger values of ∆IL allows the use of low
inductances but can result in higher output voltage ripple.
A reasonable starting point for setting ripple current is
∆IL = 0.4(IOUT)/N, where N is the number of channels and
IOUT is the total load current. Remember, the maximum
∆IL occurs at the maximum input voltage. The individual
inductor ripple currents are constant determined by the
inductor, input and output voltages.
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Because they lack a bobbin, mounting is more difficult.
However, designs for surface mount are available which
do not increase the height significantly.
1.0
0.8
0.7
IO(P-P)
VO/fL
Power MOSFET and D1, D2, D3 Selection
1-PHASE
2-PHASE
3-PHASE
4-PHASE
6-PHASE
0.9
0.6
0.5
0.4
0.3
0.2
0.1
0
0.1
0.2
0.3 0.4 0.5 0.6 0.7
DUTY FACTOR (VOUT/VIN)
0.8
0.9
3732 F04
Figure 4. Normalized Peak Output Current
vs Duty Factor [IRMS = 0.3(IO(P-P)]
Inductor Core Selection
Once the value for L1 to L3 is determined, the type of
inductor must be selected. High efficiency converters
generally cannot afford the core loss found in low cost
powdered iron cores, forcing the use of ferrite,
molypermalloy or Kool Mµ® cores. Actual core loss is
independent of core size for a fixed inductor value, but it
is very dependent on inductance selected. As inductance
increases, core losses go down. Unfortunately, increased
inductance requires more turns of wire and therefore
copper losses will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can
concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard,” which means that
inductance collapses abruptly when the peak design
current is exceeded. This results in an abrupt increase in
At least two external power MOSFETs must be selected for
each of the three output sections: One N-channel MOSFET
for the top (main) switch and one or more N-channel
MOSFET(s) for the bottom (synchronous) switch. The
number, type and “on” resistance of all MOSFETs selected
take into account the voltage step-down ratio as well as the
actual position (main or synchronous) in which the MOSFET
will be used. A much smaller and much lower input
capacitance MOSFET should be used for the top MOSFET
in applications that have an output voltage that is less than
1/3 of the input voltage. In applications where VIN >> VOUT,
the top MOSFETs’ “on” resistance is normally less important for overall efficiency than its input capacitance at
operating frequencies above 300kHz. MOSFET manufacturers have designed special purpose devices that provide
reasonably low “on” resistance with significantly reduced
input capacitance for the main switch application in switching regulators.
The peak-to-peak MOSFET gate drive levels are set by the
voltage, VCC, requiring the use of logic-level threshold
MOSFETs in most applications. Pay close attention to the
BVDSS specification for the MOSFETs as well; many of the
logic-level MOSFETs are limited to 30V or less.
Selection criteria for the power MOSFETs include the “on”
resistance RDS(ON), input capacitance, input voltage and
maximum output current.
MOSFET input capacitance is a combination of several
components but can be taken from the typical “gate
charge” curve included on most data sheets (Figure 5).
Kool Mµ is a registered trademark of Magnetics, Inc.
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The curve is generated by forcing a constant input current
into the gate of a common source, current source loaded
stage and then plotting the gate voltage versus time. The
initial slope is the effect of the gate-to-source and the gateto-drain capacitance. The flat portion of the curve is the
result of the Miller multiplication effect of the drain-to-gate
capacitance as the drain drops the voltage across the
current source load. The upper sloping line is due to the
drain-to-gate accumulation capacitance and the gate-tosource capacitance. The Miller charge (the increase in
coulombs on the horizontal axis from a to b while the curve
is flat) is specified for a given VDS drain voltage, but can be
adjusted for different VDS voltages by multiplying by the
ratio of the application VDS to the curve specified VDS
values. A way to estimate the CMILLER term is to take the
change in gate charge from points a and b on a manufacturers data sheet and divide by the stated VDS voltage
specified. CMILLER is the most important selection criteria
for determining the transition loss term in the top MOSFET
but is not directly specified on MOSFET data sheets. CRSS
and COS are specified sometimes but definitions of these
parameters are not included.
VIN
MILLER EFFECT
V
VGS
a
b
+
QIN
VGS
+V
DS
–
–
CMILLER = (QB – QA)/VDS
3732 F05
Figure 5. Gate Charge Characteristic
When the controller is operating in continuous mode the
duty cycles for the top and bottom MOSFETs are given by:
Main Switch Duty Cycle =
VOUT
VIN
V – VOUT
Synchronous Switch Duty Cycle = IN
VIN
The power dissipation for the main and synchronous
MOSFETs at maximum output current are given by:
2
I
V
PMAIN = OUT MAX 1 + δ RDS(ON) +
VIN N
2 IMAX
VIN
( )
( )(
)
()
RDR C MILLER •
2N
1
1
+
f
VCC – VTH(IL) VTH(IL)
2
PSYNC
I
V –V
= IN OUT MAX 1 + δ RDS(ON)
VIN
N
( )
where N is the number of output stages, δ is the temperature dependency of RDS(ON), RDR is the effective top driver
resistance (approximately 2Ω at VGS = VMILLER), VIN is the
drain potential and the change in drain potential in the
particular application. VTH(IL) is the data sheet specified
typical gate threshold voltage specified in the power
MOSFET data sheet at the specified drain current. CMILLER
is the calculated capacitance using the gate charge curve
from the MOSFET data sheet and the technique described
above.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which peak at the highest input voltage. For VIN < 12V, the
high current efficiency generally improves with larger
MOSFETs, while for VIN > 12V, the transition losses
rapidly increase to the point that the use of a higher
RDS(ON) device with lower CMILLER actually provides higher
efficiency. The synchronous MOSFET losses are greatest
at high input voltage when the top switch duty factor is low
or during a short circuit when the synchronous switch is
on close to 100% of the period.
The term (1 + δ ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
The Schottky diodes, D1 to D3 shown in Figure 1 conduct
during the dead time between the conduction of the two
large power MOSFETs. This prevents the body diode of the
bottom MOSFET from turning on, storing charge during
the dead time and requiring a reverse recovery period
which could cost as much as several percent in efficiency.
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CIN and COUT Selection
In continuous mode, the source current of each top
N-channel MOSFET is a square wave of duty cycle VOUT/VIN.
A low ESR input capacitor sized for the maximum RMS
current must be used. The details of a close form equation
can be found in Application Note 77. Figure 6 shows the
input capacitor ripple current for different phase configurations with the output voltage fixed and input voltage
varied. The input ripple current is normalized against the
DC output current. The graph can be used in place of
tedious calculations. The minimum input ripple current
can be achieved when the product of phase number and
output voltage, N(VOUT), is approximately equal to the
input voltage VIN or:
VOUT
k
where k = 1, 2, ..., N – 1
=
VIN
N
So the phase number can be chosen to minimize the input
capacitor size for the given input and output voltages.
In the graph of Figure 4, the local maximum input RMS
capacitor currents are reached when:
VOUT 2k – 1
=
where k = 1, 2, ..., N
VIN
N
These worst-case conditions are commonly used for design because even significant deviations do not offer much
relief. Note that capacitor manufacturer’s ripple current
ratings are often based on only 2000 hours of life. This
makes it advisable to further derate the capacitor or to
choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet
size or height requirements in the design. Always consult
the capacitor manufacturer if there is any question.
The Figure 6 graph shows that the peak RMS input current
is reduced linearly, inversely proportional to the number N
of stages used. It is important to note that the efficiency
loss is proportional to the input RMS current squared and
therefore a 3-stage implementation results in 90% less
power loss when compared to a single phase design.
Battery/input protection fuse resistance (if used), PC
board trace and connector resistance losses are also
reduced by the reduction of the input ripple current in a
PolyPhase system. The required amount of input capacitance is further reduced by the factor, N, due to the
effective increase in the frequency of the current pulses.
0.6
RMS INPUT RIPPLE CURRNET
DC LOAD CURRENT
A 2A to 8A Schottky is generally a good compromise for
both regions of operation due to the relatively small
average current. Larger diodes result in additional transition loss due to their larger junction capacitance.
0.5
1-PHASE
2-PHASE
3-PHASE
4-PHASE
6-PHASE
0.4
0.3
0.2
0.1
0
0.1
0.2
0.3 0.4 0.5 0.6 0.7
DUTY FACTOR (VOUT/VIN)
0.8
0.9
3732 F06
Figure 6. Normalized Input RMS Ripple Current
vs Duty Factor for One to Six Output Stages
Ceramic capacitors are becoming very popular for small
designs but several cautions should be observed. “X7R”,
“X5R” and “Y5V” are examples of a few of the ceramic
materials used as the dielectric layer, and these different
dielectrics have very different effect on the capacitance
value due to the voltage and temperature conditions
applied. Physically, if the capacitance value changes due
to applied voltage change, there is a concommitant piezo
effect which results in radiating sound! A load that draws
varying current at an audible rate may cause an attendant
varying input voltage on a ceramic capacitor, resulting in
an audible signal. A secondary issue relates to the energy
flowing back into a ceramic capacitor whose capacitance
value is being reduced by the increasing charge. The
voltage can increase at a considerably higher rate than the
constant current being supplied because the capacitance
value is decreasing as the voltage is increasing! Ceramic
capacitors, when properly selected and used however, can
provide the lowest overall loss due to their extremely low
ESR.
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The selection of COUT is driven by the required effective
series resistance (ESR). Typically once the ESR requirement is satisfied the capacitance is adequate for filtering.
The steady-state output ripple (∆VOUT) is determined by:
1
∆VOUT ≈ ∆IRIPPLE ESR +
8NfC OUT
where f = operating frequency of each stage, N is the
number of output stages, COUT = output capacitance and
∆IL = ripple current in each inductor. The output ripple is
highest at maximum input voltage since ∆IL increases
with input voltage. The output ripple will be less than 50mV
at max VIN with ∆IL = 0.4IOUT(MAX) assuming:
COUT required ESR < N • RSENSE
and
COUT > 1/(8Nf)(RSENSE)
The emergence of very low ESR capacitors in small,
surface mount packages makes very small physical implementations possible. The ability to externally compensate
the switching regulator loop using the ITH pin allows a
much wider selection of output capacitor types. The
impedance characteristics of each capacitor type is significantly different than an ideal capacitor and therefore
requires accurate modeling or bench evaluation during
design.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo and the Panasonic SP
surface mount types have a good (ESR)(size) product.
Once the ESR requirement for COUT has been met, the
RMS current rating generally far exceeds the IRIPPLE(P-P)
requirement. Ceramic capacitors from AVX, Taiyo Yuden,
Murata and Tokin offer high capacitance value and very
low ESR, especially applicable for low output voltage
applications.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in
surface mount configurations. New special polymer
surface mount capacitors offer very low ESR also but have
much lower capacitive density per unit volume. In the case
of tantalum, it is critical that the capacitors are surge tested
for use in switching power supplies. Several excellent
choices are the AVX TPS, AVX TPSV, the KEMET T510
series of surface-mount tantalums or the Panasonic SP
series of surface mount special polymer capacitors available in case heights ranging from 2mm to 4mm. Other
capacitor types include Sanyo POS-CAP, Sanyo OS-CON,
Nichicon PL series and Sprague 595D series. Consult the
manufacturer for other specific recommendations.
RSENSE Selection for Output Current
Once the frequency and inductor have been chosen,
RSENSE1, RSENSE2, RSENSE3 are determined based on the
required peak inductor current. The current comparator
has a maximum threshold of 75mV/RSENSE and an input
common mode range of SGND to (1.1) • VCC. The current
comparator threshold sets the peak inductor current,
yielding a maximum average output current IMAX equal to
the peak value less half the peak-to-peak ripple current,
∆IL.
Allowing a margin for variations in the IC and external
component values yields:
RSENSE = N
50mV
IMAX
The IC works well with values of RSENSE from 0.002Ω to
0.02Ω.
VCC Decoupling
The VCC pin supplies power not only to the internal circuits
of the controller but also to the top and bottom gate
drivers on the IC and therefore must be bypassed
very carefully to ground with a ceramic capacitor, type
X7R or X5R (depending upon the operating temperature
environment) of at least 1µF immediately next to the IC
and preferably an additional 10µF placed very close to
the IC due to the extremely high instantaneous currents
involved. The total capacitance, taking into account the
voltage coefficient of ceramic capacitors, should be
100 times as large as the total combined gate charge
capacitance of ALL of the MOSFETs being driven. Good
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bypassing close to the IC is necessary to supply the high
transient currents required by the MOSFET gate drivers
while keeping the 5V supply quiet enough so as not to
disturb the very small-signal high bandwidth of the current
comparators.
Topside MOSFET Driver Supply (CB, DB)
Each VID digital input is pulled up to a logical high with an
internal 3µA. The input logic threshold is approximately
1.2V but the input circuit can withstand an input voltage of
up to 7V.
Table 1. VID Output Voltage Programming
CODE
VOUT
CODE
VOUT
B4
B3 B2 B1 B0
B4
B3 B2 B1 B0
External bootstrap capacitors, CB, connected to the BOOST
pins, supply the gate drive voltages for the topside
MOSFETs. Capacitor CB in the Functional Diagram is
charged though diode DB from VCC when the SW pin is
low. When one of the topside MOSFETs turns on, the
driver places the CB voltage across the gate-source of the
desired MOSFET. This enhances the MOSFET and turns on
the topside switch. The switch node voltage, SW, rises to
VIN and the BOOST pin follows. With the topside MOSFET
on, the boost voltage is above the input supply (VBOOST =
VCC + VIN). The value of the boost capacitor CB needs to be
30 to 100 times that of the total input capacitance of the
topside MOSFET(s). The reverse breakdown of DB must be
greater than VIN(MAX).
1
0
0
0
0
1.450V
0
0
0
0
0
1.850V
1
0
0
0
1
1.425V
0
0
0
0
1
1.825V
1
0
0
1
0
1.400V
0
0
0
1
0
1.800V
1
0
0
1
1
1.375V
0
0
0
1
1
1.775V
1
0
1
0
0
1.350V
0
0
1
0
0
1.750V
1
0
1
0
1
1.325V
0
0
1
0
1
1.725V
1
0
1
1
0
1.300V
0
0
1
1
0
1.700V
1
0
1
1
1
1.275V
0
0
1
1
1
1.675V
1
1
0
0
0
1.250V
0
1
0
0
0
1.650V
1
1
0
0
1
1.225V
0
1
0
0
1
1.625V
1
1
0
1
0
1.200V
0
1
0
1
0
1.600V
1
1
0
1
1
1.175V
0
1
0
1
1
1.575V
1
1
1
0
0
1.150V
0
1
1
0
0
1.550V
1
1
1
0
1
1.125V
0
1
1
0
1
1.525V
Differential Amplifier
1
1
1
1
0
1.100V
0
1
1
1
0
1.500V
The IC has a true remote voltage sense capability. The
sensing connections should be returned from the load,
back to the differential amplifier’s inputs through a common, tightly coupled pair of PC traces. The differential
amplifier rejects common mode signals capacitively or
inductively radiated into the feedback PC traces as well as
ground loop disturbances. The differential amplifier output signal is divided down through the VID DAC and is
compared with the internal, precision 0.6V voltage reference by the error amplifier.
1
1
1
1
1
1.075V
0
1
1
1
1
1.475V
The differential amplifier has a 0 to VCC common mode
input range and an output swing range of 0 to VCC – 1.2V.
The output uses an NPN emitter follower without any
internal pull-down current. A DC resistive load to ground
is required in order to sink current.
Output Voltage
The IC includes a digitally controlled 5-bit attenuator
producing output voltages as defined in Table 1. Output
voltages with 25mV increments are produced from 1.075V
to 1.850V.
Soft-Start/Run Function
The RUN/SS pin provides three functions: 1) ON/OFF, 2)
soft-start and 3) a defeatable short-circuit latch off timer.
Soft-start reduces the input power sources’ surge currents by gradually increasing the controller’s current limit
(proportional to an internal buffered and clamped VITH).
The latchoff timer prevents very short, extreme load
transients from tripping the overcurrent latch. A small
pull-up current (>5µA) supplied to the RUN/SS pin will
prevent the overcurrent latch from operating. A maximum
pullup current of 200µA is allowed into the RUN/SS pin
even though the voltage at the pin may exceed the absolute
maximum rating for the pin. This is because the current is
limited and an internal protection circuit is provided. The
following explanation describes how this function operates.
An internal 1.5µA current source charges up the CSS
capacitor. When the voltage on RUN/SS reaches 1.5V, the
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controller is permitted to start operating. As the voltage on
RUN/SS increases from 1.5V to 3.5V, the internal current
limit is increased from 20mV/RSENSE to 75mV/RSENSE.
The output current limit ramps up slowly, taking an
additional 1s/µF to reach full current. The output current
thus ramps up slowly, eliminating the starting surge
current required from the input power supply. If RUN/SS
has been pulled all the way to ground, there is a delay
before starting of approximately:
tIRAMP
(
tLO2 >> (CSS • 3V)/(1.5µA) = 2 • 106 (CSS)
This built-in overcurrent latchoff can be overridden by
providing a pull-up resistor to the RUN/SS pin from VCC
as shown in Figure 7. When VCC is 5V, a 200k resistance
will prevent the discharge of the RUN/SS capacitor
during an overcurrent condition but also shortens the
soft-start period, so a larger RUN/SS capacitor value may
be required.
)
1.5V
C SS = 1s/µF C SS
1.5µA
3V − 1.5V
C SS = 1s/µF C SS
=
1.5µA
tDELAY =
additional time before latching off:
(
)
3.3V OR 5V
VCC
RUN/SS
PIN
5V
D1
SHDN
CSS
RUN/SS
PIN
RSS
SHDN
CSS
3732 F07
By pulling the RUN/SS pin below 0.4V the IC is put into low
current shutdown (IQ < 100 µA). The RUN/SS pin can be
driven directly from logic as shown in Figure7. Diode, D1,
in Figure 7 reduces the start delay but allows CSS to ramp
up slowly providing the soft-start function. The RUN/SS
pin has an internal 6V zener clamp (see the Functional
Diagram).
Fault Conditions: Overcurrent Latchoff
The RUN/SS pins also provide the ability to latch off the
controllers when an overcurrent condition is detected. The
RUN/SS capacitor is used initially to turn on and limit the
inrush current of all three output stages. After the controllers have been started and been given adequate time to
charge up the output capacitor and provide full load
current, the RUN/SS capacitor is used for a short-circuit
timer. If the output voltage falls to less than 70% of its
nominal value, the RUN/SS capacitor begins discharging
on the assumption that the output is in an overcurrent
condition. If the condition lasts for a long enough period,
as determined by the size of the RUN/SS capacitor, the
discharge current, and the circuit trip point, the controller
will be shut down until the RUN/SS pin voltage is recycled.
If the overload occurs during start-up, the time can be
approximated by:
tLO1 >> (CSS • 0.6V)/(1.5µA) = 4 • 105 (CSS)
If the overload occurs after start-up, the voltage on the
RUN/SS capacitor will continue charging and will provide
Figure 7. RUN/SS Pin Interfacing
Why should you defeat overcurrent latchoff? During the
prototyping stage of a design, there may be a problem with
noise pick-up or poor layout causing the protection circuit
to latch off the controller. Defeating this feature allows
troubleshooting of the circuit and PC layout. The internal
foldback current limiting still remains active, thereby
protecting the power supply system from failure. A decision can be made after the design is complete whether to
rely solely on foldback current limiting or to enable the
latchoff feature by removing the pull-up resistor.
The value of the soft-start capacitor CSS may need to be
scaled with output current, output capacitance and load
current characteristics. The minimum soft-start capacitance is given by:
CSS > (COUT )(VOUT) (10 –4) (RSENSE)
The minimum recommended soft-start capacitor of
CSS = 0.1µF will be sufficient for most applications.
Current Foldback
In certain applications, it may be desirable to defeat the
internal current foldback function. A negative impedance
is experienced when powering a switching regulator.
That is, the input current is higher at a lower VIN and
decreases as VIN is increased. Current foldback is designed to accommodate a normal, resistive load having
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increasing current draw with increasing voltage. The EAIN
pin should be artificially held 70% above its nominal
operating level of 0.6V, or 0.42V in order to prevent the IC
from “folding back” the peak current level. A suggested
circuit is shown in Figure 8.
VCC
VCC
LTC3732
The phase detector used is an edge sensitive digital type
that provides zero degrees phase shift between the external and internal oscillators. This type of phase detector will
not lock the internal oscillator to harmonics of the input
frequency. The PLL hold-in range, ∆fH, is equal to the
capture range, ∆fC:
∆fH = ∆fC = ±0.5 fO
Q1
CALCULATE FOR
0.42V TO 0.55V
approximately 400kHz. The nominal operating frequency
range of the IC is 225kHz to 680kHz.
EAIN
3732 F08
The output of the phase detector is a complementary pair
of current sources charging or discharging the external
filter components on the PLLFLTR pin. A simplified block
diagram is shown in Figure 9.
Figure 8. Foldback Current Elimination
The emitter of Q1 will hold up the EAIN pin to a voltage in
the absence of VOUT that will prevent the internal sensing
circuitry from reducing the peak output current. Removing the function in this manner eliminates the external
MOSFET’s protective feature under short-circuit conditions. This technique will also prevent the short-circuit
latchoff function from turning off the part during a shortcircuit event and the output current will only be limited to
N • 75mV/RSENSE.
Undervoltage Reset
In the event that the input power source to the IC (VCC)
drops below 4V, the RUN/SS capacitor will be discharged
to ground. When VCC rises above 4V, the RUN/SS capacitor will be allowed to recharge and initiate another softstart turn-on attempt. This may be useful in applications
that switch between two supplies that are not diode
connected, but note that this cannot make up for the
resultant interruption of the regulated output.
Phase-Locked Loop and Frequency Synchronization
The IC has a phase-locked loop comprised of an internal
voltage controlled oscillator and phase detector. This
allows the top MOSFET of output stage 1’s turn-on to be
locked to the rising edge of an external source. The
frequency range of the voltage controlled oscillator is
±50% around the center frequency fO. A voltage applied to
the PLLFLTR pin of 1.2V corresponds to a frequency of
PHASE
DETECTOR/
OSCILLATOR
EXTERNAL
OSC
RLP
10k
2.4V
CLP
OSC
PLLFLTR
PLLIN
50k
DIGITAL
PHASE/
FREQUENCY
DETECTOR
3732 F09
Figure 9. Phase-Locked Loop Block Diagram
If the external frequency (fPLLIN) is greater than the oscillator frequency, fOSC, current is sourced continuously,
pulling up the PLLFLTR pin. When the external frequency
is less than fOSC, current is sunk continuously, pulling
down the PLLFLTR pin. If the external and internal frequencies are the same, but exhibit a phase difference, the
current sources turn on for an amount of time corresponding to the phase difference. Thus, the voltage on the
PLLFLTR pin is adjusted until the phase and frequency of
the external and internal oscillators are identical. At this
stable operating point, the phase comparator output is
open and the filter capacitor CLP holds the voltage. The IC
PLLIN pin must be driven from a low impedance source
such as a logic gate located close to the pin. When using
multiple ICs for a phase-locked system, the PLLFLTR pin
of the master oscillator should be biased at a voltage that
will guarantee the slave oscillator(s) ability to lock onto the
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master’s frequency. A voltage of 1.7V or below applied to
the master oscillator’s PLLFLTR pin is recommended in
order to meet this requirement. The resultant operating
frequency will be approximately 550kHz for 1.7V.
The loop filter components (CLP, RLP) smooth out the
current pulses from the phase detector and provide a
stable input to the voltage controlled oscillator. The filter
components CLP and RLP determine how fast the loop
acquires lock. Typically RLP =10k and CLP ranges from
0.01µF to 0.1µF.
Minimum On-Time Considerations
Minimum on-time, tON(MIN), is the smallest time duration
that the IC is capable of turning on the top MOSFET. It is
determined by internal timing delays and the gate charge
of the top MOSFET. Low duty cycle applications may
approach this minimum on-time limit and care should be
taken to ensure that:
tON(MIN) <
VOUT
()
VIN f
If the duty cycle falls below what can be accommodated by
the minimum on-time, the IC will begin to skip every other
cycle, resulting in half-frequency operation. The output
voltage will continue to be regulated, but the ripple current
and ripple voltage will increase.
The minimum on-time for the IC is generally about 110ns.
However, as the peak sense voltage decreases the minimum on-time gradually increases. This is of particular
concern in forced continuous applications with low ripple
current at light loads. If the duty cycle drops below the
minimum on-time limit in this situation, a significant
amount of cycle skipping can occur with correspondingly
larger current and voltage ripple.
If an application can operate close to the minimum ontime limit, an inductor must be chosen that is low enough
in value to provide sufficient ripple amplitude to meet the
minimum on-time requirement. As a general rule, keep
the inductor ripple current equal to or greater than 30%
of IOUT(MAX) at VIN(MAX).
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, VOUT shifts by an
amount equal to ∆ILOAD • ESR, where ESR is the effective
series resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT, generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time, VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. The
availability of the ITH pin not only allows optimization of
control loop behavior, but also provides a DC coupled
and AC filtered closed-loop response test point. The DC
step, rise time and settling at this test point truly reflects
the closed-loop response. Assuming a predominantly
second order system, phase margin and/or damping
factor can be estimated using the percentage of overshoot
seen at this pin. The bandwidth can also be estimated by
examining the rise time at the pin. The ITH external components shown in the Figure 1 circuit will provide an
adequate starting point for most applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.2 to 5 times their suggested values) to maximize
transient response once the final PC layout is done and the
particular output capacitor type and value have been
determined. The output capacitors need to be decided
upon because the various types and values determine the
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loop feedback factor gain and phase. An output current
pulse of 20% to 80% of full load current having a rise time
of 1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If CLOAD is greater
than 2% of COUT , the switch rise time should be controlled
so that the load rise time is limited to approximately
1000 • RSENSE • CLOAD. Thus a 250µF capacitor and a 2mΩ
RSENSE resistor would require a 500µs rise time, limiting
the charging current to about 1A.
Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation. But before you connect, be advised: you are plugging
into the supply from hell. The main battery line in an
automobile is the source of a number of nasty potential
transients, including load dump, reverse battery and
double battery.
Load dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse battery is
just what it says, while double battery is a consequence of
tow-truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
The network shown in Figure 10 is the most straightforward approach to protect a DC/DC converter from the
ravages of an automotive battery line. The series diode
prevents current from flowing during reverse battery,
while the transient suppressor clamps the input voltage
during load dump. Note that the transient suppressor
should not conduct during double-battery operation, but
must still clamp the input voltage below breakdown of the
converter. Although the IC has a maximum input voltage
of 32V on the SW pins, most applications will be limited to
30V by the MOSFET BVDSS.
VBAT
12V
VCC
5V
+
LTC3732
3732 F10
Figure 10. Automotive Application Protection
Design Example
As a design example, assume VIN = 12V(nominal), VIN =
20V(max), VOUT = 1.3V, IMAX = 45A and f = 400kHz. The
inductance value is chosen first based upon a 30% ripple
current assumption. The highest value of ripple current in
each output stage occurs at the maximum input voltage.
L=
=
VOUT VOUT
1−
VIN
f ∆I
( )
(
1.3V
1−
20V
400kHz 30% 15A
≥ 0.68µH
1.3V
)( )( )
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Using L = 0.6µH, a commonly available value results in
34% ripple current. The worst-case output ripple for the
three stages operating in parallel will be less than 11% of
the peak output current.
RSENSE1, RSENSE2 and RSENSE3 can be calculated by using
a conservative maximum sense current threshold of 65mV
and taking into account half of the ripple current:
RSENSE =
65mV
= 0.0037Ω
34%
15A 1 +
2
Next verify the minimum on-time is not violated. The
minimum on-time occurs at maximum VCC:
VOUT
()
VIN(MAX) f
=
1.3V
(
20V 400kHz
)
= 162ns
The output voltage will be set by the VID code according
to Table 1.
The power dissipation on the topside MOSFET can be
estimated. Using a Fairchild FDS6688 for example, RDS(ON)
= 7mΩ, CMILLER = 15nC/15V = 1000pF. At maximum input
voltage with T(estimated) = 50°C:
PMAIN ≈
( )[ (
)(
)]
2
1.8V
15 1 + 0.005 50°C − 25°C
20V
2 45A
0.007Ω + 20
2Ω 1000pF
2 3
1
1
+
400kHz = 2.2W
5V – 1.8V 1.8V
( ) ( )( ) ( )(
(
)
)
The worst-case power dissipation by the synchronous
MOSFET under normal operating conditions at elevated
ambient temperature and estimated 50°C junction temperature rise is:
PSYNC =
22
( ) (1.25)(0.007Ω) = 1.84W
20V − 1.3V
15A
20V
2
ISC ≈
( ) = 7.5A
1 150ns 20V
+
2 + 3 mΩ 2 0.6µH
25mV
( )
with a typical value of RDS(ON) and d = (0.005/°C)(50°C) =
0.25. The resulting power dissipated in the bottom MOSFET
is:
PSYNC = (7.5A)2(1.25)(0.007Ω) ≈ 0.5W
Use a commonly available 0.003Ω sense resistor.
tON(MIN) =
A short circuit to ground will result in a folded back current
of:
which is less than one third of the normal, full load
conditions. Incidentally, since the load no longer dissipates any power, total system power is decreased by over
90%. Therefore, the system actually cools significantly
during a shorted condition!
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
IC. These items are also illustrated graphically in the layout
diagram of Figure 11. Check the following in the PC layout:
1) Are the signal and power ground paths isolated? Keep the
SGND at one end of a printed circuit path thus preventing
MOSFET currents from traveling under the IC. The IC signal
ground pin should be used to hook up all control circuitry
on one side of the IC, routing the copper through SGND,
under the IC covering the “shadow” of the package, connecting to the PGND pin and then continuing on to the (–) plates
of CIN and COUT. The VCC decoupling capacitor should be
placed immediately adjacent to the IC between the VCC pin
and PGND. A 1µF ceramic capacitor of the X7R or X5R type
is small enough to fit very close to the IC to minimize the ill
effects of the large current pulses drawn to drive the bottom
MOSFETs. An additional 5µF to 10uF of ceramic, tantalum
or other very low ESR capacitance is recommended in order to keep the internal IC supply quiet. The power ground
returns to the sources of the bottom N-channel MOSFETs,
anodes of the Schottky diodes and (–) plates of CIN, which
should have as short lead lengths as possible.
2) Does the IC IN+ pin connect to the (+) plates of COUT?
A 30pF to 300pF feedforward capacitor between the
DIFFOUT and EAIN pins should be placed as close as
possible to the IC.
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3) Are the SENSE– and SENSE+ printed circuit traces for
each channel routed together with minimum PC trace
spacing? The filter capacitors between SENSE+ and SENSE–
for each channel should be as close as possible to the pins
of the IC. Connect the SENSE– and SENSE+ pins to the
pads of the sense resistor as illustrated in Figure 12.
5) Keep the switching nodes, SWITCH, BOOST and TG
away from sensitive small-signal nodes (SENSE+, SENSE–
IN+, IN–, EAIN). Ideally the SWITCH, BOOST and TG
printed circuit traces should be routed away and separated
from the IC and the “quiet” side of the IC. Separate the high
dV/dt traces from sensitive small-signal nodes with ground
traces or ground planes.
4) Do the (+) plates of CPWR connect to the drains of the
topside MOSFETs as closely as possible? This capacitor
provides the pulsed current to the MOSFETs.
6) Use a low impedance source such as a logic gate to drive
the PLLIN pin and keep the lead as short as possible.
L1
SW1
RSENSE1
D1
L2
VIN
SW2
RIN
VOUT
RSENSE2
+
+
CIN
COUT
D2
BOLD LINES INDICATE HIGH,
SWITCHING CURRENT LINES.
KEEP LINES TO A MININMUM
LENGTH
RL
L3
SW3
RSENSE3
D3
3732 F11
Figure 11. Branch Current Waveforms
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Figure 11 illustrates all branch currents in a three-phase
switching regulator. It becomes very clear after studying
the current waveforms why it is critical to keep the high
switching current paths to a small physical size. High electric and magnetic fields will radiate from these “loops” just
as radio stations transmit signals. The output capacitor
ground should return to the negative terminal of the input
capacitor and not share a common ground path with any
switched current paths. The left half of the circuit gives rise
to the “noise” generated by a switching regulator. The
ground terminations of the synchronous MOSFETs and
Schottky diodes should return to the bottom plate(s) of the
input capacitor(s) with a short isolated PC trace since very
high switched currents are present. A separate isolated path
from the bottom plate(s) of the input and output capacitor(s)
should be used to tie in the IC power ground pin (PGND).
This technique keeps inherent signals generated by high
current pulses taking alternate current paths that have finite impedances during the total period of the switching
regulator. External OPTI-LOOP compensation allows overcompensation for PC layouts which are not optimized but
this is not the recommended design procedure.
reduced by, and the effective ripple frequency is increased
by the number of phases used. Figure 13 graphically
illustrates the principle.
SINGLE PHASE
SW V
ICIN
ICOUT
TRIPLE PHASE
SW1 V
SW2 V
SW3 V
IL1
IL2
IL3
ICIN
ICOUT
INDUCTOR
3732 F13
Figure 13. Single and Polyphase Current Waveforms
LTC3732
SENSE+
SENSE–
1000pF
SENSE
RESISTOR
3732 F12b
OUTPUT CAPACITOR
Figure 12. Kelvin Sensing RSENSE
Simplified Visual Explanation of How a 3-Phase
Controller Reduces Both Input and Output RMS
Ripple Current
The effect of multiphase power supply design significantly
reduces the amount of ripple current in both the input and
output capacitors. The RMS input ripple current is divided
by, and the effective ripple frequency is multiplied up by
the number of phases used (assuming that the input
voltage is greater than the number of phases used times
the output voltage). The output ripple amplitude is also
The worst-case input RMS ripple current for a single stage
design peaks at twice the value of the output voltage. The
worst-case input RMS ripple current for a two stage
design results in peaks at 1/4 and 3/4 of the input voltage,
and the worst-case input RMS ripple current for a three
stage design results in peaks at 1/6, 1/2, and 5/6 of the
input voltage. The peaks, however, are at ever decreasing
levels with the addition of more phases. A higher effective
duty factor results because the duty factors “add” as long
as the currents in each stage are balanced. Refer to AN19
for a detailed description of how to calculate RMS current
for the single stage switching regulator.
Figure 6 illustrates the RMS input current drawn from the
input capacitance versus the duty cycle as determined by
the ration of input and output voltage. The peak input RMS
current level of the single phase system is reduced by 2/3
in a 3-phase solution due to the current splitting between
the three stages.
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The output ripple current is reduced significantly when
compared to the single phase solution using the same
inductance value because the VOUT/L discharge currents
term from the stages that has their bottom MOSFETs on
subtract current from the (VCC – VOUT)/L charging current
resulting from the stage which has its top MOSFET on. The
output ripple current for a 3-phase design is:
IP-P =
(f)(L) (
VOUT
1 – 3DC
)
VIN > 3VOUT
The ripple frequency is also increased by three, further
reducing the required output capacitance when VCC < 3VOUT
as illustrated in Figure 6.
The addition of more phases by phase locking additional
controllers, always results in no net input or output ripple
at VOUT/VIN ratios equal to the number of stages
implemented. Designing a system with multiple stages
close to the VOUT/VIN ratio will significantly reduce the
ripple voltage at the input and outputs and thereby
improve efficiency, physical size and heat generation of
the overall switching power supply. Refer to Application
Note 77 for more information on Polyphase circuits.
Efficiency Calculation
To estimate efficiency, the DC loss terms include the input
and output capacitor ESR, each MOSFET RDS(ON), inductor resistance RL, the sense resistance RSENSE and the
forward drop of the Schottky rectifier at the operating
output current and temperature. Typical values for the
design example given previously in this data sheet are:
Main MOSFET RDS(ON) = 7mΩ (9mΩ at 90°C)
Sync MOSFET RDS(ON) = 7mΩ (9mΩ at 90°C)
CINESR = 20mΩ
COUTESR = 3mΩ
RL = 2.5mΩ
RSENSE = 3mΩ
VSCHOTTKY = 0.8V at 15A (0.7V at 90°C)
VOUT = 1.3V
VIN = 12V
IMAX = 45A
δ = 0.01%°C (MOSFET temperature coefficient)
N=3
f = 400kHz
The main MOSFET is on for the duty factor VOUT/VIN and
the synchronous MOSFET is on for the rest of the period
or simply (1 – VOUT/VIN). Assuming the ripple current is
small, the AC loss in the inductor can be made small if a
good quality inductor is chosen. The average current,
IOUT is used to simplify the calaculations. The equation
below is not exact but should provide a good technique
for the comparison of selected components and give a
result that is within 10% to 20% of the final application.
The temperature of the MOSFET’s die temperature may
require interative calculations if one is not familiar typical
performance. A maximum operating junction temperature of 90° to 100°C for the MOSFETs is recommended
for high reliability applications.
Common output path DC loss:
2
I
PCOMPATH ≈ N MAX RL + RSENSE
N
+ C OUTESR Loss
(
)
This totals 3.7W + COUTESR loss.
Total of all three main MOSFET’s DC loss:
2
V I
PMAIN = N OUT MAX 1 + δ RDS(ON)
VIN N
+ CINESR Loss
( )
This totals 0.66W + CINESR loss.
Total of all three synchronous MOSFET’s DC loss:
2
PSYNC
V I
= N 1 – OUT MAX 1 + δ RDS(ON)
VIN N
( )
This totals 5.4W.
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Total of all three main MOSFET’s AC loss:
The Schottky rectifier loss assuming 50ns nonoverlap
time:
45A
(2Ω)(1000pF )
(2)(3)
1
1
+
(400kHz) = 6.3 W
5V – 1.8V 1.8V
PMAIN ≈ 3(VIN )2
This totals 1W at VIN = 8V, 2.25W at VIN = 12V and 6.25W
at VIN = 20V.
Total of all three synchronous MOSFET’s AC loss:
(3)Q G
VIN
VDSSPEC
(f) = (6)(15nC )
VIN
VDSSPEC
(4E5)
This totals 0.08W at VIN = 8V, 0.12W at VIN = 12V and
0.19W at VIN = 20V. The bottom MOSFET does not
experience the Miller capacitance dissipation issue that
the main switch does because the bottom switch turns on
when its drain is close to ground.
2 • 3(0.7V)(15A)(50ns)(4E5)
This totals 1.26W.
The total output power is (1.3V)(45A) = 58.5W and the
total input power is approximately 70W so the % loss of
each component is as follows:
Main switch AC loss (VIN = 12V)
2.25W
3.75%
Main switch DC loss
0.66W
1.1%
Synchronous switch AC loss
0.19W
0.3%
Synchronous switch DC loss
5.4W
9%
Power path loss
3.7W
6.1%
The numbers above represent the values at VIN = 12V. It
can be seen from this simple example that two things can
be done to improve efficiency: 1) Use two MOSFETs on the
synchronous side and 2) use a smaller MOSFET for the
main switch with smaller CMILLER to better balance the AC
loss with the DC loss. A smaller, less expensive MOSFET
can actually perform better in the task of the main switch.
U
PACKAGE DESCRIPTIO
G Package
36-Lead Plastic SSOP (5.3mm)
(Reference LTC DWG # 05-08-1640)
12.50 – 13.10*
(.492 – .516)
1.25 ±0.12
7.8 – 8.2
36 35 34 33 32 31 30 29 28 27 26 25 24 23 22 21 20 19
5.3 – 5.7
7.40 – 8.20
(.291 – .323)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
0.42 ±0.03
0.65 BSC
1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18
RECOMMENDED SOLDER PAD LAYOUT
3. DRAWING NOT TO SCALE
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED .152mm (.006") PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE
0.09 – 0.25
(.0035 – .010)
5.00 – 5.60**
(.197 – .221)
2.0
(.079)
0° – 8°
0.55 – 0.95
(.022 – .037)
0.65
(.0256)
BSC
0.22 – 0.38
(.009 – .015)
0.05
(.002)
G36 SSOP 0802
3732f
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LTC3732
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PACKAGE DESCRIPTIO
UHF Package
38-Lead Plastic QFN (5mm × 7mm)
(Reference LTC DWG # 05-08-1701)
0.70 ± 0.05
5.50 ± 0.05
(2 SIDES)
4.10 ± 0.05
(2 SIDES)
3.20 ± 0.05
(2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
5.20 ± 0.05 (2 SIDES)
6.10 ± 0.05 (2 SIDES)
7.50 ± 0.05 (2 SIDES)
RECOMMENDED SOLDER PAD LAYOUT
5.00 ± 0.10
(2 SIDES)
3.15 ± 0.10
(2 SIDES)
0.75 ± 0.05
0.00 – 0.05
0.435 0.18
0.18
37 38
PIN 1
TOP MARK
(SEE NOTE 6)
1
0.23
2
5.15 ± 0.10
(2 SIDES)
7.00 ± 0.10
(2 SIDES)
0.40 ± 0.10
0.200 REF 0.25 ± 0.05
0.200 REF
0.00 – 0.05
0.75 ± 0.05
NOTE:
1. DRAWING CONFORMS TO JEDEC PACKAGE
OUTLINE M0-220 VARIATION WHKD
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
0.50 BSC
R = 0.115
TYP
(UH) QFN 0303
BOTTOM VIEW—EXPOSED PAD
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3732f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
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OPTIONAL FOR
SYNCHRONIZATION
1000pF
0.01µF
10k
VCC
100Ω, 1%
1
VID1 IN
SYNC IN
300kHz
VID1
2
PLLFLTR
4
5V
2N3904
BAT54
VRON
IN+
6
IN–
8
9
+
S1
0.01µF
S1–
60.4k
S2+
5V
11.8k
S2–
330pF
2.2k
1000pF
1000pF
BOOST1
TG1
FCB
5
7
27pF
PGOOD
PLLIN
3
14.7k
VID0
SW1
LTC3732
BOOST2
DIFFOUT
TG2
EAIN
SW2
SGND
VCC
10
SENSE1+
BG1
11
SENSE1–
PGND
12
SENSE2+
13
–
BG2
SENSE2
BG3
S3–
14
1000pF
S3+
SENSE3–
SW3
15
+
SENSE3
TG3
RUN/SS
BOOST3
16
3.3nF
17
VID2 IN
18
ITH
VID4
VID2
VID3
36
PGOOD
VID0 IN
47k
35
34
0.1µF
33
VCC
5V TO 7V
1Ω
VIN
M1
VOUT
L1
32
VCC
31
M2
D1
S1+
0.1µF
30
29
0.002Ω
S1–
10µF
6.3V
×3
+
10µF
25V
×5
+
COUT
VIN
28
M3
L2
27
1µF
26
0.002Ω
10µF
M4
D2
25
S2+
VIN
CIN 3.3V TO 20V
68µF
25V
S2–
24
23
VIN
22
M5
0.1µF
21
20
M6
VID4 IN
19
0.002Ω
D3
S3+
VID3 IN
3732 TA01
L3
VCC
S3–
VIN: 5V TO 20V
VOUT: 1.1V TO 1.85V, 65A
SWITCHING FREQUENCY: 300kHz
CIN: SANYO OS-CON 25SP68M
COUT: 270µF/2V ×6 PANASONIC SP EEFSDOD271R
L1 TO L3: 0.8µH SUMIDA CEP125-0R8
M1, M3, M5: IRF7811W ×2 OR FDS6688 ×2
M2, M4, M6: IRF7822 ×2 OR Si7892DP ×2
PIN #s SHOWN ARE FOR THE 36 PIN SSOP PACKAGE
Figure 14. VRM9.0/9.1 65A Power Supply for Pentium® 4 Processors
RELATED PARTS
PART NUMBER
LTC1530
LTC1628/LTC1628-PG/
LTC1628-SYNC
LTC1629/LTC1629-PG
LTC1702A
LTC1703
LTC1708-PG
LT®1709/
LT1709-8
LTC1735
LTC1736
LTC1778
LTC1929
LTC3711
LTC3729
LTC3730
LTC3731
LTC3778
DESCRIPTION
High Power Step-Down Synchronous DC/DC Controller
2-Phase, Dual Output Synchronous Step-Down
DC/DC Controllers
20A to 200A PolyPhase Synchronous Controllers
No RSENSETM 2-Phase Dual Sync Step-Down Controller
No RSENSE 2-Phase Dual Synchronous Step-Down
Controller with 5-Bit Mobile VID Control
2-Phase, Dual Synchronous Controller with Mobile VID
High Efficiency, 2-Phase Synchronous Step-Down
Switching Regulators with 5-Bit VID
High Efficiency Synchronous Step-Down Regulator
High Efficiency Synchronous Controller with VID Control
No RSENSE Current Mode Sync Step-Down Controller
2-Phase Synchronous Controllers
No RSENSE Current Mode Synchronous Step-Down
Controller with Digital 5-Bit Interface
20A to 200A, 550kHz PolyPhase Synchronous Controller
3-Phase, 5-Bit Intel Mobile VID
600kHz Synchronous Step-Down Controller
3-Phase 600kHz Synchronous Step-Down Controller
Optional RSENSE Current Mode Synchronous
Step-Down Controller
COMMENTS
High Efficiency 5V to 3.3V Conversion at Up to 15A, SO-8 Package
Reduces CIN and COUT, Power Good Output Signal, Synchronizable,
3.5V ≤ VIN ≤ 36V, IOUT up to 20A, 0.8V ≤ VOUT ≤ 5V
Expandable from 2-Phase to 12-Phase, No Heat Sink, VIN up to 36V
550kHz, No Sense Resistor
Mobile Pentium® III Processors, 550kHz,
VIN ≤ 7V
3.5V ≤ VIN ≤ 36V, VID Sets VOUT1, PGOOD
1.3V ≤ VOUT ≤ 3.5V, Current Mode Ensures
Accurate Current Sharing, 3.5V ≤ VIN ≤ 36V
Output Fault Protection, 16-Pin SSOP
Output Fault Protection, 24-Pin SSOP, 3.5V ≤ VIN ≤ 36V
≤97% Efficiency, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ (0.9) (VIN),
IOUT ≤ 20A
Up to 42A, No Heat Sinks, 3.5V ≤ VIN ≤ 36V
Up to 97% Efficiency, Ideal for Pentium III Processors,
0.925V ≤ VOUT ≤ 2V, 4V ≤ VIN ≤ 36V, IOUT up to 20A
Expandable from 2-Phase to 12-Phase, VIN up to 36V
0.6V ≤ VOUT ≤ 1.75V, IMVP3 Compatible
Up to 60A Output Current, Integrated MOSFET Drivers
0.6V ≤ VOUT ≤ 6V, 4.5V ≤ VIN ≤ 32V
IOUT up to 60A, Integrated MOSFET Drivers
4V ≤ VIN ≤ 36V, Adjustable Frequency up to 1.2MHz,
TSSOP-20 Package
No RSENSE is a trademark of Linear Technology Corporation. Pentium is a registered trademark of Intel Corporation.
3732f
28
Linear Technology Corporation
LT/TP 0503 1K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
LINEAR TECHNOLOGY CORPORATION 2002
工商网监
湘ICP备2023018690号
