LTC4006
4A, High Efficiency,
Standalone Li-Ion
Battery Charger
DESCRIPTIO
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FEATURES
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Complete Charger Controller for 2-, 3- or 4-Cell
Lithium-Ion Batteries
High Conversion Efficiency: Up to 96%
Output Currents Exceeding 4A
±0.8% Accurate Preset Voltages: 8.4V, 12.6V, 16.8V
Built-In Charge Termination with Automatic Restart
AC Adapter Current Limiting Maximizes Charge Rate*
Automatic Conditioning of Deeply Discharged
Batteries
Thermistor Input for Temperature Qualified Charging
Wide Input Voltage Range: 6V to 28V
0.5V Dropout Voltage; Maximum Duty Cycle: 98%
Programmable Charge Current: ±4% Accuracy
Indicator Outputs for Charging, C/10 Current
Detection and AC Adapter Present
Charging Current Monitor Output
16-Pin Narrow SSOP Package
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APPLICATIO S
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Notebook Computers
Portable Instruments
Battery-Backup Systems
Standalone Li-Ion Chargers
The LTC®4006 is a complete constant-current/constantvoltage charger controller for 2-, 3- or 4-cell lithium batteries in a small package using few external components.
The PWM controller is a synchronous, quasi-constant frequency, constant off-time architecture that will not generate audible noise even when using ceramic capacitors.
The LTC4006 is available in 8.4V, 12.6V and 16.8V versions
with ±0.8% voltage accuracy. Charging current is programmable with a single sense resistor to ±4% typical accuracy.
Charging current can be monitored as a representative
voltage at the IMON pin. A timer, programmed by an external
resistor, sets the total charge time or is reset to 25% of total
charge time after C/10 charging current is reached. Charging automatically resumes when the cell voltage falls below
3.9V/cell.
Fully discharged cells are automatically trickle charged at
10% of the programmed current until the cell voltage exceeds 2.5V/cell. Charging terminates if the low-battery
condition persists for more than 25% of the total charge
time.
The LTC4006 includes a thermistor sensor input that
suspends charging if an unsafe temperature condition is
detected and automatically resumes charging when the
battery temperature returns to within safe limits.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners. Protected by U.S. Patents
including 5723970.
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TYPICAL APPLICATIO
4A Li-Ion Battery Charger
DCIN
0V TO 28V
3A
INPUT SWITCH
0.1µF
5k
VLOGIC
DCIN
100k
CHG
CHG
INFET
LTC4006
CLP
ACP/SHDN
ACP
CHARGING
CURRENT MONITOR
32.4k
0.0047µF
THERMISTOR
10k
NTC
0.47µF
309k
TIMING
RESISTOR
(~2 HOURS)
6k
0.12µF
CLN
IMON
TGATE
NTC
BGATE
RT
PGND
ITH
CSP
GND
BAT
15nF
0.033Ω
TO SYSTEM LOAD
20µF
10µH
0.025Ω
BATTERY
20µF
4006 TA01
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LTC4006
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
Voltage from DCIN, CLP, CLN, TGATE, INFET,
ACP/SHDN, CHG to GND ....................... + 32V to – 0.3V
Voltage from CLP to CLN ..................................... ±0.3V
CSP, BAT to GND ................................... +28V to – 0.3V
RT to GND ................................................. +7V to – 0.3V
NTC ........................................................ +10V to – 0.3V
Operating Ambient Temperature Range
(Note 4) ............................................. – 40°C to 85°C
Operating Junction Temperature ......... – 40°C to 125°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
DCIN
1
16 INFET
CHG
2
15 BGATE
ACP/SHDN
3
14 PGND
RT
4
13 TGATE
GND
5
12 CLN
NTC
6
11 CLP
ITH
7
10 BAT
IMON
8
9
LTC4006EGN-2
LTC4006EGN-4
LTC4006EGN-6
GN PART MARKING
40062
40064
40066
CSP
GN PACKAGE
16-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 110°C/W
Order Options Tape and Reel: Add #TR
Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range (Note 4), otherwise specifications are at TA = 25°C. VDCIN = 20V, VBAT = 12V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
DCIN Operating Range
IDCIN
DCIN Operating Current
Sum of Current from CLP, CLN, DCIN
VTOL
Voltage Accuracy
(Note 2)
LTC4006-6
LTC4006-6
LTC4006-2
LTC4006-2
LTC4006-4
LTC4006-4
ITOL
TTOL
Current Accuracy (Note 3)
TYP
6
MAX
UNITS
28
V
3
5
mA
8.4
8.4
12.6
12.6
16.8
16.8
8.467
8.484
12.700
12.726
16.935
16.968
V
V
V
V
V
V
●
8.333
8.316
12.499
12.474
16.665
16.632
●
–4
–5
4
5
%
%
VBAT < 6V, VCSP – VBAT Target = 10mV
– 60
60
%
6V ≤ VBAT ≤ VLOBAT,
VCSP – VBAT Target = 10mV
– 40
40
%
–15
15
%
35
45
10
µA
µA
µA
5.5
V
VCSP – VBAT Target = 100mV
VBAT = 11.5V (LTC4006-2)
VBAT = 7.6V (LTC4006-6)
VBAT = 12V (LTC4006-4)
●
●
Termination Timer Accuracy
RRT = 270k
●
Battery Leakage Current
DCIN = 0V
DCIN = 0V
DCIN = 20V, VSHDN = 0V, VBAT = 12V
●
●
–10
20
25
0
DCIN Rising, VBAT = 0V
●
4.2
4.7
●
1
Shutdown
UVLO
Undervoltage Lockout Threshold
Shutdown Threshold at ACP/SHDN
DCIN Current in Shutdown
VSHDN = 0V, Sum of Current from CLP,
CLN, DCIN
2.5
2
3
V
mA
Current Sense Amplifier, CA1
Input Bias Current Into BAT Pin
µA
11.67
CMSL
CA1/I1 Input Common Mode Low
●
CMSH
CA1/I1 Input Common Mode High
●
0
V
VCLN – 0.2
V
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LTC4006
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range (Note 4), otherwise specifications are at TA = 25°C. VDCIN = 20V, VBAT = 12V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
140
165
200
mV
Current Comparators ICMP and IREV
ITMAX
Maximum Current Sense Threshold (VCSP – VBAT)
ITREV
Reverse Current Threshold (VCSP – VBAT)
VITH = 2.5V
●
– 30
mV
Current Sense Amplifier, CA2
Transconductance
1
mmho
Source Current
Measured at ITH, VITH = 1.4V
– 40
µA
Sink Current
Measured at ITH, VITH = 1.4V
40
µA
1.5
mmho
Current Limit Amplifier
Transconductance
VCLP
Current Limit Threshold
ICLP
CLP Input Bias Current
●
93
100
107
100
mV
nA
Voltage Error Amplifier, EA
Transconductance
Sink Current
OVSD
Measured at ITH, VITH = 1.4V
Overvoltage Shutdown Threshold as a Percent
of Programmed Charger Voltage
1
mmho
36
µA
●
102
107
110
%
●
0
0.17
0.25
V
25
50
Input P-Channel FET Driver (INFET)
DCIN Detection Threshold (VDCIN – VCLN)
DCIN Voltage Ramping Up
from VCLN – 0.1V
●
Forward Regulation Voltage (VDCIN – VCLN)
Reverse Voltage Turn-Off Voltage (VDCIN – VCLN)
DCIN Voltage Ramping Down
●
– 60
– 25
INFET “On” Clamping Voltage (VDCIN – VINFET)
IINFET = 1µA
●
5
5.8
INFET “Off” Clamping Voltage (VDCIN – VINFET)
IINFET = – 25µA
mV
mV
6.5
V
0.25
V
Thermistor
NTCVR
Reference Voltage During Sample Time
4.5
V
High Threshold
VNTC Rising
●
NTCVR
• 0.48
NTCVR
• 0.5
NTCVR
• 0.52
V
Low Threshold
VNTC Falling
●
NTCVR
• 0.115
NTCVR
• 0.125
NTCVR
• 0.135
V
Thermistor Disable Current
VNTC ≤ 10V
10
µA
Indicator Outputs (ACP/SHDN, CHG)
C10TOL
C/10 Indicator Accuracy
Voltage Falling at PROG
●
0.375
0.400
0.425
V
LBTOL
LOBAT Threshold Accuracy
LTC4006-6
LTC4006-2
LTC4006-4
●
●
●
4.70
7.27
9.70
4.93
7.5
10
5.14
7.71
10.28
V
V
V
RESTART Threshold Accuracy
LTC4006-6
LTC4006-2
LTC4006-4
●
●
●
7.5
11.35
15.15
7.8
11.7
15.6
7.96
11.94
15.92
V
V
V
VOL
Low Logic Level of ACP/SHDN, CHG
IOL = 100µA
●
0.5
V
VOH
High Logic Level of ACP/SHDN
IOH = –1µA
●
2.7
●
15
IPO
Pull-Up Current on ACP/SHDN
V = 0V
IC10
C/10 Indicator Sink Current from CHG
VOH = 3V
IOFF
Off State Leakage Current of CHG
VOH = 3V
Timer Defeat Threshold at CHG
V
µA
–10
–1
1
25
38
µA
1
µA
V
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LTC4006
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range (Note 4), otherwise specifications are at TA = 25°C. VDCIN = 20V, VBAT = 12V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
255
300
345
kHz
Oscillator
fOSC
Regulator Switching Frequency
fMIN
Regulator Switching Frequency in Drop Out
Duty Cycle ≥ 98%
20
25
kHz
DCMAX
Regulator Maximum Duty Cycle
VCSP = VBAT
98
99
%
Gate Drivers (TGATE, BGATE)
VTGATE High (VCLN – VTGATE)
ITGATE = –1mA
50
mV
VBGATE High
CLOAD = 3000pF
4.5
VTGATE Low (VCLN – VTGATE)
CLOAD = 3000pF
4.5
5.6
10
V
5.6
10
V
VBGATE Low
IBGATE = 1mA
50
mV
TGTR
TGTF
TGATE Transition Time
TGATE Rise Time
TGATE Fall Time
CLOAD = 3000pF, 10% to 90%
CLOAD = 3000pF, 10% to 90%
50
50
110
100
ns
ns
BGTR
BGTF
BGATE Transition Time
BGATE Rise Time
BGATE Fall Time
CLOAD = 3000pF, 10% to 90%
CLOAD = 3000pF, 10% to 90%
40
40
90
80
ns
ns
VTGATE at Shutdown (VCLN – VTGATE)
ITGATE = –1µA, DCIN = 0V, CLN = 12V
100
mV
VBGATE at Shutdown
IBGATE = 1µA, DCIN = 0V, CLN = 12V
100
mV
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: See Test Circuit
Note 3: Does not include tolerance of current sense resistor.
Note 4: The LTC4006E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
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TYPICAL PERFOR A CE CHARACTERISTICS
INFET Response Time to
Reverse Current
VOUT vs IOUT
Vgs OF PFET (2V/DIV)
–0.5
Id (REVERSE) OF
PFET (5A/DIV)
Id = 0A
1.25µs/DIV
TEST PERFORMED ON DEMOBOARD
LTC4006-2
VIN = 15VDC
CHARGER = ON
INFET = 1/2 Si4925DY
ICHARGE = UVLO
BAT
>2.5V/Cell
HIGH
Running
LOW
Input Current Limited Charging
>BAT
>2.5V/Cell
HIGH
Running
LOW
Charger Paused Due to Thermistor Out of Range
>BAT
X
Leakage
10% Programmed
Current
Programmed
Current
Programmed
Current
OFF
HIGH
Paused
Shut Down by ACP/SHDN Pin
Terminated by Low-Battery Fault (Note 1)
>BAT
>BAT
X
T/4 Stopped
Top-Off Charging. C/10 is Latched
>BAT
VFLOAT
OFF
HIGH
Timer is Reset by C/10 Comparator (Latched),
then Terminates After 1/4 T
>BAT
VFLOAT
OFF
HIGH
Terminated by Expired Timer
>BAT
VFLOAT**
OFF
HIGH
T/4 After C/10
Comparator Trip.
Stopped
>T Stopped
LOW or 25µA
(Faulted)
HIGH
HIGH
(Faulted)
25µA
X
X
X
X
X
HIGH
(Waiting
for Restart)
HIGH
(Waiting
for Restart)
Forced LOW
>BAT
and BAT
3.9V/Cell
*Open Drain. High when used with pull-up resistor.
**Most probable condition, X = Don’t care
Note 1: If a depleted battery is inserted while the charger is in this state,
the charger must be reset to initiate charging.
compares this current against the desired current programmed
by RIMON at the IMON pin and adjusts ITH until:
The accuracy of VIMON will range from 0% to ITOL.
therefore,
⎛ V
⎞ 3kΩ
ICHARGE = ⎜ REF – 11.67µA⎟ •
⎝ RIMON
⎠ RSENSE
The voltage at BAT is divided down by an internal resistor
divider and is used by error amp EA to decrease ITH if the
divider voltage is above the 1.19V reference. When the
charging current begins to decrease, the voltage at IMON
will decrease in direct proportion. The voltage at IMON is
then given by:
VIMON is plotted in Figure 2.
The amplifier CL1 monitors and limits the input current to
a preset level (100mV/RCL). At input current limit, CL1 will
decrease the ITH voltage, thereby reducing charging current. When this condition is detected, the C/10 indicator will
1.2
1.19V
1.0
0.8
VIMON (V)
VREF
V
–V
+ 11.67µA • 3kΩ
= CSP BAT
RIMON
3kΩ
0.6
0.4
0.309V
0.2
0
R
VIMON = (ICHARGE • RSENSE + 11.67µA • 3kΩ) • IMON
3kΩ
0
20
40
60
80
ICHARGE (% OF MAXIMUM CURRENT)
100
4006 F02
Figure 2. VIMON vs ICHARGE
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LTC4006
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OPERATIO
Table 2. Truth Table for LTC4006 Operation (Supplemental)
NUMBER
FROM
STATE
TO
STATE
NEXT C/10
LATCH
MAX BAT
CURRENT
ACP/SHDN
TIMER
STATE
1
Any
MSD
BAT
0
OFF
HIGH
Reset
HIGH
3
SD
Shut Down by Undervoltage Lockout
>BAT
and
BAT
BAT
BAT
BAT
>2.5V/Cell
Programmed
Current
HIGH
Running
12
CHARGE
CHARGE
Timer Defeated. (Low-Battery Conditioning Still
Functional)
>BAT
>2.5V/Cell
Programmed
Current
HIGH
Ignored
13
SD
CHARGE
Restart
>BAT
2.5V ≤ VBAT ≤ 3.9V
(V/Cell)
Programmed
Current
HIGH
Reset
14
CHARGE
CHARGE
Top-Off Charging
>BAT
>3.9V/Cell
Programmed
Current
HIGH
Running
LOW
15
CHARGE
CHARGE
C/10 Latch is SET when Battery Current is Less
Than 10% of Programmed Current
>BAT
>2.5V/Cell
Programmed
Current
HIGH
Reset
25µA
16
CHARGE
CHARGE
Top-Off Charging
>BAT
>3.9V/Cell
Programmed
Current
HIGH
Running
25µA
17
CHARGE
CHARGE
Input Current Limited Charging
>BAT
>2.5V/Cell
BAT
>2.5V/Cell
OFF
HIGH
Paused
LOW or 25µA
(Faulted)
19
CHARGE
SD
Shut Down by ACP/SHDN Pin
>BAT
>2.5V/Cell
0
OFF
Forced LOW
Reset
HIGH
20
CHARGE
SD
Terminated by Low-Battery Fault (Note 1)
>BAT
T/4 then Reset
HIGH
(Faulted)
21
CHARGE
SD
Terminates After 1/4 T
>BAT
VFLOAT
1
OFF
HIGH
>T/4 then Reset
HIGH
22
CHARGE
SD
Terminates After T
>BAT
VFLOAT**
0
OFF
HIGH
>T/4 then Reset
HIGH
MODE
DCIN
Shut Down by Low Adapter Voltage
SD
SD,
CONDITION,
CHARGE
4
BAT
VOLTAGE
Note 1: If a depleted battery is inserted while the charger is in this state,
the charger must be reset to initiate charging.
Note 2: See section on “Adapter Limiting”.
Note 3: The information contained in this table is supplemental to the
LTC4006 data sheet and has not been production qualified.
Note 4: Blank fields indicate no change, not considered, or other states
impact value.
*Open Drain. High when used with pull-up resistor.
** Most probable condition.
PRESENT C/10
LATCH
0
0
0
1
1
CHG*
LOW
Forced LOW
LTC4006: State Diagram (Supplemental)
1
MASTER
SHUTDOWN
ANY
2
SHUTDOWN
4
13
5,
6,
7
3, 8,
9, 10
3, 18, 19,
20, 21, 22
CONDITION
11
CHARGE
12, 14, 15, 16, 17
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LTC4006
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OPERATIO
be inhibited if it is not already active. If the charging current
decreases below 10% to 15% of programmed current,
while engaged in input current limiting, BGATE will be
forced low to prevent the charger from discharging the
battery. Audible noise can occur in this mode of operation.
LTC4006
R9
32.4k
–
6
RTH
10k
NTC
C7
0.47µF
CLK
NTC
S1
+
An overvoltage comparator guards against voltage transient overshoots (>7% of programmed value). In this
case, both MOSFETs are turned off until the overvoltage
condition is cleared. This feature is useful for batteries
which “load dump” themselves by opening their protection switch to perform functions such as calibration or
pulse mode charging.
60k
+
–
–
When the charger is enabled, it will not begin switching
until the ITH voltage exceeds a threshold that assures initial
current will be positive. This threshold is 5% to 15% of the
maximum programmed current. After the charger begins
switching, the various loops will control the current at a
level that is higher or lower than the initial current. The
duration of this transient condition depends upon the loop
compensation but is typically less than 100µs.
Thermistor Detection
The thermistor detection circuit is shown in Figure 3. It requires
an external resistor and capacitor in order to function properly.
The thermistor detector performs a sample-and-hold function. An internal clock, whose frequency is determined by
the timing resistor connected to RT, keeps switch S1
closed to sample the thermistor:
45k
+
As the voltage at BAT increases to near the input voltage
at DCIN, the converter will attempt to turn on the top
MOSFET continuously (“dropout’’). A watchdog timer
detects this condition and forces the top MOSFET to turn
off for about 300ns at 40µs intervals. This is done to
prevent audible noise when using ceramic capacitors at
the input and output.
Charger Startup
~4.5V
15k
D
TBAD
Q
C
4006 F03
Figure 3
This voltage is stored by C7. Then the switch is opened for a
short period of time to read the voltage across the thermistor.
tHOLD = 10 • RRT • 17.5pF = 54µs,
for RRT = 309k
When the tHOLD interval ends the result of the thermistor
testing is stored in the D flip-flop (DFF). If the voltage at
NTC is within the limits provided by the resistor divider
feeding the comparators, then the NOR gate output will be
low and the DFF will set TBAD to zero and charging will
continue. If the voltage at NTC is outside of the resistor
divider limits, then the DFF will set TBAD to one, the charger
will be shut down, and the timer will be suspended until
TBAD returns to zero (see Figure 4).
CLK
(NOT TO
SCALE)
tHOLD
tSAMPLE
tSAMPLE = 127.5 • 20 • RRT • 17.5pF = 13.8ms,
for RRT = 309k
The external RC network is driven to approximately 4.5V
and settles to a final value across the thermistor of:
VOLTAGE ACROSS THERMISTOR
COMPARATOR HIGH LIMIT
VNTC
COMPARATOR LOW LIMIT
4006 F04
VRTH(FINAL)
4.5V • RTH
=
RTH + R9
Figure 4
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LTC4006
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APPLICATIO S I FOR ATIO
Alternatively, a normally closed switch can be used to
detect when the battery is present (see Figure 8).
Charger Current Programming
The basic formula for charging current is:
200
180
160
Table 3. Recommended RSENSE Resistor Values
tTIMER (MINUTES)
ICHARGE(MAX)
100mV
=
RSENSE
140
120
IMAX (A)
RSENSE (Ω) 1%
RSENSE (W)
1.0
0.100
0.25
2.0
0.050
0.25
3.0
0.033
0.5
40
4.0
0.025
0.5
20
Setting the Timer Resistor
100
80
60
0
100 150 200 250 300 350 400 450 500
RRT (kΩ)
4006 F05
The charger termination timer is designed for a range of 1
hour to 3 hours with a ±15% uncertainty. The timer is
programmed by the resistor RRT using the following
equation:
tTIMER = 10 • 227 • RRT • 17.5pF (Refer to Figure 5)
(seconds)
It is important to keep the parasitic capacitance on the RT
pin to a minimum. The trace connecting RT to RRT should
be as short as possible.
Figure 5. tTIMER vs RRT
3.3V
LTC4006
VDD
µP
OUT
IN
200k
33k
CHG 2
4006 F06
Figure 6. Microprocessor Interface
CHG Status Output Pin
When the charge cycle starts, the CHG pin is pulled down
to ground by an internal N-channel MOSFET that can drive
more than 100µA. When the charge current drops to 10%
of the full-scale current (C/10), the N-channel MOSFET is
turned off and a weak 25µA current source to ground is
connected to the CHG pin. After a time out occurs, the pin
will go into a high impedance state. By using two different
value pull-up resistors, a microprocessor can detect three
states from this pin (charging, C/10 and stop charging).
See Figure 6.
Battery Detection
It is generally not good practice to connect a battery while
the charger is running. The timer is in an unknown state
and the charger could provide a large surge current into
the battery for a brief time. The circuit shown in Figure 7
keeps the charger shut down and the timer reset while a
battery is not connected.
LTC4006
ADAPTER
POWER
1 DCIN
470k
3 ACP/SHDN
4006 F07
SWITCH CLOSED IF
BATTERY CONNECTED
Figure 7
ADAPTER
POWER
LTC4006
1 DCIN
3 ACP/SHDN
4006 F08
SWITCH OPEN WHEN
BATTERY CONNECTED
Figure 8
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LTC4006
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APPLICATIO S I FOR ATIO
Soft-Start
The LTC4006 is soft started by the 0.12µF capacitor on the
ITH pin. On start-up, ITH pin voltage will rise quickly to 0.5V,
then ramp up at a rate set by the internal 40µA pull-up
current and the external capacitor. Battery charging
current starts ramping up when ITH voltage reaches 0.8V
and full current is achieved with ITH at 2V. With a 0.12µF
capacitor, time to reach full charge current is about 2ms
and it is assumed that input voltage to the charger will
reach full value in less than 2ms. The capacitor can be
increased up to 1µF if longer input start-up times are
needed.
Input and Output Capacitors
The input capacitor (C2) is assumed to absorb all input
switching ripple current in the converter, so it must have
adequate ripple current rating. Worst-case RMS ripple
current will be equal to one half of output charging current.
Actual capacitance value is not critical. Solid tantalum low
ESR capacitors have high ripple current rating in a relatively small surface mount package, but caution must be
used when tantalum capacitors are used for input or
output bypass. High input surge currents can be created
when the adapter is hot-plugged to the charger or when a
battery is connected to the charger. Solid tantalum capacitors have a known failure mechanism when subjected to
very high turn-on surge currents. Only Kemet T495 series
of “Surge Robust” low ESR tantalums are rated for high
surge conditions such as battery to ground.
The relatively high ESR of an aluminum electrolytic for C1,
located at the AC adapter input terminal, is helpful in
reducing ringing during the hot-plug event. Refer to Application Note 88 for more information.
Highest possible voltage rating on the capacitor will minimize problems. Consult with the manufacturer before use.
Alternatives include new high capacity ceramic (at least
20µF) from Tokin, United Chemi-Con/Marcon, et al. Other
alternative capacitors include OS-CON capacitors from
Sanyo.
The output capacitor (C3) is also assumed to absorb
output switching current ripple. The general formula for
capacitor current is:
IRMS
⎛
⎞
V
0.29(VBAT )⎜ 1 – BAT ⎟
⎝ VDCIN ⎠
=
(L1)( f)
For example:
VDCIN = 19V, VBAT = 12.6V, L1 = 10µH, and
f = 300kHz, IRMS = 0.41A.
EMI considerations usually make it desirable to minimize
ripple current in the battery leads, and beads or inductors
may be added to increase battery impedance at the 300kHz
switching frequency. Switching ripple current splits between the battery and the output capacitor depending on
the ESR of the output capacitor and the battery impedance.
If the ESR of C3 is 0.2Ω and the battery impedance is
raised to 4Ω with a bead or inductor, only 5% of the
current ripple will flow in the battery.
Inductor Selection
Higher operating frequencies allow the use of smaller
inductor and capacitor values. A higher frequency generally results in lower efficiency because of MOSFET gate
charge losses. In addition, the effect of inductor value on
ripple current and low current operation must also be
considered. The inductor ripple current ∆IL decreases
with higher frequency and increases with higher VIN.
∆IL =
⎛ V ⎞
1
VOUT ⎜ 1– OUT ⎟
( f)(L) ⎝ VIN ⎠
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.4(IMAX). In no case should
∆IL exceed 0.6(IMAX) due to limits imposed by IREV and
CA1. Remember the maximum ∆IL occurs at the maximum input voltage. In practice 10µH is the lowest value
recommended for use.
Lower charger currents generally call for larger inductor
values. Use Table 4 as a guide for selecting the correct
inductor value for your application.
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Table 4
MAXIMUM
AVERAGE CURRENT (A)
INPUT
VOLTAGE (V)
MINIMUM INDUCTOR
VALUE (µH)
1
≤ 20
40 ±20%
1
> 20
56 ±20%
2
≤ 20
20 ±20%
2
> 20
30 ±20%
3
≤ 20
15 ±20%
3
> 20
20 ±20%
4
≤ 20
10 ±20%
4
> 20
15 ±20%
Charger Switching Power MOSFET
and Diode Selection
Two external power MOSFETs must be selected for use
with the charger: a P-channel MOSFET for the top (main)
switch and an N-channel MOSFET for the bottom (synchronous) switch.
The peak-to-peak gate drive levels are set internally. This
voltage is typically 6V. Consequently, logic-level threshold
MOSFETs must be used. Pay close attention to the BVDSS
specification for the MOSFETs as well; many of the logic
level MOSFETs are limited to 30V or less.
Selection criteria for the power MOSFETs include the “ON”
resistance RDS(ON), total gate capacitance QG, reverse
transfer capacitance CRSS, input voltage and maximum
output current. The charger is operating in continuous
mode at moderate to high currents so the duty cycles for
the top and bottom MOSFETs are given by:
highest at high input voltages. For VIN < 20V the high
current efficiency generally improves with larger MOSFETs,
while for VIN > 20V the transition losses rapidly increase
to the point that the use of a higher RDS(ON) device with
lower CRSS actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage or during a short circuit when the duty cycle in this
switch is nearly 100%. The term (1 + δ∆T) is generally
given for a MOSFET in the form of a normalized RDS(ON) vs
temperature curve, but δ = 0.005/°C can be used as an
approximation for low voltage MOSFETs. CRSS is usually
specified in the MOSFET characteristics; if not, then CRSS
can be calculated using CRSS = QGD/∆VDS. The constant
k = 2 can be used to estimate the contributions of the two
terms in the main switch dissipation equation.
If the charger is to operate in low dropout mode or with a
high duty cycle greater than 85%, then the topside
P-channel efficiency generally improves with a larger
MOSFET. Using asymmetrical MOSFETs may achieve cost
savings or efficiency gains.
The Schottky diode D1, shown in the Typical Application
on the back page, conducts during the dead-time between
the conduction of the two power MOSFETs. This prevents
the body diode of the bottom MOSFET from turning on and
storing charge during the dead-time, which could cost as
much as 1% in efficiency. A 1A Schottky is generally a
good size for 4A regulators due to the relatively small
average current. Larger diodes can result in additional
transition losses due to their larger junction capacitance.
Main Switch Duty Cycle = VOUT/VIN
The diode may be omitted if the efficiency loss can be
tolerated.
Synchronous Switch Duty Cycle = (VIN – VOUT)/VIN.
Calculating IC Power Dissipation
The MOSFET power dissipations at maximum output
current are given by:
PMAIN = VOUT/VIN(I2MAX)(1 + δ∆T)RDS(ON)
+ k(V2IN)(IMAX)(CRSS)(fOSC)
PSYNC = (VIN – VOUT)/VIN(I2MAX)(1 + δ∆T)RDS(ON)
Where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
Both MOSFETs have I2R losses while the PMAIN equation
includes an additional term for transition losses, which are
The power dissipation of the LTC4006 is dependent upon
the gate charge of the top and bottom MOSFETs (QG1 and
QG2 respectively). The gate charge is determined from the
manufacturer’s data sheet and is dependent upon both the
gate voltage swing and the drain voltage swing of the
MOSFET. Use 6V for the gate voltage swing and VDCIN for
the drain voltage swing.
PD = VDCIN • (fOSC (QG1 + QG2) + IDCIN)
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Example:
VDCIN = 19V, fOSC = 345kHz, QG1 = QG2 = 15nC.
PD = 292mW
IDCIN = 5mA
between the CLP and DCIN pins. When this voltage exceeds 100mV, the amplifier will override programmed
charging current to limit adapter current to 100mV/RCL. A
lowpass filter formed by 5kΩ and 15nF is required to
eliminate switching noise. If the current limit is not used,
CLP should be connected to CLN.
Adapter Limiting
An important feature of the LTC4006 is the ability to
automatically adjust charging current to a level which
avoids overloading the wall adapter. This allows the product to operate at the same time that batteries are being
charged without complex load management algorithms.
Additionally, batteries will automatically be charged at the
maximum possible rate of which the adapter is capable.
This feature is created by sensing total adapter output
current and adjusting charging current downward if a
preset adapter current limit is exceeded. True analog
control is used, with closed-loop feedback ensuring that
adapter load current remains within limits. Amplifier CL1
in Figure 9 senses the voltage across RCL, connected
Setting Input Current Limit
To set the input current limit, you need to know the
minimum wall adapter current rating. Subtract 7% for the
input current limit tolerance and use that current to determine the resistor value.
RCL = 100mV/ILIM
ILIM = Adapter Min Current –
(Adapter Min Current • 7%)
As is often the case, the wall adapter will usually have at
least a +10% current limit margin and many times one can
simply set the adapter current limit value to the actual
adapter rating (see Figure 9).
Designing the Thermistor Network
LTC4006
100mV
–
+
CLP
11
15nF
CL1
5k
+
RCL*
CLN
12
*RCL =
100mV
ADAPTER CURRENT LIMIT
+
CIN
AC ADAPTER
INPUT
VIN
TO SYSTEM
LOAD
4006 F09
Figure 9. Adapter Current Limiting
There are several networks that will yield the desired
function of voltage vs temperature needed for proper
operation of the thermistor. The simplest of these is the
voltage divider shown in Figure 10. Unfortunately, since
the HIGH/LOW comparator thresholds are fixed internally,
there is only one thermistor type that can be used in this
network; the thermistor must have a HIGH/LOW resistance ratio of 1:7. If this happy circumstance is true for
Table 5. Common RCL Resistor Values
ADAPTER
RATING (A)
–7% ADAPTER
RATING (A)
RCL VALUE*
(Ω) 1%
RCL
LIMIT (A)
RCL POWER
DISSIPATION (W)
RCL POWER
RATING (W)
1.5
1.40
0.068
1.47
0.15
0.25
1.8
1.67
0.062
1.61
0.16
0.25
2.0
1.86
0.051
1.96
0.20
0.25
2.3
2.14
0.047
2.13
0.21
0.25
2.5
2.33
0.043
2.33
0.23
0.50
2.7
2.51
0.039
2.56
0.26
0.50
3.0
2.79
0.036
2.79
0.28
0.50
3.3
3.07
0.033
3.07
0.31
0.50
3.6
3.35
0.030
3.35
0.33
0.50
4.0
3.72
0.027
3.72
0.37
0.50
* Rounded to nearest 5% standard step value. Many non-standard values are popular.
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LTC4006
LTC4006
R9
NTC 6
R9
NTC 6
C7
RTH
C7
R9A
4006 F10
4006 F11
Figure 10. Voltage Divider Thermistor Network
you, then simply set R9 = RTH(LOW)
If you are using a thermistor that doesn’t have a 1:7 HIGH/
LOW ratio, or you wish to set the HIGH/LOW limits to
different temperatures, then the more generic network in
Figure 11 should work.
Once the thermistor, RTH, has been selected and the
thermistor value is known at the temperature limits, then
resistors R9 and R9A are given by:
For NTC thermistors:
R9 = 6 RTH(LOW) • RTH(HIGH)/(RTH(LOW) – RTH(HIGH))
R9A = 6 RTH(LOW) • RTH(HIGH)/(RTH(LOW) – 7 • RTH(HIGH))
where RTH(LOW) > 7 • RTH(HIGH)
For PTC thermistors:
R9 = 6 RTH(LOW) • RTH(HIGH)/(RTH(HIGH) – RTH(LOW))
R9A = 6 RTH(LOW) • RTH(HIGH)/(RTH(HIGH) – 7 • RTH(LOW))
where RTH(HIGH) > 7RTH(LOW)
Example #1: 10kΩ NTC with custom limits
TLOW = 0°C, THIGH = 50°C
RTH = 10k at 25°C,
RTH(LOW) = 32.582k at 0°C
RTH(HIGH) = 3.635k at 50°C
R9 = 24.55k → 24.3k (nearest 1% value)
R9A = 99.6k → 100k (nearest 1% value)
Example #2: 100kΩ NTC
TLOW = 5°C, THIGH = 50°C
RTH = 100k at 25°C,
RTH(LOW) = 272.05k at 5°C
RTH(HIGH) = 33.195k at 50°C
R9 = 226.9k → 226k (nearest 1% value)
R9A = 1.365M → 1.37M (nearest 1% value)
Example #3: 22kΩ PTC
TLOW = 0°C, THIGH = 50°C
RTH
Figure 11. General Thermistor Network
RTH = 22k at 25°C,
RTH(LOW) = 6.53k at 0°C
RTH(HIGH) = 61.4k at 50°C
R9 = 43.9k → 44.2k (nearest 1% value)
R9A = 154k
Sizing the Thermistor Hold Capacitor
During the hold interval, C7 must hold the voltage across
the thermistor relatively constant to avoid false readings.
A reasonable amount of ripple on NTC during the hold
interval is about 10mV to 15mV. Therefore, the value of C7
is given by:
C7 = t HOLD /(R9/7 • –ln(1 – 8 • 15mV/4.5V))
= 10 • RRT • 17.5pF/(R9/7 • – ln(1 – 8 • 15mV/4.5V)
Example:
R9 = 24.3k
RRT = 309k (~2 hour timer)
C7 = 0.57µF → 0.56µF (nearest value)
Disabling the Thermistor Function
If the thermistor is not needed, connecting a resistor
between DCIN and NTC will disable it. The resistor should
be sized to provide at least 10µA with the minimum voltage
applied to DCIN and 10V at NTC. Do not exceed 30µA into
NTC. Generally, a 301k resistor will work for DCIN less
than 15V. A 499k resistor is recommended for DCIN
between 15V and 24V.
Optional Simple Battery Discharge Path Circuit
It is NOT recommended that one permit battery current to
flow backwards through RSENSE, inductor and out the
TGATE MOSFET internal diode to reach VOUT. The TGATE
MOSFET is off when VIN < VBAT. Figure 12 shows an optional high efficiency discharge path for the battery such that
VOUT power comes from lossless “diode or” of VIN and VBAT.
Normally when VIN > VBAT, P-channel MOSFET Q1B VGS =
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selection. If the VIN supply is going to collapse very slowly
such that Q1B is not turned on quickly enough for the given
load and stay within its PD limits, you should install a suitable Schottky diode in parallel with Q1B.
100k
VIN
Q1A
PCB Layout Considerations
VOUT
ZENER
18V
TGATE
Q1B
INDUCTOR
RSENSE
VBAT
4006 F12
Figure 12. Optional Simple High
Efficiency Battery Discharge Path
0V keeping Q1B in the off state while P-channel MOSFET
Q1A is on. If VIN were to suddenly go away, Q1B internal
diode will provide a passive but instant discharge path for
battery current to reach VOUT and hold up the load. Q1B
internal diode has the same current rating as the FET itself,
but has a very high Vf of about a volt such that heat will
quickly build up in Q1B if left alone. However as VIN’s voltage
falls below VBAT by Q1B’s VGS threshold, Q1B will then turn
on shorting out its internal diode removing both the heat
and voltage losses created by the diode. When VIN falls to
zero volts, Q1B gate will be driven to the same voltage as
VBAT providing the lowest possible RDSON value. A zener
diode along with a 100k resistor in series with the Q1B gate
protects the gate from any hazardous voltage spikes that
can exceed Q1B maximum permissible VGS voltage. The
zener voltage rating must be less than Q1B VGS(MAX) voltage but greater than VBAT.
Since Q1A and Q1B are always at opposite states and share
the same load, it is often advantagous to combine both FETs
into a single package and save PCB space. The PD rate of
the FET that is on is enhanced when the other FET is off. The
choice of a combined Q1 should take into account the highest load current conditions of both paths and choose
whichever is greater as the driving force behind the MOSFET
For maximum efficiency, the switch node rise and fall times
should be minimized. To prevent magnetic and electrical
field radiation and high frequency resonant problems,
proper layout of the components connected to the IC is
essential. (See Figure 13.) Here is a PCB layout priority list
for proper layout. Layout the PCB using this specific order.
1. Input capacitors need to be placed as close as possible
to switching FET’s supply and ground connections.
Shortest copper trace connections possible. These
parts must be on the same layer of copper. Vias must
not be used to make this connection.
2. The control IC needs to be close to the switching FET’s
gate terminals. Keep the gate drive signals short for a
clean FET drive. This includes IC supply pins that connect to the switching FET source pins. The IC can be
placed on the opposite side of the PCB relative to above.
3. Place inductor input as close as possible to switching
FET’s output connection. Minimize the surface area of
this trace. Make the trace width the minimum amount
needed to support current—no copper fills or pours.
Avoid running the connection using multiple layers in
parallel. Minimize capacitance from this node to any
other trace or plane.
SWITCH NODE
L1
VBAT
VIN
C2
HIGH
FREQUENCY
CIRCULATING
PATH
D1
C3
BAT
4006 F13
Figure 13. High Speed Switching Path
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4. Place the output current sense resistor right next to
the inductor output but oriented such that the IC’s
current sense feedback traces going to resistor are not
long. The feedback traces need to be routed together
as a single pair on the same layer at any given time with
smallest trace spacing possible. Locate any filter
component on these traces next to the IC and not at the
sense resistor location.
8. Route analog ground as a trace tied back to IC ground
(analog ground pin if present) before connecting to
any other ground. Avoid using the system ground
plane. CAD trick: make analog ground a separate
ground net and use a 0Ω resistor to tie analog ground
to system ground.
5. Place output capacitors next to the sense resistor
output and ground.
10. If possible, place all the parts listed above on the same
PCB layer.
6. Output capacitor ground connections need to feed
into same copper that connects to the input capacitor
ground before tying back into system ground.
11. Copper fills or pours are good for all power connections except as noted above in Rule 3. You can also use
copper planes on multiple layers in parallel too—this
helps with thermal management and lower trace inductance improving EMI performance further.
General Rules
7. Connection of switching ground to system ground or
internal ground plane should be single point. If the
system has an internal system ground plane, a good
way to do this is to cluster vias into a single star point
to make the connection.
9. A good rule of thumb for via count for a given high
current path is to use 0.5A per via. Be consistent.
12. For best current programming accuracy provide a
Kelvin connection from RSENSE to CSP and BAT. See
Figure 13 as an example.
It is important to keep the parasitic capacitance on the RT,
CSP and BAT pins to a minimum. The traces connecting
these pins to their respective resistors should be as short
as possible.
DIRECTION OF CHARGING CURRENT
RSNS
4006 F14
CSP
BAT
Figure 14. Kelvin Sensing of Charging Current
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LTC4006
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PACKAGE DESCRIPTION
GN Package
16-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.189 – .196*
(4.801 – 4.978)
.045 ± .005
16 15 14 13 12 11 10 9
.254 MIN
.009
(0.229)
REF
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 ± .0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
RECOMMENDED SOLDER PAD LAYOUT
1
.015 ± .004
× 45°
(0.38 ± 0.10)
.007 – .0098
(0.178 – 0.249)
2 3
4
5 6
7
.0532 – .0688
(1.35 – 1.75)
8
.004 – .0098
(0.102 – 0.249)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN16 (SSOP) 0204
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
4006fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LTC4006
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TYPICAL APPLICATIO
2A Li-Ion Battery Charger
Q3
INPUT SWITCH
DCIN
0V TO 20V
2.5A
C1
0.1µF
VLOGIC
1
R3
100k
2
CHG
3
ACP
CHARGING
CURRENT MONITOR
8
R9 32.4k
6
4
THERMISTOR
10k
NTC
C5
0.0047µF
C7
0.47µF
RT
309k
TIMING
RESISTOR
(~2 HOURS)
7
R4
6.04k
C6
0.12µF
5
16
INFET
LTC4006
11
CHG
CLP
12
CLN
ACP/SHDN
13
TGATE
IMON
15
BGATE
NTC
14
PGND
RT
9
CSP
ITH
10
BAT
GND
DCIN
D1: MBRM140T3
Q1, Q2: Si7501DN
Q3: Si5435B
R1
5k
C4
15nF
RCL
0.04Ω
TO SYSTEM LOAD
C2
20µF
Q1
L1
22µH 2A
RSENSE
0.05Ω
BATTERY
Q2
D1
C3
20µF
4006 TA02
RELATED PARTS
PART NUMBER
LT1511
DESCRIPTION
3A Constant-Current/Constant-Voltage Battery Charger
LT1513
SEPIC Constant- or Programmable-Current/ConstantVoltage Battery Charger
2-Phase, Dual Synchronous Step-Down Controller
with VID
Dual Battery Charger/Selector
LTC1709
LTC1760/
LTC1960
LTC1778
LTC3711
LTC3728
LTC4002
LTC4007
LTC4008
Wide Operating Range, No RSENSETM Synchronous
Step-Down Controller
No RSENSE Synchronous Step-Down Controller
with VID
2-Phase, Dual Synchronous Step-Down Controller
Li-Ion Battery Charger Controller
High Efficiency, Programmable Voltage,
Battery Charger with Termination
High Efficiency, Programmable Voltage/Current
Battery Charger
LTC4100
Smart Battery Charger Controller
LTC4412
PowerPathTM Ideal Diode or Controller
COMMENTS
High Efficiency, Minimum External Components to Fast Charge Lithium,
NIMH and NiCd Batteries
Charger Input Voltage May be Higher, Equal to or Lower Than Battery Voltage,
500kHz Switching Frequency
Up to 42A Output, Minimum CIN and COUT, Uses Smallest Components for
Intel and AMD Processors
Simultaneous Charge or Discharge of Two Batteries, DAC Programmable
Current and Voltage, Input Current Limiting Maximizes Charge Current
2% to 90% Duty Cycle at 200kHz, Stable with Ceramic COUT
3.5V ≤ VIN ≤ 36V, 0.925V ≤ VOUT ≤ 2V, for Transmeta, AMD and Intel
Mobile Processors
Minimizes CIN and COUT, Power Good Output, 3.5V ≤ VIN ≤ 36V
1- and 2-Cell Li-Ion Batteries, VIN ≤ 22V, 500kHz Switching Frequency,
3hr Charge Termination, IOUT ≤ 4A
Complete Charger for 3- or 4-Cell Li-Ion Batteries, AC Adapter
Current Limit, Thermistor Sensor and Indicator Outputs
Constant-Current/Constant-Voltage Switching Regulator, Resistor Voltage/
Current Programming, AC Adapter Current Limit and Thermistor Sensor and
Indicator Outputs
SMBus (Rev 1.1) Compliant, 6.4V ≤ VIN ≤ 26V, SMBus Accelerator Minimizes
Bus Errors
Very Low Loss Replacement for OR’ing Diodes
No RSENSE and PowerPath are trademarks of Linear Technology Corporation.
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20 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
LT 0506 REV A • PRINTED IN USA
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