LTC4008
4A, High Efficiency,
Multi-Chemistry Battery Charger
FEATURES
DESCRIPTION
General Purpose Charger Controller
High Conversion Efficiency: Up to 96%
Output Currents Exceeding 4A
±0.8% Voltage Accuracy
AC Adapter Current Limiting Maximizes
Charge Rate*
Thermistor Input for Temperature Qualified Charging
Wide Input Voltage Range: 6V to 28V
Wide Output Voltage: 3V to 28V
0.5V Dropout Voltage; Maximum Duty Cycle: 98%
Programmable Charge Current: ±4% Accuracy
Indicator Outputs for Charging, C/10 Current
Detection, AC Adapter Present, Input Current
Limiting and Faults
n Charging Current Monitor Output
n Available in a 20-Pin Narrow SSOP Package
The LTC®4008 is a constant-current/constant-voltage
charger controller. The PWM controller uses a synchronous, quasi-constant frequency, constant off-time architecture that will not generate audible noise even when
using ceramic capacitors. Charging current is programmable with a sense resistor and programming resistor to
±4% typical accuracy. Charging current can be monitored
as a voltage across the programming resistor. An external
resistor divider and precision internal reference set the
final float voltage.
n
n
n
n
n
n
n
n
n
n
n
APPLICATIONS
Notebook Computers
Portable Instruments
n Battery Backup Systems
n
n
All registered trademarks and trademarks are the property of their respective owners. Protected
by U.S. Patents including 5723970.
TYPICAL APPLICATION
12.3V, 4A Li-Ion Charger
INPUT SWITCH
DCIN
0V TO 28V
0.1µF
140k*
VLOGIC
100k
100k
ICL
ACP
LTC4008
DCIN
BATMON
VFB
INFET
ICL
CLP
ACP/SHDN
FAULT
TGATE
FLAG
FLAG
BGATE
NTC
PGND
32.4k
RT
15k*
150k
0.47µF
ITH
6.04k
GND
0.12µF
0.1µF
0.02Ω
SYSTEM
LOAD
4.99k
20µF
CLN
FAULT
THERMISTOR
10k
NTC
The LTC4008 includes a thermistor sensor input that will
suspend charging if an unsafe temperature condition is
detected and will automatically resume charging when
battery temperature returns to within safe limits; a FAULT
pin indicates this condition. A FLAG pin indicates when
charging current has decreased below 10% of the programmed current. An external sense resistor programs
AC adapter current limiting. The ICL pin indicates when the
charging current is being reduced by input current limiting
so that the charging algorithm can adapt.
Q1
10µH
Q2
0.025Ω
20µF
Li-Ion
BATTERY
CSP
BAT
3.01k
3.01k
PROG
0.0047µF
26.7k
NOTE: * 0.25% TOLERANCE
ALL OTHER RESISTORS ARE 1% TOLERANCE
Q1: Si4431BDY
Q2: FDC645N
CHARGING
CURRENT
MONITOR
4008 TA01
Rev C
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1
LTC4008
ABSOLUTE MAXIMUM RATINGS (Note 1)
Voltage from DCIN, CLP, CLN to GND.......... +32V/–0.3 V
PGND with Respect to GND....................................±0.3V
CSP, BAT to GND........................................... +28V/–0.3V
VFB, RT to GND................................................ +7V/–0.3V
NTC................................................................+10V/–0.3V
ACP/SHDN, FLAG, FAULT, ICL ....................... +32V/–0.3V
CLP to CLN ...........................................................+0.5V
Operating Ambient Temperature Range
(Note 4)................................................–40°C to 85°C
Operating Junction Temperature............. –40°C to 125°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................... 300°C
PIN CONFIGURATION
LTC4008
LTC4008-1*
TOP VIEW
TOP VIEW
DCIN
1
20 INFET
DCIN
1
20 NC
ICL
2
19 BGATE
ICL
2
19 BGATE
ACP/SHDN
3
18 PGND
SHDN
3
18 PGND
RT
4
17 TGATE
RT
4
17 TGATE
FAULT
5
16 CLP
FAULT
5
16 CLP
GND
6
15 CLN
GND
6
15 CLN
VFB
7
14 FLAG
VFB
7
14 FLAG
NTC
8
13 BATMON
NTC
8
13 BATMON
ITH
9
12 BAT
ITH
9
12 BAT
PROG 10
11 CSP
PROG 10
11 CSP
GN PACKAGE
20-LEAD NARROW PLASTIC SSOP
TJMAX = 125°C, θJA = 90°C/W
GN PACKAGE
20-LEAD NARROW PLASTIC SSOP
TJMAX = 125°C, θJA = 90°C/W
*The LTC4008EGN-1 does not have the Input FET function
ORDER INFORMATION
http://www.linear.com/product/LTC4008#orderinfo
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC4008EGN#PBF
LTC4008EGN#TRPBF
LTC4008EGN
20-Lead Narrow Plastic SSOP
–40°C to 125°C
LTC4008EGN-1#PBF
LTC4008EGN-1#TRPBF
LTC4008EGN-1
20-Lead Narrow Plastic SSOP
–40°C to 125°C
Consult ADI Marketing for parts specified with wider operating temperature ranges.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
Rev C
2
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LTC4008
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VDCIN = 20V, VBAT = 12V unless otherwise noted.
SYMBOL PARAMETER
CONDITIONS
MIN
DCIN Operating Range
IQ
Operating Current
Charging Sum of Current from CLP, CLN, DCIN
VTOL
Voltage Accuracy
(Notes 2, 5)
3
l
ITOL
TYP
6
BATMON Error (Note 5)
Measured from BAT to BATMON,
RLOAD = 100k
Charge Current Accuracy (Note 3)
VCSP – VBAT Target = 100mV
l
–4
–5
UNITS
28
V
5
mA
0.8
1.0
%
%
80
mV
4
5
%
%
20
35
10
µA
µA
–0.8
–1.0
0
MAX
35
Shutdown
UVLO
Battery Leakage Current
DCIN = 0V (LTC4008 Only)
ACP/SHDN = 0V
l
l
–10
Undervoltage Lockout Threshold
DCIN Rising, VBAT = 0V
l
4.2
4.7
5.5
V
l
1
1.6
2.5
V
2
3
Shutdown Threshold at ACP/SHDN
VSHDN = 0V, Sum of Current from CLP,
CLN, DCIN
Operating Current in Shutdown
mA
Current Sense Amplifier, CA1
Input Bias Current Into BAT Pin
CMSL
CA1/I1 Input Common Mode Low
CMSH
CA1/I1 Input Common Mode High
11.66
l
VDCIN ≤ 28V
Input Voltage Offset
VOS
Current Comparators ICMP and IREV
ITMAX
Maximum Current Sense Threshold (VCSP – VBAT) VITH = 2.5V
ITREV
0
V
VCLN – 0.2
l
–3.5
l
µA
140
Reverse Current Threshold (VCSP – VBAT)
165
V
3.5
mV
200
mV
–30
mV
Current Sense Amplifier, CA2
Transconductance
1
mmho
Source Current
Measured at ITH, VITH = 1.4V
–40
µA
Sink Current
Measured at ITH, VITH = 1.4V
40
µA
1.4
mmho
Current Limit Amplifier
Transconductance
VCLP
Current Limit Threshold
ICLN
CLN Input Bias Current
l
93
100
107
100
mV
nA
Voltage Error Amplifier, EA
Transconductance
1
VREF
Reference Voltage Used to Calculate VFLOAT
IBEA
Input Bias Current
OVSD
Overvoltage Shutdown Threshold as a Percent of
Programmed Charger Voltage
mmho
1.19
±4
Measured at ITH, VITH = 1.4V
Sink Current
V
±25
36
l
102
107
nA
µA
110
%
Rev C
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3
LTC4008
ELECTRICAL
CHARACTERISTICS
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VDCIN = 20V, VBAT = 12V unless otherwise noted.
SYMBOL PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
0
0.17
0.25
V
25
50
Input P-Channel FET Driver (INFET) (LTC4008 Only)
DCIN Detection Threshold (VDCIN – VCLP)
DCIN Voltage Ramping Up
from VCLP – 0.1V
Forward Regulation Voltage (VDCIN – VCLP)
l
l
Reverse Voltage Turn-Off Voltage (VDCIN – VCLP)
DCIN Voltage Ramping Down
l
–60
–25
INFET “On” Clamping Voltage (VCLP – VINFET)
IINFET = 1µA
l
5
5.8
INFET “Off” Clamping Voltage (VCLP – VINFET)
IINFET = –25µA
mV
mV
6.5
V
0.25
V
Thermistor
NTCVR
Reference Voltage During Sample Time
4.5
V
High Threshold
VNTC Rising
l
NTCVR
• 0.48
NTCVR
• 0.5
NTCVR
• 0.52
V
Low Threshold
VNTC Falling
l
NTCVR
• 0.115
NTCVR
• 0.125
NTCVR
• 0.135
V
Thermistor Disable Current
VNTC ≤ 10V
10
µA
V
Indicator Outputs (ACP/SHDN, FLAG, ICL, FAULT
C10TOL
FLAG (C/10) Accuracy
Voltage Falling at PROG
ICL Threshold Accuracy
VCLP – VCLN
VOL
Low Logic Level of ACP/SHDN, FLAG, ICL, FAULT IOL = 100µA
VOH
High Logic Level of ACP/SHDN, ICL
IOH = –1µA
IOFF
Off State Leakage Current of FLAG, FAULT
VOH = 3V
IPO
Pull-Up Current on ACP/SHDN, ICL
V = 0V
l
l
0.375
0.397
0.420
83
93
105
mV
0.5
V
2.7
V
–1
1
–10
µA
µA
Oscillator
fOSC
Regulator Switching Frequency
fMIN
Regulator Switching Frequency in Drop Out
DCMAX
Regulator Maximum Duty Cycle
255
300
345
kHz
Duty Cycle ≥ 98%
20
25
kHz
VCSP = VBAT
98
99
%
Gate Drivers (TGATE, BGATE)
VTGATE High (VCLP – VTGATE)
ITGATE = –1mA
50
mV
VBGATE High
CLOAD = 3000pF
5.6
10
V
VTGATE Low (VCLP – VTGATE)
CLOAD = 3000pF
5.6
10
V
VBGATE Low
IBGATE = 1mA
50
mV
TGTR
TGTF
TGATE Transition Time
TGATE Rise Time
TGATE Fall Time
CLOAD = 3000pF, 10% to 90%
CLOAD = 3000pF, 10% to 90%
50
50
110
100
ns
ns
BGTR
BGTF
BGATE Transition Time
BGATE Rise Time
BGATE Fall Time
CLOAD = 3000pF, 10% to 90%
CLOAD = 3000pF, 10% to 90%
40
40
90
80
ns
ns
VTGATE at Shutdown (VCLP – VTGATE)
ITGATE = –1µA, DCIN = 0V, CLP = 12V
100
mV
VBGATE at Shutdown
IBGATE = 1µA, DCIN = 0V, CLP = 12V
100
mV
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: See “Test Circuit”.
Note 3: Does not include tolerance of current sense resistor or current
programming resistor.
Note 4: The LTC4008E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 5: Voltage accuracy includes BATMON error and voltage reference
error. Does not include error of external resistor divider.
Rev C
4
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LTC4008
TYPICAL PERFORMANCE CHARACTERISTICS (TA = 25°C unless otherwise noted)
INFET Response Time to
Reverse Current
VFB vs DCIN
0.05
Vgs OF PFET (2V/DIV)
Vgs = 0
Vs OF PFET (5V/DIV)
VFB (%)
0
Vs = 0V
–0.05
–0.10
–0.15
Id (REVERSE) OF
PFET (5A/DIV)
–0.20
Id = 0A
1.25µs/DIV
6
11
TEST PERFORMED ON DEMOBOARD
VCHARGE = 12.6V
VIN = 15VDC
CHARGER = ON
INFET = 1/2 Si4925DY
ICHARGE = 3.3V, add an external pull-up.
ACP/SHDN (Pin 3): Open-drain output used to indicate if
the AC adapter voltage is adequate for charging. Active high
digital output. Internal 10µA pull-up to 3.5V. The charger
can also be shutdown by pulling this pin below 1V. The pin
is capable of sinking at least 100µA. If VLOGIC > 3.3V, add
an external pull-up. (LTC4008-1: ACP function disabled.)
RT (Pin 4): Thermistor Clocking Resistor. Use a 150k resistor as a nominal value. This resistor is always required.
If this resistor is not present, the charger will not start.
FAULT (Pin 5): Active low open-drain output that indicates that charger operation has suspended due to the
thermistor exceeding allowed values. A pull-up resistor
is required if this function is used. The pin is capable of
sinking at least 100µA.
GND (Pin 6): Ground for Low Power Circuitry.
VFB (Pin 7): Input of Voltage Feedback Error Amplifier,
EA, in the “Block Diagram”.
NTC (Pin 8): A thermistor network is connected from NTC
to GND. This pin determines if the battery temperature is
safe for charging. The charger and timer are suspended
and the FAULT pin is driven low if the thermistor indicates
a temperature that is unsafe for charging. The thermistor
function may be disabled with a 300k to 500k resistor
from DCIN to NTC.
resistor. The voltage at this pin provides a linear indication of charging current. Peak current is equivalent to
1.19V. Zero current is approximately 0.309V. A capacitor from PROG to ground is required to filter higher frequency components. The maximum program resistance
to ground is 100k. Values higher than 100k can cause the
charger to shut down.
CSP (Pin 11): Current Amplifier CA1 Input. The CSP and
BAT pins measure the voltage across the sense resistor, RSENSE, to provide the instantaneous current signals
required for both peak and average current mode operation.
BAT (Pin 12): Battery Sense Input and the Negative
Reference for the Current Sense Resistor.
BATMON (Pin 13): Output Voltage Representing Battery
Voltage. Switched off to reduce standby current drain
when AC is not present. An external voltage divider
from BATMON to VFB sets the charger float voltage.
Recommended minimum load resistance is 100k.
FLAG (Pin 14): Active low open-drain output that indicates when charging current has declined to 10% of max
programmed current. A pull-up resistor is required if this
function is used. The pin is capable of sinking at least
100µA. This function is latching. To clear it, user must
cycle the ACP/SHDN pin.
CLN (Pin 15): Negative Input to the Input Current Limiting
Amplifier CL1. The threshold is set at 100mV below the
voltage at the CLP pin. When used to limit input current,
a filter is needed to filter out the switching noise. If no
current limit function is desired, connect this pin to CLP.
CLP (Pin 16): This pin serves as a positive reference for
the input current limit amplifier, CL1. It also serves as the
power supply for the IC.
ITH (Pin 9): Control Signal of the Inner Loop of the Current
Mode PWM. Higher ITH voltage corresponds to higher
charging current in normal operation. A 6k resistor in
series with a capacitor of at least 0.1µF to GND provides
loop compensation. Typical full-scale output current is
40µA. Nominal voltage range for this pin is 0V to 3V.
TGATE (Pin 17): Drives the top external PMOSFET of the
battery charger buck converter.
PROG (Pin 10): Current Programming/Monitoring Input/
Output. An external resistor to GND programs the peak
charging current in conjunction with the current sensing
INFET (Pin 20): Drives the gate of the external input
P-MOSFET. (LTC4008-1: No Connection)
PGND (Pin 18): High Current Ground Return for BGATE
Driver.
BGATE (Pin 19): Drives the bottom external N-MOSFET
of the battery charger buck converter.
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Rev C
7
LTC4008
BLOCK DIAGRAM
0.1µF
VIN
DCIN
INFET*
Q3
1
5.8V
20
*
*
–
+
*NOT USED IN THE LTC4008-1
CLP
ACP/SHDN 3
CONTROL
BLOCK
FAULT 5
TBAD
OSCILLATOR
4
THERMISTOR
8
150k
RT
NTC
32.4k
10k
NTC
0.47µF
C/10
FLAG 14
BATMON
397mV
+
35mV
6
11.67µA
–
GND
–
13
+
CLN
–
16
100mV
15
gm = 1.4m
11
CSP
20µF
3k
9k
CL1
+
gm = 1m
–
Ω
0.1µF
5k
–
7
3k
RSENSE
CA1
Ω
RCL
EA
+
CLP
Ω
VFB
gm = 1m
BAT
–
+
1.19V
12
CA2
ICL 2
+
DCIN
OSCILLATOR
WATCHDOG
DETECT tOFF
20µF
1.19V
9
+
1.28V
–
OV
÷5
ITH
6K
BUFFERED ITH
0.12µF
CLP
BGATE
PGND
17
19
Q
PWM
LOGIC
S
R
ICMP
–+
–
Q2
TGATE
+
Q1
CHARGE
18
IREV
–
+
L1
17mV
10
PROG
4.7nF
RPROG
26.7k
4008 BD
Rev C
8
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LTC4008
TEST CIRCUIT
7
VFB
LTC4008
+
VREF
EA
–
13
BATMON
+
–
11
CSP
12
BAT
9
90.325k
ITH
+
LT1055
–
9.675k
0.6V
4008 TC
OPERATION
OVERVIEW
The LTC4008 is a synchronous current mode PWM step
down (buck) switcher battery charger controller. The
charge current is programmed by the combination of a
program resistor (RPROG) from the PROG pin to ground
and a sense resistor (RSENSE) between the CSP and BAT
pins. The final float voltage is programmed with an external resistor divider and the internal 1.19V reference voltage. Charging begins when the potential at the DCIN pin
rises above the voltage at BAT (and the UVLO voltage)
and the ACP/SHDN pin is high. An external thermistor
network is sampled at regular intervals. If the thermistor
value exceeds design limits, charging is suspended and
the FAULT pin is set low. If the thermistor value returns
to an acceptable value, charging resumes and the FAULT
pin is set high. An external resistor on the RT pin sets the
sampling interval for the thermistor.
As the battery approaches the final float voltage, the
charge current will begin to decrease. When the current
drops to 10% of the full-scale charge current, an internal
C/10 comparator will indicate this condition by latching
the FLAG pin low. If this condition is caused by an input
current limit condition, described below, then the FLAG
indicator will be inhibited. When the input voltage is not
present, the charger goes into a sleep mode, dropping
battery current drain to 15µA. This greatly reduces the
current drain on the battery and increases the standby
time. The charger can be inhibited at any time by forcing
the ACP/SHDN pin to a low voltage. Forcing ACP/SHDN
low, or removing the voltage from DCIN, will also clear
the FLAG pin if it is low.
Table 1. Truth Table For Indicator States
MODE
DCIN
ACP/SHDN
FLAG**
FAULT**
ICL
Shutdown by low adapter voltage (Disabled on LTC4008-1)
BAT
HIGH
HIGH
HIGH*
HIGH*
Input current limited charging
>BAT
HIGH
HIGH*
HIGH*
LOW
Charger shut down due to thermistor out of range
>BAT
HIGH
X
LOW
HIGH
X
Forced LOW
HIGH
HIGH
LOW
>BAT + 7% of programmed value). In this
case, both MOSFETs are turned off until the overvoltage
condition is cleared. This feature is useful for batteries
which “load dump” themselves by opening their protection switch to perform functions such as calibration or
pulse mode charging.
The thermistor detection circuit is shown in Figure 3. It
requires an external resistor and capacitor in order to
function properly.
LTC4008
R10
32.4k
RTH
10k
NTC
C7
0.47µF
8
CLK
–
NTC
S1
+
+
60k
–
–
PWM Watchdog Timer
There is a watchdog timer that observes the activity on
the BGATE and TGATE pins. If TGATE stops switching for
more than 40µs, the watchdog activates and turns off the
top MOSFET for about 400ns. The watchdog engages to
prevent very low frequency operation in dropout which
is a potential source of audible noise when using ceramic
input and output capacitors.
~4.5V
45k
+
15k
D
Q
TBAD
C
4008 F03
Figure 3.
Rev C
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11
LTC4008
OPERATION
The thermistor detector performs a sample-and-hold
function. An internal clock, whose frequency is determined by the timing resistor connected to RT , keeps
switch S1 closed to sample the thermistor:
This voltage is stored by C7. Then the switch is opened
for a short period of time to read the voltage across the
thermistor.
tSAMPLE = 127.5 • 20 • RRT • 17.5pF = 6.7ms,
for RRT = 150k
for RRT = 150k
The external RC network is driven to approximately 4.5V
and settles to a final value across the thermistor of:
VRTH(FINAL) =
4.5V • R TH
R TH +R10
tHOLD = 10 • RRT • 17.5pF = 26µs,
When the tHOLD interval ends the result of the thermistor
testing is stored in the D flip-flop (DFF). If the voltage at
NTC is within the limits provided by the resistor divider
feeding the comparators, then the NOR gate output will
be low and the DFF will set TBAD to zero and charging will
continue. If the voltage at NTC is outside of the resistor
divider limits, then the DFF will set TBAD to one, the charger
will be shut down, FAULT pin is set low and the timer will
be suspended until TBAD returns to zero (Figure 4).
CLK
(NOT TO
SCALE)
tHOLD
VOLTAGE ACROSS THERMISTOR
tSAMPLE
COMPARATOR HIGH LIMIT
VNTC
COMPARATOR LOW LIMIT
4008 F04
Figure 4.
Rev C
12
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LTC4008
APPLICATIONS INFORMATION
Charger Current Programming
The basic formula for charging current is:
V
• 3.01kΩ / R PROG – 0.035V
ICHARGE(MAX) = REF
R SENSE
VREF = 1.19V. This leaves two degrees of freedom: RSENSE
and RPROG. The 3.01k input resistors must not be altered
since internal currents and voltages are trimmed for this
value. Pick RSENSE by setting the average voltage between
CSP and BAT to be close to 100mV during maximum
charger current. Then RPROG can be determined by solving the above equation for RPROG.
R PROG =
VREF • 3.01kΩ
R SENSE •ICHARGE(MAX) + 0.035V
Table 2. Recommended RSNS and RPROG Resistor Values
IMAX (A)
RSENSE (Ω) 1%
RSENSE (W)
RPROG (kΩ) 1%
1.0
0.100
0.25
26.7
2.0
0.050
0.25
26.7
3.0
0.033
0.5
26.7
4.0
0.025
0.5
26.7
Charging current can be programmed by pulse width modulating RPROG with a switch Q1 to RPROG at a frequency
higher than a few kHz (Figure 5). CPROG must be increased
to reduce the ripple caused by the RPROG switching. The
compensation capacitor at ITH will probably need to be
LTC4008
PROG
10
CPROG
RPROG
5V
0V
RZ
102k
Q1
2N7002
4008 F05
Figure 5. PWM Current Programming
increased also to improve stability and prevent large
overshoot currents during start-up conditions. Charging
current will be proportional to the duty cycle of the switch
with full current at 100% duty cycle and zero current when
Q1 is off.
Maintaining C/10 Accuracy
The C/10 comparator threshold that drives the FLAG pin
has a fixed threshold of approximately VPROG = 400mV.
This threshold works well when RPROG is 26.7k, but will
not yield a 10% charging current indication if RPROG
is a different value. There are situations where a standard value of RSENSE will not allow the desired value of
charging current when using the preferred RPROG value.
In these cases, where the full-scale voltage across RSENSE
is within ±20mV of the 100mV full-scale target, the input
resistors connected to CSP and BAT can be adjusted to
provide the desired maximum programming current as
well as the correct FLAG trip point.
For example, the desired max charging current is 2.5A but
the best RSENSE value is 0.033Ω. In this case, the voltage across RSENSE at maximum charging current is only
82.5mV, normally RPROG would be 30.1k but the nominal
FLAG trip point is only 5% of maximum charging current.
If the input resistors are reduced by the same amount as
the full-scale voltage is reduced then, R4 = R5 = 2.49k and
RPROG = 26.7k, the maximum charging current is still 2.5A
but the FLAG trip point is maintained at 10% of full scale.
There are other effects to consider. The voltage across the
current comparator is scaled to obtain the same values
as the 100mV sense voltage target, but the input referred
sense voltage is reduced, causing some careful consideration of the ripple current. Input referred maximum
comparator threshold is 117mV, which is the same ratio
of 1.4x the DC target. Input referred IREV threshold is
scaled back to –24mV. The current at which the switcher
starts will be reduced as well so there is some risk of
boost activity. These concerns can be addressed by using
a slightly larger inductor to compensate for the reduction
of tolerance to ripple current.
Rev C
For more information www.analog.com
13
LTC4008
APPLICATIONS INFORMATION
Battery Conditioning
Table 3.
Some batteries require a small charging current to condition them when they are severely depleted. The charging
current is switched to a high rate after the battery voltage has reached a “safe” voltage to do so. Figure 6 illustrates how to do this 2-level charging. When Q1 is on, the
charger current is set to maximum. When Q1 is off, the
charging current is set to 10% of the maximum.
R9 (kΩ) 0.25%
R8 (kΩ) 0.25%
8.2
24.9
147
8.4
26.1
158
12.3
15
140
12.6
16.9
162
16.4
11.5
147
16.8
13.3
174
Soft-Start
LTC4008
PROG
10
CPROG
0.0047µF
R1
26.7k
Q1
2N7002
FLOAT VOLTAGE (V)
R2
53.6k
4008 F06
Figure 6. 2-Level Current Programming
Charger Voltage Programming
A resistor divider, R8 and R9 (see Figure 10), programs
the final float voltage of the charger. The equation for float
voltage is (the input bias current of EA is typically –4nA
and can be ignored):
VFLOAT = VREF (1 + R8/R9)
It is recommended that the sum of R8 and R9 not be less
than 100k. Accuracy of the LTC4008 voltage reference is
±0.8% at 25°C, and ±1% over the full temperature range.
This leads to the possibility that very accurate (0.1%)
resistors might be needed for R8 and R9. Actually, the
temperature of the LTC4008 will rarely exceed 50°C near
the float voltage because charging currents have tapered
to a low level, so 0.25% resistors will normally provide the
required level of overall accuracy. Table 3 contains recommended values for R8 and R9 for popular float voltages.
The LTC4008 is soft started by the 0.12µF capacitor on the
ITH pin. On start-up, ITH pin voltage will rise quickly to 0.5V,
then ramp up at a rate set by the internal 40µA pull-up current and the external capacitor. Battery charging current
starts ramping up when ITH voltage reaches 0.8V and full
current is achieved with ITH at 2V. With a 0.12µF capacitor,
time to reach full charge current is about 2ms and it is
assumed that input voltage to the charger will reach full
value in less than 2ms. The capacitor can be increased up
to 1µF if longer input start-up times are needed.
Input and Output Capacitors
The input capacitor (C2) is assumed to absorb all input
switching ripple current in the converter, so it must have
adequate ripple current rating. Worst-case RMS ripple
current will be equal to one-half of output charging current. Actual capacitance value is not critical. Solid tantalum low ESR capacitors have high ripple current rating
in a relatively small surface mount package, but caution
must be used when tantalum capacitors are used for input
or output bypass. High input surge currents can be created when the adapter is hot-plugged to the charger or
when a battery is connected to the charger. Solid tantalum
capacitors have a known failure mechanism when subjected to very high turn-on surge currents. Kemet T495
series of “Surge Robust” low ESR tantalums are rated for
high surge conditions such as battery to ground.
Rev C
14
For more information www.analog.com
LTC4008
APPLICATIONS INFORMATION
The relatively high ESR of an aluminum electrolytic for
C1, located at the AC adapter input terminal, is helpful
in reducing ringing during the hot-plug event. Refer to
Application Note 88 for more information.
Highest possible voltage rating on the capacitor will minimize problems. Consult with the manufacturer before use.
Alternatives include high capacity ceramic (at least 20µF)
from Tokin, United Chemi-Con/Marcon, et al. Other alternative capacitors include OS-CON capacitors from Sanyo.
The output capacitor (C3) is also assumed to absorb
output switching current ripple. The general formula for
capacitor current is:
⎛
0.29 ( VBAT ) ⎜ 1–
⎝
IRMS =
(L1) ( f )
VBAT ⎞
⎟
VDCIN ⎠
∆IL =
⎞
⎛ V
VOUT ⎜ 1– OUT ⎟
VIN ⎠
( f ) (L )
⎝
1
Accepting larger values of ∆IL allows the use of low inductances, but results in higher output voltage ripple and
greater core losses. A reasonable starting point for setting
ripple current is ∆IL = 0.4(IMAX). In no case should ∆IL
exceed 0.6(IMAX) due to limits imposed by IREV and CA1.
Remember the maximum ∆IL occurs at the maximum
input voltage. In practice 10µH is the lowest value recommended for use.
Lower charger currents generally call for larger inductor
values. Use Table 4 as a guide for selecting the correct
inductor value for your application.
Table 4.
For example:
MAXIMUM AVERAGE
CURRENT (A)
INPUT
VOLTAGE (V)
MINIMUM INDUCTOR
VALUE (µH)
1
≤20
40 ±20%
VDCIN = 19V, VBAT = 12.6V, L1 = 10µH, and
f = 300kHz, IRMS = 0.41A.
1
>20
56 ±20%
2
≤20
20 ±20%
EMI considerations usually make it desirable to minimize ripple current in the battery leads, and beads or
inductors may be added to increase battery impedance
at the 300kHz switching frequency. Switching ripple current splits between the battery and the output capacitor
depending on the ESR of the output capacitor and the battery impedance. If the ESR of C3 is 0.2Ω and the battery
impedance is raised to 4Ω with a bead or inductor, only
5% of the current ripple will flow in the battery.
2
>20
30 ±20%
3
≤20
15 ±20%
3
>20
20 ±20%
4
≤20
10 ±20%
4
>20
15 ±20%
Inductor Selection
Higher operating frequencies allow the use of smaller
inductor and capacitor values. A higher frequency generally results in lower efficiency because of MOSFET gate
charge losses. In addition, the effect of inductor value
on ripple current and low current operation must also
be considered. The inductor ripple current ∆IL decreases
with higher frequency and increases with higher VIN.
Charger Switching Power MOSFET
and Diode Selection
Two external power MOSFETs must be selected for use
with the charger: a P-channel MOSFET for the top (main)
switch and an N-channel MOSFET for the bottom (synchronous) switch.
The peak-to-peak gate drive levels are set internally. This
voltage is typically 6V. Consequently, logic-level threshold
MOSFETs must be used. Pay close attention to the BVDSS
specification for the MOSFETs as well; many of the logic
level MOSFETs are limited to 30V or less.
Rev C
For more information www.analog.com
15
LTC4008
APPLICATIONS INFORMATION
Selection criteria for the power MOSFETs include the “ON”
resistance RDS(ON), total gate capacitance QG, reverse
transfer capacitance CRSS , input voltage and maximum
output current. The charger is operating in continuous
mode so the duty cycles for the top and bottom MOSFETs
are given by:
Synchronous Switch Duty Cycle = (VIN – VOUT)/VIN.
The Schottky diode D1, shown in the “Typical Application”
on the back page, conducts during the dead-time between
the conduction of the two power MOSFETs. This prevents
the body diode of the bottom MOSFET from turning on
and storing charge during the dead-time, which could cost
as much as 1% in efficiency. A 1A Schottky is generally
a good size for 4A regulators due to the relatively small
average current. Larger diodes can result in additional
transition losses due to their larger junction capacitance.
The MOSFET power dissipations at maximum output current are given by:
The diode may be omitted if the efficiency loss can be
tolerated.
Main Switch Duty Cycle = VOUT/VIN
PMAIN = VOUT/VIN(IMAX)2(1 + δ∆T)RDS(ON)
+ k(VIN)2(IMAX)(CRSS)(fOSC)
PSYNC = (VIN – VOUT)/VIN(IMAX)2(1 + δ∆T)RDS(ON)
Where δ∆T is the temperature dependency of RDS(ON)
and k is a constant inversely related to the gate drive
current. Both MOSFETs have I2R losses while the PMAIN
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON)
device with lower CRSS actually provides higher efficiency. The synchronous MOSFET losses are greatest at
high input voltage or during a short circuit when the duty
cycle in this switch in nearly 100%. The term (1 + δ∆T) is
generally given for a MOSFET in the form of a normalized
RDS(ON) vs temperature curve, but δ = 0.005/°C can be
used as an approximation for low voltage MOSFETs. CRSS
= QGD/∆VDS is usually specified in the MOSFET characteristics. The constant k = 2 can be used to estimate the
contributions of the two terms in the main switch dissipation equation.
If the charger is to operate in low dropout mode or with
a high duty cycle greater than 85%, then the topside
P-channel efficiency generally improves with a larger
MOSFET. Using asymmetrical MOSFETs may achieve cost
savings or efficiency gains.
Calculating IC Power Dissipation
The power dissipation of the LTC4008 is dependent upon
the gate charge of the top and bottom MOSFETs (QG1 &
QG2 respectively) The gate charge is determined from the
manufacturer’s data sheet and is dependent upon both
the gate voltage swing and the drain voltage swing of the
MOSFET. Use 6V for the gate voltage swing and VDCIN for
the drain voltage swing.
PD = VDCIN • (fOSC (QG1 + QG2) + IQ)
Example:
VDCIN = 19V, fOSC = 345kHz, QG1 = QG2 = 15nC,
IQ = 5mA
PD = 292mW
Adapter Limiting
An important feature of the LTC4008 is the ability to
automatically adjust charging current to a level which
avoids overloading the wall adapter. This allows the product to operate at the same time that batteries are being
charged without complex load management algorithms.
Additionally, batteries will automatically be charged at the
maximum possible rate of which the adapter is capable.
This feature is created by sensing total adapter output current and adjusting charging current downward if a preset
adapter current limit is exceeded. True analog control is
Rev C
16
For more information www.analog.com
LTC4008
APPLICATIONS INFORMATION
used, with closed-loop feedback ensuring that adapter load
current remains within limits. Amplifier CL1 in Figure 7
senses the voltage across RCL, connected between the
CLP and CLN pins. When this voltage exceeds 100mV,
the amplifier will override programmed charging current
to limit adapter current to 100mV/RCL. A lowpass filter
formed by 5kΩ and 15nF is required to eliminate switching noise. If the current limit is not used, CLN should be
connected to CLP.
Note that the ICL pin will be asserted when the voltage
across RCL is 93mV, before the adapter limit regulation
threshold.
VIN
LTC4008
–
CL1
16
100mV
+
15
CLP
CLN
RCL*
15nF
5k
+
CIN
TO
SYSTEM
LOAD
NTC 8
R9
C7
RTH
4008 F08
Figure 8. Voltage Divider Thermistor Network
Table 5. Common RCL Resistor Values
ADAPTER
RATING (A)
RCL VALUE*
(Ω) 1%
RCL POWER
DISSIPATION (W)
RCL POWER
RATING (W)
1.5
0.06
0.135
0.25
1.8
0.05
0.162
0.25
2
0.045
0.18
0.25
2.3
0.039
0.206
0.25
2.5
0.036
0.225
0.5
2.7
0.033
0.241
0.5
3
0.03
0.27
0.5
*Values shown above are rounded to nearest standard value.
4008 F07
*RCL =
LTC4008
100mV
ADAPTER CURRENT LIMIT
As is often the case, the wall adapter will usually have at
least a +10% current limit margin and many times one
can simply set the adapter current limit value to the actual
adapter rating (see Table 5).
Figure 7. Adapter Current Limiting
Setting Input Current Limit
To set the input current limit, you need to know the minimum wall adapter current rating. Subtract 7% for the
input current limit tolerance and use that current to determine the resistor value.
Designing the Thermistor Network
There are several networks that will yield the desired function of voltage vs temperature needed for proper operation of the thermistor. The simplest of these is the voltage
RCL = 100mV/ILIM
divider shown in Figure 8. Unfortunately, since the HIGH/
ILIM = Adapter Min Current –
LOW comparator thresholds are fixed internally, there is
(Adapter Min Current • 7%)
only one thermistor type that can be used in this network;
the thermistor must have a HIGH/LOW resistance ratio of
1:7. If this happy circumstance is true for you, then simply
set R9 = RTH(LOW)
Rev C
For more information www.analog.com
17
LTC4008
APPLICATIONS INFORMATION
LTC4008
NTC 8
Example #2: 100kΩ NTC
R9
C7
R9A
RTH
4008 F09
Figure 9. General Thermistor Network
If you are using a thermistor that doesn’t have a 1:7 HIGH/
LOW ratio, or you wish to set the HIGH/LOW limits to
different temperatures, then the more generic network in
Figure 9 should work.
Once the thermistor, RTH , has been selected and the
thermistor value is known at the temperature limits, then
resistors R9 and R9A are given by:
For NTC thermistors:
R9 = 6 RTH(LOW) • RTH(HIGH)/(RTH(LOW) – RTH(HIGH))
R9A = 6 RTH(LOW) • RTH(HIGH)/(RTH(LOW) – 7 •
RTH(HIGH))
Where RTH(LOW) > 7 • RTH(HIGH)
For PTC thermistors:
R9 = 6 RTH(LOW) • RTH(HIGH)/(RTH(HIGH) – RTH(LOW))
R9A = 6 RTH(LOW) • RTH(HIGH)/(RTH(HIGH) – 7 •
RTH(LOW))
Where RTH(HIGH) > 7 • RTH(LOW)
Example #1: 10kΩ NTC with custom limits
TLOW = 5°C, THIGH = 50°C
RTH = 100k at 25°C,
RTH(LOW) = 272.05k at 5°C
RTH(HIGH) = 33.195k at 50°C
R9 = 226.9k → 226k (nearest 1% value)
R9A = 1.365M → 1.37M (nearest 1% value)
Example #3: 22kΩ PTC
TLOW = 0°C, THIGH = 50°C
RTH = 22k at 25°C,
RTH(LOW) = 6.53k at 0°C
RTH(HIGH) = 61.4k at 50°C
R9 = 43.9k → 44.2k (nearest 1% value)
R9A = 154k
Sizing the Thermistor Hold Capacitor
During the hold interval, C7 must hold the voltage across
the thermistor relatively constant to avoid false readings.
A reasonable amount of ripple on NTC during the hold
interval is about 10mV to 15mV. Therefore, the value of
C7 is given by:
C7 = tHOLD/(R9/7 • – ln(1 – 8 • 15mV/4.5V))
= 10 • RRT • 17.5pF/(R9/7 • – ln(1 – 8 • 15mV/4.5V)
Example:
R9 = 24.3k
RRT = 150k
C7 = 0.28µF → 0.27µF (nearest value)
TLOW = 0°C, THIGH = 50°C
RTH = 10k at 25°C,
RTH(LOW) = 32.582k at 0°C
RTH(HIGH) = 3.635k at 50°C
R9 = 24.55k → 24.3k (nearest 1% value)
R9A = 99.6k → 100k (nearest 1% value)
Rev C
18
For more information www.analog.com
LTC4008
APPLICATIONS INFORMATION
Disabling the Thermistor Function
If the thermistor is not needed, connecting a resistor
between DCIN and NTC will disable it. The resistor should
be sized to provide at least 10µA with the minimum voltage applied to DCIN and 10V at NTC. Do not exceed 30µA.
Generally, a 301k resistor will work for DCIN less than
15V. A 499k resistor is recommended for DCIN between
15V and 24V.
Using the LTC4008-1 (Refer to Figure 10)
The LTC4008-1 is intended for applications where the
battery power is fully isolated from the charger and wall
adapter connections. An example application is a system with multiple batteries such that the charger’s output power passes through a downstream power path or
selector system. Typically these systems also provide isolation and control the wall adapter power. To reduce cost
in such systems, the LTC4008-1 removes the requirement
for the wall adapter INFET function or blocking diode.
Wall adapter or ACP detection is also removed along with
micropower shutdown mode. Asserting of the SHDN pin
only puts the charger into standby mode. Failure to isolate
the battery power from ANY of the LTC4008-1 pins when
wall adapter power is removed or lost will only drain the
battery at the IC quiescent current rate. More specifically,
high current is drawn from the DCIN, CLP and CLN pins.
Suggested devices to isolate power from the charger
include simple diodes, electrical or mechanical switches
or power path control devices such as the LTC4412 low
loss PowerPath™ controller.
Because the switcher operation is continuous under nearly
all conditions, precautions must be taken to prevent the
charger from boosting the input voltage above maximum
voltage values on the input capacitors or adapter. Z1 and
Q3 will shut down the charger if the input voltage exceeds
a safe value.
Rev C
For more information www.analog.com
19
LTC4008
APPLICATIONS INFORMATION
DCIN
0V TO 28V
R8
140k
0.25%
VLOGIC
R11
100k
ICL
R12
100k
FAULT
FLAG
R10 32.4k 1%
Z1
Q3
2N7002
R13
1.5k
C7
0.47µF
THERMISTOR
10k
NTC
RT
150k
R9
15k
0.25%
R7
6.04k
1%
C6
0.12µF
C1
0.1µF
DCIN
BATMON
LTC4008-1
VFB
ICL
CLP
SHDN
CLN
FAULT
TGATE
FLAG
BGATE
NTC
PGND
RT
CSP
ITH
BAT
GND
C4
15nF R1
4.99k
1%
RCL
0.02Ω
1%
SYSTEM
LOAD
C2
20µF
L1
10µH
Q1
Q2
D2
RSENSE
0.025Ω
1%
Q4
D1
C3
20µF
R4 3.01k 1%
R26
150k
Li-Ion
BATTERY
R5 3.01k 1%
PROG
C5
0.0047µF
R6
28.7k
1%
D1: MBRS130T3
D2: SBM540
Q1: Si4431BDY
Q2: FDC645N
Q4: Si7423DN
Z1 VALUE SIZED FOR ABSOLUTE MAXIMUM ADAPTER VOLTAGE
Q5
2N7002
CHARGE
4008 F10
Figure 10. Typical LTC4008-1 Application (12.3V/4A)
Rev C
20
For more information www.analog.com
LTC4008
APPLICATIONS INFORMATION
PCB Layout Considerations
For maximum efficiency, the switch node rise and fall
times should be minimized. To prevent magnetic and
electrical field radiation and high frequency resonant
problems, proper layout of the components connected
to the IC is essential. (See Figure 11.) Here is a PCB layout
priority list for proper layout. Layout the PCB using this
specific order.
1. Input capacitors need to be placed as close as possible to switching FET’s supply and ground connections. Shortest copper trace connections possible.
These parts must be on the same layer of copper. Vias
must not be used to make this connection.
2. The control IC needs to be close to the switching
FET ’s gate terminals. Keep the gate drive signals short
for a clean FET drive. This includes IC supply pins that
connect to the switching FET source pins. The IC can
be placed on the opposite side of the PCB relative to
above.
3. Place inductor input as close as possible to switching
FET’s output connection. Minimize the surface area of
this trace. Make the trace width the minimum amount
needed to support current—no copper fills or pours.
Avoid running the connection using multiple layers in
parallel. Minimize capacitance from this node to any
other trace or plane.
4. Place the output current sense resistor right next to
the inductor output but oriented such that the IC’s current sense feedback traces going to resistor are not
long. The feedback traces need to be routed together
as a single pair on the same layer at any given time
with smallest trace spacing possible. Locate any filter
component on these traces next to the IC and not at
the sense resistor location.
5. Place output capacitors next to the sense resistor
output and ground.
6. Output capacitor ground connections need to feed
into same copper that connects to the input capacitor
ground before tying back into system ground.
7. Connection of switching ground to system ground or
internal ground plane should be single point. If the
system has an internal system ground plane, a good
way to do this is to cluster vias into a single star point
to make the connection.
8. Route analog ground as a trace tied back to IC ground
(analog ground pin if present) before connecting to
any other ground. Avoid using the system ground
plane. CAD trick: make analog ground a separate
ground net and use a 0Ω resistor to tie analog ground
to system ground.
9. A good rule of thumb for via count for a given high
current path is to use 0.5A per via. Be consistent.
10. If possible, place all the parts listed above on the same
PCB layer.
11. Copper fills or pours are good for all power connections except as noted above in Rule 3. You can also
use copper planes on multiple layers in parallel too—
this helps with thermal management and lower trace
inductance improving EMI performance further.
12. For best current programming accuracy provide a
Kelvin connection from RSENSE to CSP and BAT. See
Figure 12 as an example.
It is important to keep the parasitic capacitance on the RT ,
CSP and BAT pins to a minimum. The traces connecting
these pins to their respective resistors should be as short
as possible.
Rev C
For more information www.analog.com
21
LTC4008
APPLICATIONS INFORMATION
DIRECTION OF CHARGING CURRENT
SWITCH NODE
L1
VBAT
VIN
C2
HIGH
FREQUENCY
CIRCULATING
PATH
RSENSE
D1
C3
BAT
4008 F12
BAT
CSP
4008 F11
Figure 11. High Speed Switching Path
Figure 12. Kelvin Sensing of Charging Current
PACKAGE DESCRIPTION
GN Package
20-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.337 – .344*
(8.560 – 8.738)
.045 ±.005
20 19 18 17 16 15 14 13 12
.254 MIN
.150 – .165
.0165 ±.0015
11
.229 – .244
(5.817 – 6.198)
.058
(1.473)
REF
.150 – .157**
(3.810 – 3.988)
.0250 BSC
1
RECOMMENDED SOLDER PAD LAYOUT
.015 ± .004
× 45°
(0.38 ± 0.10)
.0075 – .0098
(0.19 – 0.25)
2 3
4
5 6
7
8
.0532 – .0688
(1.35 – 1.75)
9 10
.004 – .0098
(0.102 – 0.249)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
.0250
(0.635)
BSC
GN20 (SSOP) 0204
Rev C
22
For more information www.analog.com
LTC4008
REVISION HISTORY
(Revision history begins at Rev B)
REV
DATE
DESCRIPTION
B
7/10
Updated Figure 5 and Figure 6.
C
4/18
Changed LTC4008 pin configuration and lowered TJMAX and θJA values
PAGE NUMBER
13, 14
2
Rev C
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog
Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications
subject to change without notice. No license For
is granted
implication or
otherwise under any patent or patent rights of Analog Devices.
more by
information
www.analog.com
23
LTC4008
TYPICAL APPLICATION
NiMH/4A Battery Charger
Q3
INPUT SWITCH
DCIN
0V TO 20V
VLOGIC
ICL
R8
147k
0.25%
R11
100k
R12
100k
ACP
C1
0.1µF
BATMON
DCIN
VFB
INFET
ICL
LTC4008 CLP
FLAG
R10 32.4k 1%
FAULT
TGATE
FLAG
BGATE
NTC
PGND
BAT
ITH
THERMISTOR
10k
NTC
RT
150k
R7
6.04k
1%
GND
SYSTEM
LOAD
C2
20µF
L1
10µH
Q1
Q2
RSENSE
0.025Ω
1%
D1
C3
20µF
CSP
RT
R9
C7
13.3k
0.47µF 0.25%
RCL
0.02Ω
1%
CLN
ACP/SHDN
FAULT
C4
0.1µF
R1 4.99k 1%
R4 3.01k 1%
R5 3.01k 1%
PROG
C5
0.0047µF
R6
26.7k
1%
C6
0.12µF
NiMH
BATTERY
PACK
CHARGING
CURRENT
MONITOR
D1: MBRS130T3
Q1: Si4431BDY
Q2: FDC645N
4008 TA02
RELATED PARTS
PART NUMBER DESCRIPTION
COMMENTS
LT®1511
3A Constant-Current/Constant-Voltage Battery Charger
High Efficiency, Minimum External Components to Fast Charge Lithium,
NIMH and NiCd Batteries
LT1513
Sepic Constant- or Programmable-Current/ConstantVoltage Battery Charger
Charger Input Voltage May be Higher, Equal to or Lower Than Battery Voltage,
500kHz Switching Frequency
LT1571
1.5A Switching Charger
1- or 2-Cell Li-Ion, 500kHz or 200kHz Switching Frequency, Termination Flag
LTC1628-PG
2-Phase, Dual Synchronous Step-Down Controller
Minimizes CIN and COUT , Power Good Output, 3.5V ≤ VIN ≤ 36V
LTC1709 Family 2-Phase, Dual Synchronous Step-Down Controller
with VID
Up to 42A Output, Minimum CIN and COUT , Uses Smallest Components for
Intel and AMD Processors
LTC1729
Li-Ion Battery Charger Termination Controller
Trickle Charge Preconditioning, Temperature Charge Qualification, Time or
Charge Current Termination, Automatic Charger and Battery Detection, and
Status Output
LT1769
2A Switching Battery Charger
Constant-Current/Constant-Voltage Switching Regulator, Input Current
Limiting Maximizes Charge Current
LTC1778
Wide Operating Range, No RSENSE™ Synchronous
Step-Down Controller
2% to 90% Duty Cycle at 200kHz, Stable with Ceramic COUT
LTC1960
Dual Battery Charger/Selector with SPI Interface
Simultaneous Charge or Discharge of Two Batteries, DAC Programmable
Current and Voltage, Input Current Limiting Maximizes Charge Current
LTC3711
No RSENSE Synchronous Step-Down Controller
with VID
3.5V ≤ VIN ≤ 36V, 0.925V ≤ VOUT ≤ 2V, for Transmeta, AMD and Intel Mobile
Processors
LTC4006
Small, High Efficiency, Fixed Voltage, Lithium-Ion
Battery Charger with Termination
Complete Charger for 3- or 4-Cell Li-Ion Batteries, AC Adapter Current Limit
and Thermistor Sensor, 16-Pin Narrow SSOP Package
LTC4007
High Efficiency, Programmable Voltage
Battery Charger with Termination
Complete Charger for 3- or 4-Cell Li-Ion Batteries, AC Adapter Current Limit,
Thermistor Sensor and Indicator Outputs
LTC4100
Smart Battery Charger Controller
SMBus Rev 1.1 Compliant
Rev C
24
D16841-0-4/18(C)
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