1977IFE

LT1977 High Voltage 1.5A, 500kHz Step-Down Switching Regulator with 100µA Quiescent Current FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO Wide Input Range: 3.3V to 60V 1.5A Peak Switch Current Burst Mode® Operation: 100µA Quiescent Current** Low Shutdown Current: IQ < 1µA Power Good Flag with Programmable Threshold Load Dump Protection to 60V 500kHz Switching Frequency Saturating Switch Design: 0.2Ω On-Resistance Peak Switch Current Maintained Over Full Duty Cycle Range* 1.25V Feedback Reference Voltage Easily Synchronizable Soft-Start Capability Small 16-Pin Thermally Enhanced TSSOP Package The LT®1977 is a 500kHz monolithic buck switching regulator that accepts input voltages up to 60V. A high efficiency 1.5A, 0.2Ω switch is included on the die along with all the necessary oscillator, control and logic circuitry. Current mode topology is used for fast transient response and good loop stability. Innovative design techniques along with a new high voltage process achieve high efficiency over a wide input range. Efficiency is maintained over a wide output current range by employing Burst Mode operation at low currents, utilizing the output to bias the internal circuitry, and by using a supply boost capacitor to fully saturate the power switch. Patented circuitry maintains peak switch current over the full duty cycle range.* Shutdown reduces input supply current to less than 1µA. External synchronization can be implemented by driving the SYNC pin with logic-level inputs. A single capacitor from the CSS pin to the output provides a controlled output voltage ramp (soft-start). The device also has a power good flag with a programmable threshold and time-out and thermal shutdown protection. The LT1977 is available in a 16-pin TSSOP package with exposed pad leadframe for low thermal resistance. The LT1976, a 200kHz reduced switch frequency version of the LT1977, is also available. See the Applications Information section for selection criteria between the LT1976 and LT1977. Efficiency and Power Loss vs Load Current 100 VIN = 12V EFFICIENCY 5V 75 EFFICIENCY (%) APPLICATIO S ■ ■ ■ ■ ■ High Voltage Power Conversion 14V and 42V Automotive Systems Industrial Power Systems Distributed Power Systems Battery-Powered Systems , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a registered trademark of Linear Technology Corporation. *U.S. Patent 6,498,466 and 6,531,909 **See Burst Mode Operation section for conditions TYPICAL APPLICATIO Supply Current vs Input Voltage 150 125 SUPPLY CURRENT (µA) VOUT = 3.3V TA = 25°C 14V to 3.3V Step-Down Converter with 100µA No Load Quiescent Current VIN 2.2µF 100V CER VIN SHDN LT1977 VC 1500pF 26k 1µF CT SYNC GND 330pF CSS VBIAS FB PGFB PG 100k 1% 10pF 165k 1% + 100µF 6.3V TANT BOOST 0.1µF SW 0.1µF 10µH 1N4148 VOUT 3.3V 1A 100 75 50 25 0 0 10 30 40 20 INPUT VOLTAGE (V) 50 60 1977 F05 10MQ100N 1977 TA01 U 10 3.3V 1 POWER LOSS (W) U U 50 TYPICAL POWER LOSS 25 0.1 0.01 0 0.0001 0.001 0.1 0.01 LOAD CURRENT (A) 1 0.001 10 1977 TA02 1977f 1 LT1977 ABSOLUTE (Note 1) AXI U RATI GS PACKAGE/ORDER I FOR ATIO TOP VIEW NC SW NC VIN NC BOOST CT GND 1 2 3 4 5 6 7 8 17 16 PG 15 SHDN 14 SYNC 13 PGFB 12 FB 11 VC 10 BIAS 9 CSS VIN, SHDN, BIAS ..................................................... 60V BOOST Pin Above SW ............................................ 35V BOOST Pin Voltage ................................................. 68V SYNC, CSS, PGFB, FB ................................................ 6V Operating JunctionTemperature Range LT1977EFE (Note 2) ........................ – 40°C to 125°C LT1977IFE (Note 2) ......................... – 40°C to 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER LT1977EFE LT1977IFE FE PART MARKING 1977EFE 1977IFE FE PACKAGE 16-LEAD PLASTIC TSSOP TJMAX = 125°C, θJA = 45°C/W, θJC(PAD) = 10°C/W EXPOSED PAD IS GND (PIN 17) MUST BE SOLDERED TO GND (PIN 8) Consult LTC Marketing for parts specified with wider operating temperature ranges. The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TJ = 25°C. VIN = 12V, SHDN = 12V, BIAS = 5V, FB/PGFB = 1.25V, CSS/SYNC = 0V unless otherwise noted. SYMBOL VSHDN ISHDN IVINS PARAMETER SHDN Threshold SHDN Input Current Minimum Input Voltage (Note 3) Supply Shutdown Current Supply Sleep Current (Note 4) IVIN Supply Quiescent Current Minimum BIAS Voltage (Note 5) IBIASS IBIAS BIAS Sleep Current (Note 4) BIAS Quiescent Current Minimum Boost Voltage (Note 6) Input Boost Current (Note 7) VREF IFB Reference Voltage (VREF) FB Input Bias Current EA Voltage Gain (Note 8) EA Voltage gm EA Source Current EA Sink Current VC to SW gm VC High Clamp IPK SW Current Limit ● ELECTRICAL CHARACTERISTICS CONDITIONS SHDN = 12V SHDN = 0V, BOOST = 0V, FB/PGFB = 0V BIAS = 0V, FB = 1.35V FB = 1.35V BIAS = 0V, FB = 1.15V BIAS = 5V, FB = 1.15V ● ● ● ● ● ● MIN 1.15 TYP 1.3 5 2.4 0.1 170 45 MAX 1.45 20 3 2 250 75 4.10 3.25 UNITS V µA V µA µA µA mA mA V µA µA V mA 2.7 110 700 1.8 40 3 180 900 SYNC = 3.3V ISW = 1.5A ISW = 1.5A 3.3V < VVIN < 60V ● 1.225 1.25 75 900 1.275 200 900 55 40 2.4 3.5 dI(VC)= ± 10µA FB = 1.15V FB = 1.35V 450 20 15 2.1 1.5 650 40 30 3 2.2 2.4 µMho µA µA A/V V A 2 U V nA V/V 1977f W U U WW W LT1977 The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TJ = 25°C. VIN = 12V, SHDN = 12V, BIAS = 5V, FB/PGFB = 1.25V, CSS/SYNC = 0V unless otherwise noted. SYMBOL PARAMETER Switch On Resistance (Note 9) Switching Frequency Maximum Duty Cycle Minimum SYNC Amplitude SYNC Frequency Range SYNC Input Impedance ICSS IPGFB VPGFB ICT VCT CSS Current Threshold (Note 10) PGFB Input Current PGFB Voltage Threshold (Note 11) CT Source Current (Note 11) CT Sink Current (Note 11) CT Voltage Threshold (Note 11) PG Leakage (Note 11) PG Sink Current (Note 11) VPG = 12V PGFB = 1V, PG = 400mV 100 ● ELECTRICAL CHARACTERISTICS CONDITIONS ● ● MIN 425 86 575 7 88 2 1 1.16 TYP 0.2 500 92 1.5 700 85 13 25 90 3.6 2 1.2 0.1 200 MAX 0.4 575 2.0 UNITS Ω kHz % V kHz kΩ 20 100 92 5.5 1.26 1 µA nA % µA mA V µA µA Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LT1977EFE is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT1977IFE is guaranteed and tested over the full –40°C to 125°C operating junction temperature range. Note 3: Minimum input voltage is defined as the voltage where switching starts. Actual minimum input voltage to maintain a regulated output will depend upon output voltage and load current. See Applications Information. Note 4: Supply input current is the quiescent current drawn by the input pin. Its typical value depends on the voltage on the BIAS pin and operating state of the LT1977. With the BIAS pin at 0V, all of the quiescent current required to operate the LT1977 will be provided by the VIN pin. With the BIAS voltage above its minimum input voltage, a portion of the total quiescent current will be supplied by the BIAS pin. Supply sleep current is defined as the quiescent current during the “sleep” portion of Burst Mode operation. See Applications Information for determining application supply currents. Note 5: Minimum BIAS voltage is the voltage on the BIAS pin when IBIAS is sourced into the pin. Note 6: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. Note 7: Boost current is the current flowing into the BOOST pin with the pin held 3.3V above input voltage. It flows only during switch on time. Note 8: Gain is measured with a VC swing from 1.15V to 750mV. Note 9: Switch on resistance is calculated by dividing VIN to SW voltage by the forced current (1.5A). See Typical Performance Characteristics for the graph of switch voltage at other currents. Note 10: The CSS threshold is defined as the value of current sourced into the CSS pin which results in an increase in sink current from the VC pin. See the Soft-Start section in Applications Information. Note 11: The PGFB threshold is defined as the percentage of VREF voltage which causes the current source output of the CT pin to change from sinking (below threshold) to sourcing current (above threshold). When sourcing current, the voltage on the CT pin rises until it is clamped internally. When the clamp is activated, the output of the PG pin will be set to a high impedance state. When the CT clamp is inactive the PG pin will be set active low with a current sink capability of 200µA. 1977f 3 LT1977 TYPICAL PERFOR A CE CHARACTERISTICS FB Voltage 1.30 1.29 1.28 FREQUENCY (kHz) 1.27 VOLTAGE (V) 1.26 1.25 1.24 1.23 1.22 1.21 1.20 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 510 500 490 480 470 460 450 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 VOLTAGE (V) SHDN Pin Current 5.5 5.0 4.5 4.0 CURRENT (µA) TJ = 25°C CURRENT (µA) 3.5 3.0 2.5 2.0 1.5 1.0 0 0 10 30 40 20 SHDN VOLTAGE (V) 50 60 1977 G04 15 CURRENT (µA) Bias Sleep Current 200 180 160 140 1.20 1.18 1.16 1.14 1.12 1.10 1.08 1.06 1.04 1.02 50 25 0 75 TEMPERATURE (°C) 100 125 CURRENT (µA) 120 100 80 60 40 20 0 –50 –25 CURRENT (µA) VOLTAGE (V) 4 UW 1977 G01 1977 G07 Oscillator Frequency 550 540 530 520 SHDN Threshold 1.40 1.35 1.30 1.25 1.20 0.15 1.10 1.05 1.00 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 1977 G02 1977 G03 Shutdown Supply Current 25 200 180 20 VIN = 60V 160 140 120 100 80 60 5 VIN = 42V 0 –50 –25 VIN = 12V 100 125 40 20 Sleep Mode Supply Current VBIAS = 0V 10 VBIAS = 5V 50 25 0 75 TEMPERATURE (°C) 0 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 1977 G05 1977 G06 PGFB Threshold 250 PG Sink Current 200 150 100 50 1.00 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 0 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 1977 G08 1977 G09 1977f LT1977 TYPICAL PERFOR A CE CHARACTERISTICS Switch Peak Current Limit 3.5 50 45 PEAK SWITCH CURRENT (A) 3.0 CURRENT (µA) 40 35 30 25 20 15 10 5 1.5 –50 –25 75 –0 25 50 TEMPERATURE (°C) 100 125 0 0 0.2 0.6 0.8 0.4 FB VOLTAGE (V) 1.0 1.2 1977 G11 FOLDBACK PERCENTAGE (%) 2.5 2.0 Switch On Voltage (VCESAT) 500 450 400 SUPPLY CURRENT (µA) 350 VOLTAGE (mV) 100 75 50 25 0 INPUT VOLTAGE (V) 300 250 200 150 100 50 0 –0.1 0.1 0.3 TJ = – 50°C TJ = 125°C TJ = 25°C 0.5 0.7 0.9 1.1 LOAD CURRENT (A) Burst Mode Threshold vs Input Voltage 500 VOUT = 3.3V 450 L = 10µH = 100µF C 400 OUT LOAD CURRENT (mA) 350 ON TIME (ns) 350 Burst Mode EXIT (INCREASING LOAD) LOAD CURRENT = 0.5A BOOST CURRENT (mA) 300 250 200 150 100 50 0 Burst Mode ENTER (DECREASING LOAD) 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 INPUT VOLTAGE (V) 1977 G20 UW 1977 G10 Soft-Start Current Threshold vs FB Voltage 100 Frequency Foldback Percentage 90 80 70 60 50 40 30 20 10 0 0 0.25 0.75 0.5 FB PIN VOLTAGE (V) 1 1.25 1977 G12 TJ = 25°C SOFT-START DEFEATED Supply Current vs Input Voltage 150 125 8.0 Minimum Input Voltage 7.5 7.0 6.5 6.0 5.5 5.0 4.5 4.0 3.5 3.0 VOUT = 3.3V START-UP RUNNING VOUT = 5V START-UP RUNNING VOUT = 3.3V TA = 25°C 1.3 1.5 0 10 30 40 20 INPUT VOLTAGE (V) 50 60 1977 F05 0 0.25 0.75 0.5 LOAD CURRENT (A) 1 1.25 1977 G19 1977 G13 Minimum On Time 500 450 400 45 40 35 30 25 20 15 10 5 Boost Current vs Load Current 300 250 200 150 100 50 0 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 LOAD CURRENT = 1A 0 0 0.25 0.5 0.75 1 LOAD CURRENT (A) 1.25 1.5 1977 G21 1977 G22 1977f 5 LT1977 TYPICAL PERFOR A CE CHARACTERISTICS Burst Mode Operation Burst Mode Operation VOUT 20mV/DIV ISW 500mA/DIV VIN = 12V VOUT = 3.3V IQ = 100µA No Load 1A Step Response VOUT 50mV/DIV IOUT 500mA/DIV VIN = 12V VOUT = 3.3V COUT = 100µF IDC = 0mA PI FU CTIO S NC (Pins 1, 3, 5): No Connection. SW (Pin 2): The SW pin is the emitter of the on-chip power NPN switch. This pin is driven up to the input pin voltage during switch on time. Inductor current drives the SW pin negative during switch off time. Negative voltage is clamped with the external catch diode. Maximum negative switch voltage allowed is –0.8V. VIN (Pin 4): This is the collector of the on-chip power NPN switch. VIN powers the internal control circuitry when a voltage on the BIAS pin is not present. High di/dt edges occur on this pin during switch turn on and off. Keep the path short from the VIN pin through the input bypass capacitor, through the catch diode back to SW. All trace inductance on this path will create a voltage spike at switch off, adding to the VCE voltage across the internal NPN. BOOST (Pin 6): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. Without this added voltage, the typical switch voltage loss would be about 1.5V. The additional BOOST voltage allows the switch to saturate and its voltage loss approximates that of a 0.2Ω FET structure. CT (Pin 7): A capacitor on the CT pin determines the amount of delay time between the PGFB pin exceeding its threshold (VPGFB) and the PG pin set to a high impedance state. When the PGFB pin rises above VPGFB, current is sourced from the CT pin into the external capacitor. When the voltage on the external capacitor reaches an internal clamp (VCT), the PG pin becomes a high impedance node. The resultant PG delay time is given by t = CCT • VCT/ICT. If the 1977f 6 UW VOUT 20mV/DIV ISW 500mA/DIV 5ms/DIV 1977 G14 VIN = 12V VOUT = 3.3V IQ = 100µA 2µs/DIV 1977 G15 Step Response VOUT 50mV/DIV IOUT 500mA/DIV 500µs/DIV 1977 G17 VIN = 12V VOUT = 3.3V COUT = 100µF IDC = 350mA 500µs/DIV 1977 G18 U U U LT1977 PI FU CTIO S voltage on the PGFB pin drops below VPGFB, CCT will be discharged rapidly to 0V and PG will be active low with a 200µA sink capability. If the CT pin is clamped (Power Good condition) during normal operation and SHDN is taken low, the CT pin will be discharged and a delay period will occur when SHDN is returned high. See the Power Good section in Applications Information for details. GND (Pins 8, 17): The GND pin connection acts as the reference for the regulated output, so load regulation will suffer if the “ground” end of the load is not at the same voltage as the GND pin of the IC. This condition will occur when load current or other currents flow through metal paths between the GND pin and the load ground. Keep the path between the GND pin and the load ground short and use a ground plane when possible. The GND pin also acts as a heat sink and should be soldered (along with the exposed leadframe) to the copper ground plane to reduce thermal resistance (see Applications Information). CSS (Pin 9): A capacitor from the CSS pin to the regulated output voltage determines the output voltage ramp rate during start-up. When the current through the CSS capacitor exceeds the CSS threshold (ICSS), the voltage ramp of the output is limited. The CSS threshold is proportional to the FB voltage (see Typical Performance Characteristics) and is defeated for FB voltage greater than 0.9V (typical). See Soft-Start section in Applications Information for details. BIAS (Pin 10): The BIAS pin is used to improve efficiency when operating at higher input voltages and light load current. Connecting this pin to the regulated output voltage forces most of the internal circuitry to draw its operating current from the output voltage rather than the input supply. This architecture increases efficiency especially when the input voltage is much higher than the output. Minimum output voltage setting for this mode of operation is 3V. VC (Pin 11): The VC pin is the output of the error amplifier and the input of the peak switch current comparator. It is normally used for frequency compensation, but can also serve as a current clamp or control loop override. VC sits at about 0.45V for light loads and 2.2V at maximum load. During the sleep portion of Burst Mode operation, the VC pin is held at a voltage slightly below the burst threshold for better transient response. Driving the VC pin to ground will disable switching and place the IC into sleep mode. FB (Pin 12): The feedback pin is used to determine the output voltage using an external voltage divider from the output that generates 1.25V at the FB pin . When the FB pin drops below 0.9V, switching frequency is reduced, the SYNC function is disabled and output ramp rate control is enabled via the CSS pin. See the Feedback section in Applications Information for details. PGFB (PIN 13): The PGFB pin is the positive input to a comparator whose negative input is set at VPGFB. When PGFB is taken above VPGFB, current (ICSS) is sourced into the CT pin starting the PG delay period. When the voltage on the PGFB pin drops below VPGFB, the CT pin is rapidly discharged resetting the PG delay period. The PGFB voltage is typically generated by a resistive divider from the regulated output or input supply. See Power Good section in Applications Information for details. SYNC (Pin 14): The SYNC pin is used to synchronize the internal oscillator to an external signal. It is directly logic compatible and can be driven with any signal between 30% and 70% duty cycle. The synchronizing range is equal to maximum initial operating frequency up to 700kHz. When the voltage on the FB pin is below 0.9V the SYNC function is disabled. See the Synchronizing section in Applications Information for details. SHDN (Pin 15): The SHDN pin is used to turn off the regulator and to reduce input current to less than 1µA. The SHDN pin requires a voltage above 1.2V with a typical source current of 3µA to take the IC out of the shutdown state. PG (Pin 16): The PG pin is functional only when the SHDN pin is above its threshold, and is active low when the internal clamp on the CT pin is below its clamp level and high impedance when the clamp is active. The PG pin has a typical sink capability of 200µA. See the Power Good section in Applications Information for details. U U U 1977f 7 LT1977 BLOCK DIAGRA 4 VIN 10 BIAS 14 15 SYNC SHDN ANTISLOPE COMP R S + SHDN COMP – 1.3V CSS BURST MODE DETECT 9 12 FB 1.25V 11 13 VC PG PGFB 16 1.12V 7 CT The LT1977 is a constant frequency, current mode buck converter. This means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. In addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. A switch cycle starts with an oscillator pulse which sets the RS latch to turn the switch on. When switch current reaches a level set by the current comparator the latch is reset and the switch turns off. Output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. This technique means that the error amplifier commands current to be 8 W INTERNAL REF UNDERVOLTAGE LOCKOUT THERMAL SHUTDOWN 2.4V SLOPE COMP Σ 500kHz OSCILLATOR + CURRENT COMP – BOOST 6 SWITCH Q LATCH DRIVER CIRCUITRY SW 2 SOFT-START FOLDBACK DETECT VC CLAMP + ERROR AMP – + PG COMP 1.2V CT CLAMP GND 17 PGND 8 – 1977 BD Figure 1. LT1977 Block Diagram delivered to the output rather than voltage. A voltage fed system will have low phase shift up to the resonant frequency of the inductor and output capacitor, then an abrupt 180° shift will occur. The current fed system will have 90° phase shift at a much lower frequency, but will not have the additional 90° shift until well beyond the LC resonant frequency. This makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response. Most of the circuitry of the LT1977 operates from an internal 2.4V bias line. The bias regulator normally draws 1977f LT1977 BLOCK DIAGRA power from the VIN pin, but if the BIAS pin is connected to an external voltage higher than 3V bias power will be drawn from the external source (typically the regulated output voltage). This improves efficiency. High switch efficiency is achieved by using the BOOST pin to provide a voltage to the switch driver which is higher than the input voltage, allowing switch to be saturated. This boosted voltage is generated with an external capacitor and diode. APPLICATIO S I FOR ATIO CHOOSING THE LT1976 OR LT1977 The LT1976 and LT1977 are both high voltage 1.5A stepdown Burst Mode switching regulators with a typical quiescent current of 100µA. The difference between the two is that the fixed switching frequency of the LT1976 is 200kHz versus 500kHz for the LT1977. The switching frequency affects: inductor size, input voltage range in continuous mode operation, efficiency, thermal loss and EMI. OUTPUT RIPPLE AND INDUCTOR SIZE Output ripple current is determined by the input to output voltage ratio, inductor value and switch frequency. Since the switch frequency of the LT1977 is 2.5 times greater than that of the LT1976, the inductance used in the LT1977 application can be 2.5 times lower than the LT1976 while maintaining the same output ripple current. The lower value used in the LT1977 application allows the use of a physically smaller inductor. INPUT VOLTAGE RANGE The LT1976 and LT1977 minimum on and off times are equivalent. This results in a narrower range of continuous mode operation for the LT1977. Typical minimum and maximum duty cycles are 6% to 95% for the LT1976 and 15% to 90% for the LT1977. Both parts will regulate up to an input voltage of 60V but the LT1977 will transistion into pulse skipping/Burst Mode operation when the input voltage is above 30V for a 5V output. At outputs above 10V U W W To further optimize efficiency, the LT1977 automatically switches to Burst Mode operation in light load situations. In Burst Mode operation, all circuitry associated with controlling the output switch is shut down reducing the input supply current to 45µA. The LT1977 contains a power good flag with a programmable threshold and delay time. A logic-level low on the SHDN pin disables the IC and reduces input suppy current to less than 1µA. the LT1977’s input range will be similar to the LT1976. Lowering the input voltage below the maximum duty cycle limitation will cause a dropout in regulation. Table 1. LT1976/LT1977 Comparison PARAMETER Minimum Duty Cycle Maximum Duty Cycle Inductor Size Output Capacitor Size Efficiency EMI Input Range Output Ripple ADVANTAGE LT1976 LT1976 LT1977 LT1977 LT1976 LT1976 LT1976 LT1977 UU FEEDBACK PIN FUNCTIONS The feedback (FB) pin on the LT1977 is used to set output voltage and provide several overload protection features. The first part of this section deals with selecting resistors to set output voltage and the remaining part talks about frequency foldback and soft-start features. Please read both parts before committing to a final design. Referring to Figure 2, the output voltage is determined by a voltage divider from VOUT to ground which generates 1.25V at the FB pin. Since the output divider is a load on the output care must be taken when choosing the resistor divider values. For light load applications the resistor values should be as large as possible to achieve peak efficiency in Burst Mode operation. Extremely large values for resistor R1 will cause an output voltage error due to the 1977f 9 LT1977 APPLICATIO S I FOR ATIO LT1977 SW 2 SOFT-START CSS 9 500kHz OSCILLATOR FOLDBACK DETECT + ERROR AMP FB 12 R2 – 1.25V VC 11 1977 F02 Figure 2. Feedback Network Table 2 OUTPUT VOLTAGE (V) 2.5 3 3.3 5 6 8 10 12 R2 (kΩ, 1%) 100 100 100 100 100 100 100 100 R1 NEAREST (1%) (kΩ) 100 140 165 300 383 536 698 866 OUTPUT ERROR (%) 0 0 0.38 0 0.63 – 0.63 – 0.25 0.63 50nA FB pin input current. The suggested value for the output divider resistor (see Figure 2) from FB to ground (R2) is 100k or less. A formula for R1 is shown below. A table of standard 1% values is shown in Table 2 for common output voltages. R1 = R2 • VOUT – 1.25 1.25 + R2 • 50nA More Than Just Voltage Feedback The FB pin is used for more than just output voltage sensing. It also reduces switching frequency and controls the soft-start voltage ramp rate when output voltage is below the regulated level (see the Frequency Foldback 10 U VOUT C1 W UU and Soft-Start Current graphs in Typical Performance Characteristics). Frequency foldback is used to control power dissipation in both the IC and in the external diode and inductor during short-circuit conditions. A shorted output requires the switching regulator to operate at very low duty cycles. As a result the average current through the diode and inductor is equal to the short-circuit current limit of the switch (typically 2A for the LT1977). Minimum switch on time limitations would prevent the switcher from operating at a sufficiently low duty cycle if switching frequency were maintained at 500kHz, so frequency is reduced by about 4:1 when the FB pin voltage drops below 0.4V (see Frequency Foldback graph). In addition, if the current in the switch exceeds 1.5 times the current limitations specified by the VC pin, due to minimum switch on time, the LT1977 will skip the next switch cycle. As the feedback voltage rises, the switching frequency increases to 500kHz with 0.95V on the FB pin. During frequency foldback, external synchronization is disabled to prevent interference with foldback operation. Frequency foldback does not affect operation during normal load conditions. In addition to lowering switching frequency the soft-start ramp rate is also affected by the feedback voltage. Large capacitive loads or high input voltages can cause a high input current surge during start-up. The soft-start function reduces input current surge by regulating switch current via the VC pin to maintain a constant voltage ramp rate (dV/dt) at the output. A capacitor (C1 in Figure 2) from the CSS pin to the output determines the maximum output dV/dt. When the feedback voltage is below 0.4V, the VC pin will rise, resulting in an increase in switch current and output voltage. If the dV/dt of the output causes the current through the CSS capacitor to exceed ICSS the VC voltage is reduced resulting in a constant dV/dt at the output. As the feedback voltage increases ICSS increases, resulting in an increased dV/dt until the soft-start function is defeated with 0.9V present at the FB pin. The soft-start function does not affect operation during normal load conditions. However, if a momentary short (brown out condition) is present at the output which causes the FB voltage to drop below 0.9V, the soft-start circuitry will become active. 1977f R1 LT1977 APPLICATIO S I FOR ATIO INPUT CAPACITOR Step-down regulators draw current from the input supply in pulses. The rise and fall times of these pulses are very fast. The input capacitor is required to reduce the voltage ripple this causes at the input of LT1977 and force the switching current into a tight local loop, thereby minimizing EMI. The RMS ripple current can be calculated from: IRIPPLE(RMS) = IOUT VOUT ( VIN – VOUT ) VIN Ceramic capacitors are ideal for input bypassing. At 500kHz switching frequency input capacitor values in the range of 4.7µF to 20µF are suitable for most applications. If operation is required close to the minimum input required by the LT1977 a larger value may be required. This is to prevent excessive ripple causing dips below the minimum operating voltage resulting in erratic operation. Input voltage transients caused by input voltage steps or by hot plugging the LT1977 to a pre-powered source such as a wall adapter can exceed maximum VIN ratings. The sudden application of input voltage will cause a large surge of current in the input leads that will store energy in the parasitic inductance of the leads. This energy will cause the input voltage to swing above the DC level of input power source and it may exceed the maximum voltage rating of the input capacitor and LT1977. All input voltage transient sequences should be observed at the VIN pin of the LT1977 to ensure that absolute maximum voltage ratings are not violated. The easiest way to suppress input voltage transients is to add a small aluminum electrolytic capacitor in parallel with the low ESR input capacitor. The selected capacitor needs to have the right amount of ESR to critically damp the resonant circuit formed by the input lead inductance and the input capacitor. The typical values of ESR will fall in the range of 0.5Ω to 2Ω and capacitance will fall in the range of 5µF to 50µF. If tantalum capacitors are used, values in the 22µF to 470µF range are generally needed to minimize ESR and meet ripple current and surge ratings. Care should be taken to ensure the ripple and surge ratings are not U exceeded. The AVX TPS and Kemet T495 series are surge rated. AVX recommends derating capacitor operating voltage by 2:1 for high surge applications. OUTPUT CAPACITOR The output capacitor is normally chosen by its effective series resistance (ESR) because this is what determines output ripple voltage. To get low ESR takes volume, so physically smaller capacitors have higher ESR. The ESR range for typical LT1977 applications is 0.05Ω to 0.2Ω. A typical output capacitor is an AVX type TPS, 100µF at 10V, with a guaranteed ESR less than 0.1Ω. This is a “D” size surface mount solid tantalum capacitor. TPS capacitors are specially constructed and tested for low ESR, so they give the lowest ESR for a given volume. The value in microfarads is not particularly critical and values from 22µF to greater than 500µF work well, but you cannot cheat Mother Nature on ESR. If you find a tiny 22µF solid tantalum capacitor, it will have high ESR and output ripple voltage could be unacceptable. Table 3 shows some typical solid tantalum surface mount capacitors. Table 3. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current E CASE SIZE AVX TPS D CASE SIZE AVX TPS C CASE SIZE AVX TPS 0.2 0.5 0.1 to 0.3 0.7 to 1.1 ESR MAX (Ω) 0.1 to 0.3 RIPPLE CURRENT (A) 0.7 to 1.1 W UU Many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents. This is historically true and type TPS capacitors are specially tested for surge capability but surge ruggedness is not a critical issue with the output capacitor. Solid tantalum capacitors fail during very high turn-on surges which do not occur at the output of regulators. High discharge surges, such as when the regulator output is dead shorted, do not harm the capacitors. Unlike the input capacitor RMS, ripple current in the output capacitor is normally low enough that ripple current rating is not an issue. The current waveform is 1977f 11 LT1977 APPLICATIO S I FOR ATIO Output capacitor ripple current (RMS) IRIPPLE(RMS) = triangular with a typical value of 200mARMS. The formula to calculate this is: 0.29( VOUT )( VIN – VOUT ) IP-P = 12 (L)( f)( VIN ) CERAMIC CAPACITORS Higher value, lower cost ceramic capacitors are now becoming available. They are generally chosen for their good high frequency operation, small size and very low ESR (effective series resistance). Low ESR reduces output ripple voltage but also removes a useful zero in the loop frequency response, common to tantalum capacitors. To compensate for this a resistor RC can be placed in series with the VC compensation capacitor CC (Figure 10). Care must be taken however since this resistor sets the high frequency gain of the error amplifier including the gain at the switching frequency. If the gain of the error amplifier is high enough at the switching frequency output ripple voltage (although smaller for a ceramic output capacitor) may still affect the proper operation of the regulator. A filter capacitor CF in parallel with the RC/CC network, along with a small feedforward capacitor CFB, is suggested to control possible ripple at the VC pin. The LT1977 can be stabilized using a 100µF ceramic output capacitor and VC component values of CC = 1500pF, RC = 10k, CF = 330pF and CFB = 10pF. OUTPUT RIPPLE VOLTAGE Figure 3 shows a typical output ripple voltage waveform for the LT1977. Ripple voltage is determined by the impedance of the output capacitor and ripple current through the inductor. Peak-to-peak ripple current through the inductor into the output capacitor is: IP-P = VOUT ( VIN – VOUT ) ( VIN )(L)( f) For high frequency switchers the ripple current slew rate is also relevant and can be calculated from: di VIN = dt L 12 U VOUT 10mV/DIV 100µF 75mΩ TANTALUM VOUT 10mV/DIV 100µF CERAMIC VSW 10V/DIV VIN = 12V VOUT = 3.3V IL= 1A 500ns/DIV 1977 F03 W UU Figure 3. LT1977 Ripple Voltage Waveform Peak-to-peak output ripple voltage is the sum of a triwave created by peak-to-peak ripple current times ESR and a square wave created by parasitic inductance (ESL) and ripple current slew rate. Capacitive reactance is assumed to be small compared to ESR or ESL. VRIPPLE = (IP-P )(ESR) + (ESL) di dt Example: with VIN = 12V, VOUT = 3.3V, L = 15µH, ESR = 0.08Ω, ESL = 10nH: IP-P = (3.3)(12 – 3.3) = 0.319A (12)(15e − 6)(500e3) di 12 = = 0.8e6 dt 15e – 6 VRIPPLE = (0.319A)(0.08) + (10e – 9)(0.8e6) = 0.026 + 0.008 = 34mVP-P MAXIMUM OUTPUT LOAD CURRENT Maximum load current for a buck converter is limited by the maximum switch current rating (IPK). The minimum specified current rating for the LT1977 is 1.5A. Unlike most current mode converters, the LT1977 maximum switch current limit does not fall off at high duty cycles. Most current mode converters suffer a drop off of peak switch current for duty cycles above 50%. This is due to the effects of slope compensation required to prevent subharmonic oscillations in current mode converters. (For detailed analysis, see Application Note 19.) 1977f LT1977 APPLICATIO S I FOR ATIO The LT1977 is able to maintain peak switch current limit over the full duty cycle range by using patented circuitry to cancel the effects of slope compensation on peak switch current without affecting the frequency compensation it provides. Maximum load current would be equal to maximum switch current for an infinitely large inductor, but with finite inductor size, maximum load current is reduced by one-half peak-to-peak inductor current. The following formula assumes continuous mode operation, implying that the term on the right (IP-P/2) is less than IOUT. ( VOUT )( VIN – VOUT ) = I – IP-P IOUT(MAX) = IPK – PK 2(L)( f)( VIN ) 2 Discontinuous operation occurs when: IOUT(DIS) ≤ VOUT ( VIN – VOUT ) 2(L)( f)( VIN ) For VOUT = 5V, VIN = 8V and L = 15µH: IOUT(MAX) = 1.5 – (5)(8 – 5) 2(15e – 6)(500e3)(8) = 1.5 – 0.125 = 1.375A Note that there is less load current available at the higher input voltage because inductor ripple current increases. At VIN = 15V, duty cycle is 33% and for the same set of conditions: (5)(15 – 5) IOUT(MAX) = 1.5 – 2(15e – 6)(500e3)(15) = 1.5 – 0.22 = 1.28 A To calculate actual peak switch current in continuous mode with a given set of conditions, use: ISW (PK) = IOUT + VOUT VIN – VOUT 2 L f VIN ( )( )( ) ( ) If a small inductor is chosen which results in discontinous mode operation over the entire load range, the maximum load current is equal to: U IPK2 2( f)(L)( VIN ) IOUT(MAX) = 2( VOUT )( VIN – VOUT ) CHOOSING THE INDUCTOR For most applications the output inductor will fall in the range of 5µH to 33µH. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output current because they reduce peak current seen by the LT1977 switch, which has a 1.5A limit. Higher values also reduce output ripple voltage and reduce core loss. When choosing an inductor you might have to consider maximum load current, core and copper losses, allowable component height, output voltage ripple, EMI, fault current in the inductor, saturation and of course cost. The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. Choose a value in microhenries such that the maximum load current plus half the ripple current is less than the minimum peak switch current (IPK). Choosing a small inductor with lighter loads may result in discontinuous mode of operation, but the LT1977 is designed to work well in either mode. Assume that the average inductor current is equal to load current and decide whether or not the inductor must withstand continuous fault conditions. If maximum load current is 0.5A, for instance, a 0.5A inductor may not survive a continuous 2A overload condition. For applications with a duty cycle above 50%, the inductor value should be chosen to obtain an inductor ripple current of less than 40% of the peak switch current. 2. Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t omit this step. Powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core materials fall somewhere in between. The following formula assumes continuous mode of operation, but it errs only slightly on the high side for discontinuous mode, so it can be used for all conditions. 1977f W UU 13 LT1977 APPLICATIO S I FOR ATIO Table 4. Inductor Selection Criteria VENDOR/ PART NO. Coiltronics UP1B-100 UP1B-220 UP2B-220 UP2B-330 UP1B-150 Coilcraft D01813P-153HC D01813P-103HC D53316P-223 D53316P-333 LP025060B-682 Sumida CDRH4D28-4R7 CDRH5D28-100 CDRH6D28-150 CDRH6D28-180 CDRH6D28-220 CDRH6D38-220 4.7 10 15 18 22 22 1.32 1.30 1.40 1.32 1.20 1.30 0.072 0.065 0.084 0.095 0.128 0.096 15 10 22 33 6.8 1.5 1.9 1.6 1.4 1.3 0.170 0.111 0.207 0.334 0.165 10 22 22 33 15 1.9 1.2 2.0 1.7 1.5 0.111 0.254 0.062 0.092 0.175 VALUE (µH) IDC(MAX) (Amps) DCR (Ohms) IPEAK = IOUT + VOUT ( VIN – VOUT ) 2( f)(L)( VIN ) VIN = maximum input voltage f = switching frequency, 500kHz 3. Decide if the design can tolerate an “open” core geometry like a rod or barrel, which have high magnetic field radiation, or whether it needs a closed core like a toroid to prevent EMI problems. This is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. 4. After making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. Use the experts in the Linear Technology’s applications department if you feel uncertain about the final choice. They have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. where: 14 U Short-Circuit Considerations HEIGHT (mm) 5.0 5.0 6.0 6.0 5.0 5.0 5.0 5.1 5.1 1.65 3.0 3.0 3.0 3.0 3.0 4.0 W UU The LT1977 is a current mode controller. It uses the VC node voltage as an input to a current comparator which turns off the output switch on a cycle-by-cycle basis as this peak current is reached. The internal clamp on the VC node, nominally 2.2V, then acts as an output switch peak current limit. This action becomes the switch current limit specification. The maximum available output power is then determined by the maximum specified switch current limit. A potential control problem could occur under shortcircuit conditions. If the power supply output is short circuited, the feedback amplifier responds to the low output voltage by raising the control voltage, VC, to its peak current limit value. Ideally, the output switch would be turned on, and then turned off as its current exceeded the value indicated by VC. However, there is finite response time involved in both the current comparator and turn-off of the output switch. These result in a typical minimum on time of 300ns (see Typical Performance Characteristics). When combined with the large ratio of VIN to (VF + I • R), the diode forward voltage plus inductor I • R voltage drop, the potential exists for a loss of control. Expressed mathematically the requirement to maintain control is: f • tON ≤ VF + I • R VIN f = switching frequency tON = switch on time VF = diode forward voltage VIN = Input voltage I • R = inductor I • R voltage drop If this condition is not observed, the current will not be limited at IPK but will cycle-by-cycle ratchet up to some higher value. Using the nominal LT1977 clock frequency of 500kHz, a VIN of 12V and a (VF + I • R) of say 0.7V, the maximum tON to maintain control would be approximately 116ns, an unacceptably short time. The solution to this dilemma is to slow down the oscillator to allow the current in the inductor to drop to a sufficiently 1977f LT1977 APPLICATIO S I FOR ATIO low value such that the current doesn’t continue to ratchet higher. When the FB pin voltage is abnormally low thereby indicating some sort of short-circuit condition, the oscillator frequency will be reduced. Oscillator frequency is reduced by a factor of 4 when the FB pin voltage is below 0.4V and increases linearly to its typical value of 500kHz at a FB voltage of 0.95V (see Typical Performance Characteristics). In addition, if the current in the switch exceeds 1.5 • IPK current demanded by the VC pin, the LT1977 will skip the next on cycle effectively reducing the oscillator frequency by a factor of 2. These oscillator frequency reductions during short-circuit conditions allow the LT1977 to maintain current control. SOFT-START For applications where [VIN/(VOUT + VF)] ratios > 10 or large input surge currents can’t be tolerated, the LT1977 soft-start feature should be used to control the output capacitor charge rate during start-up, or during recovery from an output short circuit thereby adding additional control over peak inductor current. The soft-start function limits the switch current via the VC pin to maintain a constant voltage ramp rate (dV/dt) at the output capacitor. A capacitor (C1 in Figure 2) from the CSS pin to the regulated output voltage determines the output voltage ramp rate. When the current through the CSS capacitor exceeds the CSS threshold (ICSS), the voltage ramp of the output capacitor is limited by reducing the VC pin voltage. The CSS threshold is proportional to the FB voltage (see Typical Performance Characteristics) and is defeated for FB voltages greater than 0.9V (typical). The output dV/dt can be approximated by: dV ICSS = dt CSS but actual values will vary due to start-up load conditions, compensation values and output capacitor selection. Burst Mode OPERATION To enhance efficiency at light loads, the LT1977 automatically switches to Burst Mode operation (see Typical Performance Characteristics) which keeps the output U CCSS = 1000pF VOUT 1V/DIV CCSS = 0.01µF CCSS = 0.1µF VIN = 12V VOUT = 3.3V 1ms/DIV 1977 F04 W UU Figure 4. VOUT dV/dt capacitor charged to the proper voltage while minimizing the input quiescent current. During Burst Mode operation, the LT1977 delivers short bursts of current to the output capacitor followed by sleep periods where the output power is delivered to the load by the output capacitor. In addition, VIN and BIAS quiescent currents are reduced to typically 45µA and 125µA respectively during the sleep time. As the load current decreases towards a no load condition, the percentage of time that the LT1977 operates in sleep mode increases and the average input current is greatly reduced resulting in higher efficiency. The minimum average input current depends on the VIN to VOUT ratio, VC frequency compensation, feedback divider network and Schottky diode leakage. It can be approximated by the following equation: ⎛ V ⎞ (IBIASS + IFB + IS ) IIN(AVG) ≅ IVINS + ISHDN + ⎜ OUT ⎟ ⎝ VIN ⎠ ( η) where: IVINS = input pin current in sleep mode VOUT = output voltage VIN = input voltage IBIASS = BIAS pin current in sleep mode IFB = feedback network current IS = catch diode reverse leakage at VOUT η = low current efficiency (non Burst Mode operation) 1977f 15 LT1977 APPLICATIO S I FOR ATIO Example: For VOUT = 3.3V, VIN = 12V ⎛ 3.3 ⎞ (125µA + 12.5µA + 0.5µA ) IIN(AVG) = 45µA + 5µA + ⎜ ⎟ ⎝ 12 ⎠ (0.8) = 45µA + 5µA + 44µA = 99µA 150 125 SUPPLY CURRENT (µA) 100 75 50 25 0 0 10 30 40 20 INPUT VOLTAGE (V) 50 60 1977 F05 VOUT = 3.3V TA = 25°C ISW 500mA/DIV Figure 5. IQ vs VIN During the sleep portion of the Burst Mode cycle, the VC pin voltage is held just below the level needed for normal operation to improve transient response. See the Typical Performance Characteristics section for burst and transient response waveforms. If a no load condition can be anticipated, the supply current can be further reduced by cycling the SHDN pin at a rate higher than the natural no load burst frequency. Figure 6 shows Burst Mode operation with the SHDN pin. VOUT burst ripple is maintained while the average supply current drops to 15µA. The PG pin will be active low during the “on” portion of the SHDN waveform due to the CT capacitor discharge when SHDN is taken low. See the Power Good section for further information. CATCH DIODE The catch diode carries load current during the SW off time. The average diode current is therefore dependent on the switch duty cycle. At high input to output voltage ratios the diode conducts most of the time. As the ratio approaches unity the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short circuited. Under this condition the diode must safely handle IPEAK at maximum duty cycle. 16 U VOUT 50mV/DIV VSHDN 2V/DIV VIN = 12V VOUT = 3.3V IQ = 15µA TIME (50ms/DIV) 1977 G16 W UU Figure 6. Burst Mode with Shutdown Pin To maximize high and low load current efficiency a fast switching diode with low forward drop and low reverse leakage should be used. Low reverse leakage is critical to maximize low current efficiency since its value over temperature can potentially exceed the magnitude of the LT1977 supply current. Low forward drop is critical for high current efficiency since the loss is proportional to forward drop. These requirements result in the use of a Schottky type diode. DC switching losses are minimized due to its low forward voltage drop and AC behavior is benign due to its lack of a significant reverse recovery time. Schottky diodes are generally available with reverse voltage ratings of 60V and even 100V and are price competitive with other types. The effect of reverse leakage and forward drop on efficiency for various Schottky diodes is shown in Table 5. As can be seen these are conflicting parameters and the user Table 5. Catch Diode Selection Criteria LEAKAGE VOUT = 3.3V DIODE IR 10BQ100 Diodes Inc. B260SMA Diodes Inc. B360SMB IR MBRS360TR IR 30BQ100 25°C 0.0µA 0.1µA 0.2µA 1µA 0.5µA 125°C 59µA 242µA 440µA 1.81mA 225µA IQ at 125°C EFFICIENCY VIN =12V VIN =12V VOUT = 3.3 VOUT = 3.3V I L = 0A 125µA 215µA 270µA 821µA 206µA I L = 1A 76.1% 80.4% 80.8% 81.4% 78.8% 1977f VF AT 1A 25°C 0.72V 0.48V 0.45V 0.42V 0.59V LT1977 APPLICATIO S I FOR ATIO must weigh the importance of each specification in choosing the best diode for the application. The use of so-called “ultrafast” recovery diodes is generally not recommended. When operating in continuous mode, the reverse recovery time exhibited by “ultrafast” diodes will result in a slingshot type effect. The power internal switch will ramp up VIN current into the diode in an attempt to get it to recover. Then, when the diode has finally turned off, some tens of nanoseconds later, the VSW node voltage ramps up at an extremely high dV/dt, perhaps 5V to even 10V/ns! With real world lead inductances the VSW node can easily overshoot the VIN rail. This can result in poor RFI behavior and, if the overshoot is severe enough, damage the IC itself. BOOST PIN For most applications the boost components are a 0.1µF capacitor and a MMSD914 diode. The anode is typically connected to the regulated output voltage to generate a voltage approximately VOUT above VIN to drive the output stage (Figure 7a). However, the output stage discharges the boost capacitor during the on time of the switch. The output driver requires at least 2.5V of headroom throughout this period to keep the switch fully saturated. If the output voltage is less than 3.3V it is recommended that an alternate boost supply is used. The boost diode can be connected to the input (Figure 7b) but care must be taken to prevent the boost voltage (VBOOST = VIN • 2) from exceeding the BOOST pin absolute maximum rating. The additional voltage across the switch driver also increases power loss and reduces efficiency. If available, an independent supply can be used to generate the required BOOST voltage (Figure 7c). Tying BOOST to VIN or an independent supply may reduce efficiency but it will reduce the minimum VIN required to start-up with light loads. If the generated BOOST voltage dissipates too much power at maximum load, the BOOST voltage the LT1977 sees can be reduced by placing a Zener diode in series with the BOOST diode (Figure 7a option). A 0.1µF boost capacitor is recommended for most applications. Almost any type of film or ceramic capacitor is suitable but the ESR should be 4V with 100ms Delay VIN 200k PG LT1977 PGFB 511k 200k 165k FB CT 0.27µF 100k VOUT = 3.3V COUT W UU VOUT Disconnect 3.3V Logic Signal with 100µs Delay VIN 200k PG LT1977 COUT PGFB 866k FB CT 270pF 1977 F10 VOUT = 12V 100k 19 LT1977 APPLICATIO S I FOR ATIO electrical, thermal and noise performance. For maximum efficiency switch rise and fall times are typically in the nanosecond range. To prevent noise both radiated and conducted the high speed switching current path, shown in Figure 11, must be kept as short as possible. This is implemented in the suggested layout of Figure 12. Shortening this path will also reduce the parasitic trace inductance of approximately 25nH/inch. At switch off, this parasitic inductance produces a flyback spike across the LT1977 switch. When operating at higher currents and input voltages, with poor layout, this spike can generate voltages across the LT1977 that may exceed its absolute maximum rating. A ground plane should always be used under the switcher circuitry to prevent interplane coupling and overall noise. LT1977 4 VIN VIN C2 SW 2 L1 VOUT + HIGH FREQUENCY CIRCULATION PATH D1 C1 LOAD 1977 F11 Figure 11. High Speed Switching Path C2 VOUT C1 L1 D2 CONNECT PIN 8 GND TO THE PIN 17 EXPOSED PAD GND PLACE VIA's UNDER EXPOSED PAD TO A BOTTOM PLANE TO ENHANCE THERMAL CONDUCTIVITY KELVIN SENSE FEEDBACK TRACE AND KEEP SEPARATE FROM BIAS TRACE R3 D1 MINIMIZE D1-C3 LOOP GND 1 NC 2 SW 3 NC PGOOD 16 SHDN 15 LT1977 SYNC 14 PGFB 13 FB 12 VC 11 BIAS 10 CSS 9 C5 R1 R2 C2 C3 VIN 4 VIN 5 NC 6 BOOST 7 TCAP C4 8 GND GND Figure 12. Suggested Layout 20 U The VC and FB components should be kept as far away as possible from the switch and boost nodes. The LT1977 pinout has been designed to aid in this. The ground for these components should be separated from the switch current path. Failure to do so will result in poor stability or subharmonic like oscillation. Board layout also has a significant effect on thermal resistance. Pin 8 and the exposed die pad, Pin 17, are a continuous copper plate that runs under the LT1977 die. This is the best thermal path for heat out of the package. Reducing the thermal resistance from Pin 8 and exposed pad onto the board will reduce die temperature and increase the power capability of the LT1977. This is achieved by providing as much copper area as possible around the exposed pad. Adding multiple solder filled feedthroughs under and around this pad to an internal ground plane will also help. Similar treatment to the catch diode and coil terminations will reduce any additional heating effects. THERMAL CALCULATIONS Power dissipation in the LT1977 chip comes from four sources: switch DC loss, switch AC loss, boost circuit current, and input quiescent current. The following formulas show how to calculate each of these losses. These formulas assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. Switch loss: 2 W UU PSW RSW (IOUT ) ( VOUT ) = + tEFF (1/2)(IOUT )( VIN )( f) VIN Boost current loss: PBOOST ( VOUT )2 (IOUT /32) = VIN Quiescent current loss: PQ = VIN (0.0015) + VOUT (0.003) RSW = switch resistance (≈ 0.3 when hot ) tEFF = effective switch current/voltage overlap time (tr + tf + tIR + tIF) 1977f 1977 F12 LT1977 APPLICATIO S I FOR ATIO tr = (VIN/1.1)ns tf = (VIN/1.8)ns tIR = tIF = (IOUT/0.05)ns f = switch frequency Example: with VIN = 12V, VOUT = 5V and IOUT = 1A: PSW 12 0.04 + TBD = 0.125 + 0.172 = 0.297W (0.3)(1)2 (5) + = (57.6e−9 )(1/2)(1)(12)(500e3) PBOOST 12 PQ = 12(0.0015) + 5(0.003) = 0.033W (5)2 (1/32) = 0.002W = Total power dissipation is: PTOT = 0.297 + 0.065 + 0.033 = 0.40W Thermal resistance for the LT1977 package is influenced by the presence of internal or backside planes. With a full plane under the FE16 package, thermal resistance will be about 45°C/W. No plane will increase resistance to about 150°C/W. To calculate die temperature, use the proper thermal resistance number for the desired package and add in worst-case ambient temperature: TJ = TA + QJA (PTOT) With the FE16 package (QJA = 45°C/W) at an ambient temperature of 70°C: TJ = 70 + 45(0.40) = 98°C Input Voltage vs Operating Frequency Considerations The absolute maximum input supply voltage for the LT1977 is specified at 60V. This is based solely on internal semiconductor junction breakdown effects. Due to internal power dissipation the actual maximum VIN achievable in a particular application may be less than this. A detailed theoretical basis for estimating internal power loss is given in the section Thermal Considerations. Note that AC switching loss is proportional to both operating frequency and output current. The majority of AC switching loss is also proportional to the square of input voltage. U For example, while the combination of VIN = 40V, VOUT = 5V at 1A and fOSC = 500kHz may be easily achievable, simultaneously raising VIN to 60V and fOSC to 700kHz is not possible. Nevertheless, input voltage transients up to 60V can usually be accommodated, assuming the resulting increase in internal dissipation is of insufficient time duration to raise die temperature significantly. A second consideration is control. A potential limitation occurs with a high step-down ratio of VIN to VOUT, as this requires a correspondingly narrow minimum switch on time. An approximate expression for this (assuming continuous mode operation) is given as follows: tON(MIN) = VOUT + VF/VIN(fOSC) where: VIN = input voltage VOUT = output voltage VF = Schottky diode forward drop fOSC = switching frequency A potential control problem arises if the LT1977 is called upon to produce an on time shorter than it is able to produce. Feedback loop action will lower then reduce the VC control voltage to the point where some sort of cycleskipping or Burst Mode behavior is exhibited. In summary: 1. Be aware that the simultaneous requirements of high VIN, high IOUT and high fOSC may not be achievable in practice due to internal dissipation. The Thermal Considerations section offers a basis to estimate internal power. In questionable cases a prototype supply should be built and exercised to verify acceptable operation. 2. The simultaneous requirements of high VIN, low VOUT and high fOSC can result in an unacceptably short minimum switch on time. Cycle skipping and/or Burst Mode behavior will result causing an increase in output voltage ripple while maintaining the correct output voltage. FREQUENCY COMPENSATION Before starting on the theoretical analysis of frequency response the following should be remembered—the worse 1977f W UU 21 LT1977 APPLICATIO S I FOR ATIO the board layout, the more difficult the circuit will be to stabilize. This is true of almost all high frequency analog circuits. Read the Layout Considerations section first. Common layout errors that appear as stability problems are distant placement of input decoupling capacitor and/or catch diode and connecting the VC compensation to a ground track carrying significant switch current. In addition the theoretical analysis considers only first order nonideal component behavior. For these reasons, it is important that a final stability check is made with production layout and components. The LT1977 uses current mode control. This alleviates many of the phase shift problems associated with the inductor. The basic regulator loop is shown in Figure 12. The LT1977 can be considered as two gm blocks, the error amplifier and the power stage. Figure 13 shows the overall loop response with a 330pF VC capacitor and a typical 100µF tantalum output capacitor. The response is set by the following terms: Error amplifier: DC gain is set by gm and RO: EA Gain = 650µ • 1.5M = 975 The pole set by CF and RL: EA Pole = 1/(2π • 1.5M • 330pF) = 322Hz Unity gain frequency is set by CF and gm: EA Unity Gain Frequency = 650µ /(2π • 330pF) = 313kHz Powerstage: DC gain is set by gm and RL (assume 10Ω): PS DC Gain = 3 • 10 = 30 Pole set by COUT and RL: GAIN (dB) CC 11 RC CF gm = 650µ VC 1.5M ERROR AMP Figure 13. Model for Loop Response 100 VOUT = 3.3V COUT = 100µF, 0.1Ω CF = 330pF RC/CC = NC ILOAD = 350mA 100 PS Pole = 1/(2π • 100µF • 10) = 159Hz Unity gain set by COUT and gm: PS Unity Gain Freq = 3/(2π • 100µF) = 4.7kHz. Tantalum output capacitor zero is set by COUT and COUT ESR 50 0 45 –50 Output Capacitor Zero = 1/(2π • 100µF • 0.1) = 159kHz The zero produced by the ESR of the tantalum output capacitor is very useful in maintaining stability. If better 10 100 1k 10k FREQUENCY (Hz) Figure 14. Overall Loop Response 1977f 22 Ω Ω U transient response is required, a zero can be added to the loop using a resistor (RC) in series with a compensation capacitor(s). As the value of RC is increased, transient response will generally improve but two effects limit its value. First, the combination of output capacitor ESR and a large RC may stop loop gain rolling off altogether. Second, if the loop gain is not rolled off sufficiently at the switching frequency output ripple will perturb the VC pin enough to cause unstable duty cycle switching similar to subharmonic oscillation. This may not be apparent at the output. Smallsignal analysis will not show this since a continuous time system is assumed. If needed, an additional capacitor (CF) can be added to form a pole at below the switching frequency (if RC = 26k, CC = 1500pF, CF = 330pF). When checking loop stability the circuit should be operated over the application’s full voltage, current and temperature range. Any transient loads should be applied and the output voltage monitored for a well-damped behavior. LT1977 CURRENT MODE POWER STAGE gm = 3 SW 2 R1 CFB OUTPUT W Ω UU Ω + – FB 12 R2 ESR 1.25V COUT 1977 F13 135 PHASE (DEG) 90 100k 0 1M 1977 F14 LT1977 PACKAGE DESCRIPTIO 3.58 (.141) 6.60 ± 0.10 4.50 ± 0.10 SEE NOTE 4 0.65 BSC RECOMMENDED SOLDER PAD LAYOUT 4.30 – 4.50* (.169 – .177) 0.09 – 0.20 (.0035 – .0079) 0.50 – 0.75 (.020 – .030) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U FE Package 16-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663) Exposed Pad Variation BC 4.90 – 5.10* (.193 – .201) 3.58 (.141) 16 1514 13 12 1110 9 2.94 (.116) 0.45 ± 0.05 1.05 ± 0.10 6.40 2.94 (.252) (.116) BSC 12345678 1.10 (.0433) MAX 0° – 8° 0.25 REF 0.65 (.0256) BSC 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE16 (BC) TSSOP 0204 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 1977f 23 LT1977 RELATED PARTS PART NUMBER LT1074/LT1074HV LT1076/LT1076HV LT1676 LT1765 LT1766 LT1767 LT1776 LTC®1875 LT1940 LT1956 LT1976 LT3010 LTC3407 LTC3412 LTC3414 LT3430 LT3431 LT3433 DESCRIPTION 4.4A (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converters 1.6A (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converters 60V, 440mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter 25V, 3A (IOUT), 1.25MHz, High Efficiency Step-Down DC/DC Converter 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 25V, 1.5A (IOUT), 1.25MHz, High Efficiency Step-Down DC/DC Converter 40V, 550mA (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 1.5A (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter Dual 1.2A (IOUT), 1.1MHz, High Efficiency Step-Down DC/DC Converter 60V, 1.2A (IOUT), 500kHz, High Efficiency Step-Down DC/DC Converter 60V, 1.5A (IOUT), 200kHz High Efficiency Step-Down DC/DC Converter 80V, 50mA, Low Noise Linear Regulator Dual 600mA (IOUT), 1.5MHz, High Efficiency Step-Down DC/DC Converter 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 4A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 60V, 2.5A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 60V, 2.5A (IOUT), 500kHz, High Efficiency Step-Down DC/DC Converter 60V, 400mA (IOUT), 200kHz, Buck-Boost DC/DC Converter COMMENTS VIN: 7.3V to 45V/64V, VOUT(MIN) = 2.21V, IQ = 8.5mA, ISD < 10µA, DD5/7, TO220-5/7 VIN: 7.3V to 45V/64V, VOUT(MIN) = 2.21V, IQ = 8.5mA, ISD < 10µA, DD5/7, TO220-5/7 VIN: 7.4V to 60V, VOUT(MIN) = 1.24V, IQ = 3.2mA, ISD < 2.5µA, S8 VIN: 3V to 25V, VOUT(MIN) = 1.20V, IQ = 1mA, ISD < 15µA, SO-8, TSSOP16/E VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD < 25µA, TSSOP16E VIN: 3V to 25V, VOUT(MIN) = 1.20V, IQ = 1mA, ISD < 6µA, MS8E VIN: 7.4V to 40V, VOUT(MIN) = 1.24V, IQ = 3.2mA, ISD < 30µA, N8, S8 VIN: 2.7V to 6V, VOUT(MIN) = 0.8V, IQ = 15µA, ISD < 1µA, TSSOP16 VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 3.8mA, MS10 VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD < 25µA, TSSOP16E VIN: 3.3V to 60V, IQ = 100µA, ISD < 1µA, TSSOP16E VIN: 1.5V to 80V, VOUT(MIN) = 1.28V, IQ = 30µA, ISD < 1µA, MS8E VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA, MS10 VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA, ISD < 1µA, TSSOP16E VIN: 2.25V to 5.5V, VOUT(MIN) = 0.8V, IQ = 64µA, ISD < 1µA, TSSOP20E VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD < 30µA, TSSOP16E VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD < 30µA, TSSOP16E VIN: 5V to 60V, VOUT: 3.3V to 20V, IQ = 100µA, TSSOP16E VIN: 4V to 36V, VOUT(MIN) = 0.8V, IQ = 670µA, ISD < 20µA, QFN-32, SSOP-28 LTC3727/LTC3727-1 36V, 500kHz, High Efficiency Step-Down DC/DC Controllers 1977f 24 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● LT/TP 0604 1K PRINTED IN USA www.linear.com © LINEAR TECHNOLOGY CORPORATION 2004