3506AIFE

LT3506/LT3506A Dual Monolithic 1.6A Step-Down Switching Regulator ■ ■ ■ ■ ■ ■ ■ ■ ■ Wide Input Voltage Range, 3.6V to 25V Two 1.6A Output Switching Regulators with Internal Power Switches Constant Switching Frequency LT3506: 575kHz LT3506A: 1.1MHz Anti-Phase Switching Reduces Ripple Accurate 0.8V Reference, ±1% Independent Shutdown/Soft-Start Pins Independent Power Good Indicators Ease Supply Sequencing Uses Small Inductors and Ceramic Capacitors Small 16-Lead Thermally Enhanced 5mm × 4mm DFN and TSSOP Surface Mount Packages The LT®3506 is a dual current mode PWM step-down DC/DC converter with internal 2A power switches. Both converters are synchronized to a single oscillator and run with opposite phases, reducing input ripple current. The output voltages are set with external resistor dividers, and each regulator has independent shutdown and soft-start circuits. Each regulator generates a power-good signal when its output is in regulation, easing power supply sequencing and interfacing with microcontrollers and DSPs. The LT3506 switching frequency is 575kHz and the LT3506A is 1.1MHz. These high switching frequencies allow the use of tiny inductors and capacitors, resulting in a very small dual 1.6A output solution. Constant frequency and ceramic capacitors combine to produce low, predictable output ripple voltage. With its wide input range of 3.6V to 25V, the LT3506 regulates a wide variety of power sources, from 4-cell batteries and 5V logic rails to unregulated wall transformers, lead acid batteries and distributed-power supplies. Current mode PWM architecture provides fast transient response with simple compensation components and cycle-by-cycle current limiting. Frequency foldback and thermal shutdown provide additional protection. , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. APPLICATIO S ■ ■ ■ ■ ■ ■ Disk Drives DSP Power Supplies Wall Transformer Regulation Distributed Power Regulation DSL Modems Cable Modems VIN 4.5V TO 25V 22µF VIN1 VIN2 BOOST1 VOUT1 1.8V 1.6A 4.7µH 18.7k 1000pF 0.22µF SW1 FB1 VC1 47µF 15k D1 15k LT3506 RUN/SS1 RUN/SS2 100k 100k PGOOD1 PGOOD2 1.5nF PGOOD1 PGOOD2 GND 1.5nF SW2 FB2 VC2 10k BOOST2 0.22µF EFFICIENCY (%) U TYPICAL APPLICATIO 1/2 BAT-54A 1/2 BAT-54A 6.4µH 33.2k VOUT2 3.3V 1.6A 2200pF D2 10.7k 22µF 3506 F01 D1, D2: ON SEMI MBR5230LT3 U Efficiency 100 VIN = 5V VOUT = 3.3V VOUT = 1.8V 90 80 70 60 50 0 0.5 1.0 IOUT (A) 1.5 2.0 3506 TA01b FEATURES DESCRIPTIO U 3506afb  LT3506/LT3506A Absolute MAxiMuM RAtings (Note 1) VIN Voltage ................................................. –0.3V to 25V BOOST Pin Voltage ................................................... 50V BOOST Pin Above SW Pin......................................... 25V PG Pin Voltage .......................................................... 25V RUN/SS, FB, VC Pins ................................................ 5.5V Maximum Junction Temperature........................... 125°C Operating Temperature Range (Note 2) E Grade ................................................ –40°C to 85°C I Grade ............................................... –40°C to 125°C Storage Temperature Range................... –65°C to 125°C TOP VIEW BOOST1 SW1 VIN1 VIN1 VIN2 VIN2 SW2 BOOST2 1 2 3 4 5 6 7 8 17 16 FB1 15 VC1 14 PG1 13 RUN/SS1 12 RUN/SS2 11 PG2 10 VC2 9 FB2 BOOST1 SW1 VIN1 VIN1 VIN2 VIN2 SW2 BOOST2 1 2 3 4 5 6 7 8 DHD PACKAGE 16-LEAD PLASTIC DFN TJMAX = 125°C, θJA = 43°C/W, θJC = 4.3°C/W EXPOSED PAD (PIN 17) IS GND MUST BE SOLDERED TO PCB LEAD FREE FINISH LT3506EDHD#PBF LT3506AEDHD#PBF LT3506IDHD#PBF LT3506AIDHD#PBF LT3506EFE#PBF LT3506AEFE#PBF LT3506IFE#PBF LT3506AIFE#PBF TAPE AND REEL LT3506EDHD#TRPBF LT3506AEDHD#TRPBF LT3506IDHD#TRPBF LT3506AIDHD#TRPBF LT3506EFE#TRPBF LT3506AEFE#TRPBF LT3506IFE#TRPBF LT3506AIFE#TRPBF LEAD BASED FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3506EDHD LT3506EDHD#TR 3506 16-Lead (5mm x 4mm) Plastic DFN –40°C to 85°C LT3506AEDHD LT3506AEDHD#TR 3506A 16-Lead (5mm x 4mm) Plastic DFN –40°C to 85°C LT3506IDHD LT3506IDHD#TR 3506 16-Lead (5mm x 4mm) Plastic DFN –40°C to 125°C LT3506AIDHD LT3506AIDHD#TR 3506A 16-Lead (5mm x 4mm) Plastic DFN –40°C to 125°C LT3506EFE LT3506EFE#TR 3506EFE 16-Lead Plastic TSSOP Narrow –40°C to 85°C LT3506AEFE LT3506AEFE#TR 3506AEFE 16-Lead Plastic TSSOP Narrow –40°C to 85°C LT3506IFE LT3506IFE#TR 3506IFE 16-Lead Plastic TSSOP Narrow –40°C to 125°C LT3506AIFE LT3506AIFE#TR 3506AIFE 16-Lead Plastic TSSOP Narrow –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3506afb  U ORDER I FOR ATIO W U U TOP VIEW 16 FB1 15 VC1 14 PG1 17 13 RUN/SS1 12 RUN/SS2 11 PG2 10 VC2 9 FB2 FE PACKAGE 16-LEAD PLASTIC TSSOP NARROW TJMAX = 125°C, θJA = 45°C/W, θJC = 10°C/W EXPOSED PAD (PIN 17) IS GND MUST BE SOLDERED TO PCB PI CO FIGURATIO U U PART MARKING* 3506 3506A 3506 3506A 3506EFE 3506AEFE 3506IFE 3506AIFE PACKAGE DESCRIPTION 16-Lead (5mm x 4mm) Plastic DFN 16-Lead (5mm x 4mm) Plastic DFN 16-Lead (5mm x 4mm) Plastic DFN 16-Lead (5mm x 4mm) Plastic DFN 16-Lead Plastic TSSOP Narrow 16-Lead Plastic TSSOP Narrow 16-Lead Plastic TSSOP Narrow 16-Lead Plastic TSSOP Narrow TEMPERATURE RANGE –40°C to 85°C –40°C to 85°C –40°C to 125°C –40°C to 125°C –40°C to 85°C –40°C to 85°C –40°C to 125°C –40°C to 125°C LT3506/LT3506A ELECTRICAL CHARACTERISTICS SYMBOL VIN(MIN) IINQ IINSD VFB IFB VFB(REG) gmEA AV IVC VVC(THRESH) VVC(CLAMP) fSW PARAMETER Undervoltage Lockout Quiescent Current Shutdown Current Feedback Voltage Not Switching VRUNSS = 0V –40°C to 85°C, EDHD –40°C to 85°C, EFE –40°C to 125°C, IFE, IDHD VFB = 800mV, VC = 0.4V VIN = 5V to 25V ● ● ● ● The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, VBOOST = 8V, unless otherwise noted. (Note 2) CONDITIONS ● MIN TYP 3.4 3.8 30 MAX 3.6 4.8 45 808 816 816 100 UNITS V mA µA mV mV mV nA %/V uMhos µA µA V V 792 784 784 800 800 800 40 0.005 350 400 FB Pin Bias Current Reference Line Regulation Error Amp GM Error Amp Voltage Gain VC Source Current VC Sink Current VC Switching Threshold VC Clamp Voltage Switching Frequency Switching Phase VFB = 0.6V, VC = 0V VFB = 1.2V, VC = 1100mV 30 30 0.7 1.9 LT3506 LT3506A (Note 5) LT3506 LT3506A VFB = 0V (Note 3) ISW = 1A ISW = 1A ISW = 1A 500 1 89 78 575 1.1 180 93 88 0.4 170 650 1.2 kHz MHz Deg % % V kHz DC VFB(SWTHRESH) fFOLD ISW VSW(SAT) ILSW VBOOST(MIN) IBOOST IRUN/SS VRUN/SS(THRESH) VFB(PGTHRESH) VPG(LOW) ILPG Maximum Duty Cycle Frequency Shift Threshold on FB Foldback Frequency Switch Current Limit Switch VCESAT (Note 4) Switch Leakage Current Minimum Boost Voltage Above Switch BOOST Pin Current RUN/SS Current RUN/SS Threshold VFB PG Threshold PG Voltage Output Low PG Pin Leakage 2.0 2.6 210 3.6 10 A mV µA V mA µA V mV 1.5 20 2.1 0.3 0.8 720 0.22 0.1 2.5 30 VFB Rising VFB = 640mV, IPG = 250µA VPG = 2V 0.4 1 V µA Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3506E/LT3506AE are guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LT3506I/LT3506AI are guaranteed and tested over the full –40°C to 125°C operating temperature range. Note 3: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at high duty cycle. Note 4: Switch VCESAT guaranteed by design. Note 5: Switching phase is guaranteed by design. 3506afb  LT3506/LT3506A TYPICAL PERFOR A CE CHARACTERISTICS 100 Efficiency, VOUT = 1.8V (LT3506A) VOUT = 1.8V L = 4.7µH (COILCRAFT MSS6122-472MLB) 90 TA = 25°C 80 EFFICIENCY (%) 70 60 50 40 VIN = 4.5V VIN = 12V VIN = 25V 0 0.2 0.4 0.6 0.8 1.0 1.2 OUTPUT CURRENT (A) 1.4 1.6 EFFICIENCY (%) 80 75 70 65 60 55 50 0 0.4 0.8 IOUT (A) 1.2 1.6 3506 G02 EFFICIENCY (%) 30 Maximum Load Current, VOUT = 1.8V (LT3506A) 1.8 TA = 25°C L = 2.2µH LOAD CURRENT (A) 1.8 L = 1.5µH 1.4 L = 1µH 1.2 L = 3.3µH 1.4 L = 2.2µH SWITCH VOLTAGE (mV) 1.6 LOAD CURRENT (A) 1.0 0 2 4 8 10 12 6 INPUT VOLTAGE (V)* Boost Pin Current 40 TA = 25°C 3.0 2.5 BOOST CURRENT (mA) 30 CURRENT LIMIT (A) 2.0 20 MINIMUM 1.5 1.0 0.5 FREQUENCY (kHz) 10 0 0 1.0 1.5 0.5 SWITCH CURRENT (A)  UW 3506 G01 100 Efficiency, VOUT = 3.3V (LT3506) 100 Efficiency, VOUT = 5V (LT3506) VOUT = 5V 95 L = 10µH (COOPER UP1B-100) TA = 25°C 90 85 80 75 70 65 60 55 50 0 0.4 0.8 IOUT (A) 1.2 1.6 3506 G03 VOUT = 3.3V 95 L = 6.4µH (SUMIDA CR54-6R4) TA = 25°C 90 85 VIN = 5V VIN = 8V VIN = 12V VIN = 15V VIN = 25V VIN = 25V Maximum Load Current, VOUT = 3.3V (LT3506A) SLOPE COMPENSATION REQUIRES L > 2.2µH FOR VIN < 7 WITH VOUT = 3.3V TA = 25°C L = 4.7µH 400 Switch VCESAT TA = 25°C 1.6 300 200 1.2 100 14 16 1.0 0 5 15 20 10 INPUT VOLTAGE (V)* 25 3506 G05 0 0 0.5 1.0 SW CURRENT (A) 1.5 2.0 3506 G06 3506 G04 Current Limit vs Duty Cycle TA = 25°C TYPICAL 650 700 Frequency vs Temperature 1.20 1.15 FREQUENCY (MHz) LT3506A 600 LT3506 550 1.10 1.05 2.0 3506 G07 0 0 20 60 40 DUTY CYCLE (%) 80 100 3506 G08 500 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 1.00 125 3506 G10 3506afb LT3506/LT3506A TYPICAL PERFOR A CE CHARACTERISTICS IRUN/SS vs Temperature 3.0 2.5 RUNN/SS THRESHOLDS (V) RUN/SS CURRENT (µA) 2.0 1.5 1.0 0.5 0 –50 1.4 1.2 1.0 0.8 0.6 TO RUN 0.4 0.2 –25 0 25 50 75 TEMPERATURE (°C) 100 125 0 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 125 TO SWITCH PI FU CTIO S BOOST1 (Pin 1), BOOST2 (Pin 8): The BOOST pins are used to provide drive voltages, higher than the input voltage, to the internal bipolar NPN power switches. Tie through a diode from VOUT or from VIN. SW1 (Pin 2), SW2 (Pin 7): The SW pins are the outputs of the internal power switches. Connect these pins to the inductors, catch diodes and boost capacitors. VIN1 (Pins 3, 4): The VIN1 pins supply current to the LT3506’s internal regulator and to the internal power switch connected to SW1. These pins must be locally bypassed. VIN2 (Pins 5, 6): The VIN2 pins supply current to the internal power switch connected to SW2 and must be locally bypassed. Connect these pins directly to VIN1 unless power for Channel 2 is coming from a different source. RUN/SS1 (Pin 13), RUN/SS2 (Pin 12): The RUN/SS pins are used to shut down the individual switching regulators and the internal bias circuits. They also provide a soft-start function. To shut down either regulator, pull the RUN/SS pin to ground with an open drain or collector. Tie a capacitor from these pins to ground to limit switch current during start-up. If neither feature is used, leave these pins unconnected. UW RUN/SS Thresholds vs Temperature 3506 G12 3506 G13 U U U PG1 (Pin 14), PG2 (Pin 11): The Power Good pins are the open collector outputs of an internal comparator. PG remains low until the FB pin is within 10% of the final regulation voltage. As well as indicating output regulation, the PG pins can be used to sequence the two switching regulators. These pins can be left unconnected. The PG outputs are valid when VIN is greater than 3.4V and either of the RUN/SS pins is high. The PG comparators are disabled in shutdown. VC1 (Pin 15), VC2 (Pin 10): The VC pins are the outputs of the internal error amps. The voltages on these pins control the peak switch currents. These pins are normally used to compensate the control loops, but can also be used to override the loops. Pull these pins to ground with an open drain to shut down each switching regulator. FB1 (Pin 16), FB2 (Pin 9): The LT3506 regulates each feedback pin to 800mV. Connect the feedback resistor divider taps to these pins. Exposed Pad (Pin 17): The Exposed Pad of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The Exposed Pad must be soldered to the circuit board for proper operation. 3506afb  LT3506/LT3506A W VIN RUN/SS2 2µA INT REG AND REF MASTER OSC CLK1 CLK2 VIN VIN CIN 0.75V C1 R S FOLDBACK LOGIC Q C3 SW L1 OUT D1 C1 BOOST D2 FB R1 R2 ERROR AMP ILIMIT CLAMP 80mV BLOCK DIAGRA RUN/SS1 2µA ∑ SLOPE CLK VC CF CC RUN/SS PG GND Figure 2. Block Diagram of the LT3506 with Associated External Components (1 of 2 Regulators Shown)  – + + – RC – + 800mV 3506 F02 3506afb LT3506/LT3506A The LT3506 is a dual, constant frequency, current mode buck regulator with internal 2A power switches. The two regulators share common circuitry including voltage reference and oscillator. In addition, the analog blocks on both regulators share the VIN1 supply voltage, but are otherwise independent. This section describes the operation of the LT3506. If the RUN/SS (run/soft-start) pins are both tied to ground, the LT3506 is shut down and draws 30μA from VIN1. Internal 2μA current sources charge external soft-start capacitors, generating voltage ramps at these pins. If either RUN/SS pin exceeds 0.6V, the internal bias circuits turn on, including the internal regulator, 800mV reference and 575kHz master oscillator. In this state, the LT3506 draws 1.8mA from VIN1, whether one or both RUN/SS pins are high. Neither switching regulator will begin to operate until its RUN/SS pin reaches ~0.8V. The master oscillator generates two clock signals of opposite phase. The two switchers are current mode, step-down regulators. This means that instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. This current mode control improves loop dynamics and provides cycle-by-cycle current limit. The Block Diagram in Figure 2 shows only one of the two switching regulators. A pulse from the slave oscillator sets the RS flip-flop and turns on the internal NPN bipolar power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level determined by the voltage at VC, current comparator C1 resets the flip-flop, turning off the switch. The current in U OPERATIO (Refer to the Block Diagram) the inductor flows through the external Schottky diode, and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way the voltage on the VC pin controls the current through the inductor to the output. The internal error amplifier regulates the output voltage by continually adjusting the VC pin voltage. The threshold for switching on the VC pin is 0.75V, and an active clamp of 1.9V limits the output current. The VC pin is also clamped to the RUN/SS pin voltage. As the internal current source charges the external soft-start capacitor, the current limit increases slowly. Each switcher contains an independent oscillator. This slave oscillator is normally synchronized to the master oscillator. However, during start-up, short-circuit or overload conditions, the FB pin voltage will be near zero and an internal comparator gates the master oscillator clock signal. This allows the slave oscillator to run the regulator at a lower frequency. This frequency foldback behavior helps to limit switch current and power dissipation under fault conditions. The switch driver operates from either the input or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to fully saturate the internal bipolar NPN power switch for efficient operation. A power good comparator trips when the FB pin is at 90% of its regulated value. The PG output is an open collector transistor that is off when the output is in regulation, allowing an external resistor to pull the PG pin high. Power good is valid when the LT3506 is enabled (either RUN/SS pin is high) and VIN is greater than ~3.4V. 3506afb  LT3506/LT3506A FB Resistor Network The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the 1% resistors according to: R1 = R2(VOUT/0.8 – 1) The parallel combination of R1 and R2 should be 10k or less to avoid bias current errors. Reference designators refer to the Block Diagram in Figure 2. Input Voltage Range The minimum input voltage is determined by either the LT3506’s minimum operating voltage of ~3.6V, or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: DC = (VOUT + VD)/(VIN – VSW + VD) where VD is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.3V at maximum load). This leads to a minimum input voltage of: VIN(MIN) = (VOUT + VD)/DCMAX - VD + VSW with DCMAX = 0.89 (0.78 for the LT3506A). A more detailed analysis includes inductor loss and the dependence of the diode and switch drop on operating current. A common application where the maximum duty cycle limits the input voltage range is the conversion of 5V to 3.3V. The maximum load current that the LT3506 can deliver at 3.3V depends on the accuracy of the 5V input supply. With a low loss inductor (DCR less than 80mW), the LT3506 can deliver 1.2A for VIN > 4.7V and 1.6A for VIN > 4.85V. The maximum input voltage is determined by the absolute maximum ratings of the VIN and BOOST pins and by the minimum duty cycle DCMIN = 0.08 (0.15 for the LT3506A): VIN(MAX) = (VOUT + VD)/DCMIN – VD + VSW. This limits the maximum input voltage to ~21V with VOUT = 1.2V and ~15V with VOUT = 0.8V. For the LT3506A the  U maximum input voltage is ~8V with VOUT=0.8V. Note that this is a restriction on the operating input voltage; the circuit will tolerate transient inputs up to the absolute maximum rating. Inductor Selection and Maximum Output Current A good first choice for the inductor value is: L = 2 • (VOUT + VD) for the LT3506 L = (VOUT + VD) for the LT3506A where VD is the voltage drop of the catch diode (~0.4V) and L is in μH. With this value the maximum load current will be ~1.6A, independent of input voltage. The inductor’s RMS current rating must be greater than your maximum load current and its saturation current should be about 30% higher. To keep efficiency high, the series resistance (DCR) should be less than 0.1W. Table 1 lists several vendors and types that are suitable. Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger value provides a slightly higher maximum load current, and will reduce the output voltage ripple. If your load is lower than 1.6A, then you can decrease the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. There are several graphs in the Typical Performance Characteristics section of this data sheet that show the maximum load current as a function of input voltage and inductor value for several popular output voltages. Also, low inductance may result in discontinuous mode operation, which may be acceptable, but further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally, for duty cycles greater than 50%(VOUT/VIN < 0.5), there is a minimum inductance required to avoid subharmonic oscillations. See Application Note 19 for detailed information on subharmonic oscillations. The following discussion assumes continuous inductor current. 3506afb APPLICATIO S I FOR ATIO W U U LT3506/LT3506A The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-topeak inductor ripple current. The LT3506 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3506 will deliver depends on the current limit, the inductor value and the input and output voltages. L is chosen based on output current requirements, output voltage ripple requirements, size restrictions and efficiency goals. When the switch is off, the inductor sees the output voltage plus the catch diode drop. This gives the peak-topeak ripple current in the inductor: ΔIL = (1 – DC)(VOUT + VD)/(L • f) where f is the switching frequency of the LT3506 and L is the value of the inductor. The peak inductor and switch current is ISWPK = ILPK = IOUT + ΔIL/2. To maintain output regulation, this peak current must be less than the LT3506’s switch current limit ILIM. ILIM is at least 2A at low duty cycle and decreases linearly to 1.7A at DC = 0.8. The maximum output current is a function of the chosen inductor value: IOUT(MAX) = ILIM – ΔIL/2 = 2A • (1 – 0.21 • DC) – ΔIL/2 If the inductor value is chosen so that the ripple current is small, then the available output current will be near the switch current limit. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then use these equations to check that the LT3506 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT is less than ΔIL/2 as calculated above. U Table 1. Inductors Part Number Sumida CR43-3R3 CR43-4R7 CDC5d23-2R2 CDRH5D28-2R6 CDRH6D26-5R6 CDH113-100 Coilcraft DO1606T-152 DO1606T-222 DO1608C-332 DO1608C-472 DO1813P-682HC Cooper SD414-2R2 SD414-6R8 UP1B-100 Toko (D62F)847FY-2R4M (D73LF)817FY2R2M 2.4 2.2 2.5 2.7 0.037 0.03 2.7 3.0 2.2 6.8 10 2.73 1.64 1.90 0.061 0.135 0.111 1.35 1.35 5.0 1.5 2.2 3.3 4.7 6.8 2.10 1.70 2.00 1.50 2.20 0.060 0.070 0.080 0.090 0.080 2.0 2.0 2.9 2.9 5.0 3.3 4.7 2.2 2.6 5.6 10 1.44 1.15 2.16 2.60 2.00 2.00 0.086 0.109 0.030 0.013 0.027 0.047 3.5 3.5 2.5 3.0 2.8 3.7 Value (μH) ISAT (A) DCR (W) Height (mm) APPLICATIO S I FOR ATIO W U U Input Capacitor Selection Bypass the input of the LT3506 circuit with a 4.7μF or higher ceramic capacitor of X7R or X5R type. A lower value or a less expensive Y5V type can be used if there is additional bypassing provided by bulk electrolytic or tantalum capacitors. The following paragraphs describe the input capacitor considerations in more detail. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3506 and to force this very high frequency switching current into a tight local loop, minimizing EMI. The input capaci- 3506afb  LT3506/LT3506A tor must have low impedance at the switching frequency to do this effectively, and it must have an adequate ripple current rating. With two switchers operating at the same frequency but with different phases and duty cycles, calculating the input capacitor RMS current is not simple. However, a conservative value is the RMS input current for the channel that is delivering most power (VOUT • IOUT). This is given by: IINRMS = IOUT VOUT • ( VIN − VOUT ) VIN < IOUT 2 and is largest when VIN = 2VOUT (50% duty cycle). As the second, lower power channel draws input current, the input capacitor’s RMS current actually decreases as the out-of-phase current cancels the current drawn by the higher power channel. Considering that the maximum load current from a single channel is ~1.6A, RMS ripple current will always be less than 0.8A. The high frequency of the LT3506 reduces the energy storage requirements of the input capacitor, so that the capacitance required is less than 22μF (less than 10μF for the LT3506A). The combination of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes them the preferred choice. The low ESR results in very low voltage ripple and the capacitors can handle plenty of ripple current. They are also comparatively robust and can be used in this application at their rated voltage. X5R and X7R types are stable over temperature and applied voltage, and give dependable service. Other types (Y5V and Z5U) have very large temperature and voltage coefficients of capacitance, so they may have only a small fraction of their nominal capacitance in your application. While they will still handle the RMS ripple current, the input voltage ripple may become fairly large, and the ripple current may end up flowing from your input supply or from other bypass capacitors in your system, as opposed to being fully sourced from the local input capacitor. An alternative to a high value ceramic capacitor is a lower value along with a larger electrolytic capacitor, for example a 1μF ceramic capacitor in parallel with a low ESR tantalum capacitor. For the electrolytic capacitor, a value larger than 22mF (10mF for the LT3506A) will be required to meet the 0 U ESR and ripple current requirements. Because the input capacitor is likely to see high surge currents when the input source is applied, tantalum capacitors should be surge rated. The manufacturer may also recommend operation below the rated voltage of the capacitor. Be sure to place the 1μF ceramic as close as possible to the VIN and GND pins on the IC for optimal noise immunity. A final caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plugging the circuit into a live power source) this tank can ring, doubling the input voltage and damaging the LT3506. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor in parallel with the ceramic capacitor. For details, see Application Note 88. Output Capacitor Selection The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order satisfy transient loads and to stabilize the LT3506’s control loop. Because the LT3506 operates at a high frequency, you don’t need much output capacitance. Also, the current mode control loop doesn’t require the presence of output capacitor series resistance (ESR). For these reasons, you are free to use ceramic capacitors to achieve very low output ripple and small circuit size. Estimate output ripple with the following equations: VRIPPLE = ΔIL/(8 • f • COUT) for ceramic capacitors, and VRIPPLE = ΔIL • ESR for electrolytic capacitors (tantalum and aluminum); where ΔIL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low, and the RMS current rating of the output capacitor is usually not of concern. Another constraint on the output capacitor is that it must have greater energy storage than the inductor; if the stored energy in the inductor is transferred to the output, you would like the resulting voltage step to be small compared to the regulation voltage. For a 5% overshoot, this requirement becomes COUT > 10L(ILIM/VOUT)2. 3506afb APPLICATIO S I FOR ATIO W U U LT3506/LT3506A Finally, there must be enough capacitance for good transient performance. The last equation gives a good starting point. Alternatively, you can start with one of the designs in this data sheet and experiment to get the desired performance. This topic is covered more thoroughly in the section on loop compensation. For 5V and 3.3V outputs with greater than 1A output, a 22μF 6.3V ceramic capacitor (X5R or X7R) at the output results in very low output voltage ripple and good transient response. For lower voltages, 22μF is adequate but increasing COUT will improve transient performance. For the LT3506A, 10μF of output capacitance is sufficient at VOUT between 3.3V and 5V. Other types and values can be used. The following discusses tradeoffs in output ripple and transient performance. The high performance (low ESR), small size and robustness of ceramic capacitors make them the preferred type for LT3506 applications. However, all ceramic capacitors are not the same. As mentioned above, many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and temperature extremes. Because the loop stability and transient response depend on the value of COUT, you may not be able to tolerate this loss. Use X7R and X5R types. You can also use electrolytic capacitors. The ESRs of most aluminum electrolytics are too large to deliver low output ripple. Tantalum and newer, lower ESR organic electrolytic capacitors intended for power supply use are suitable, and the manufacturers will specify the ESR. The choice of capacitor value will be based on the ESR required for low ripple. Because the volume of the capacitor determines its ESR, both the size and the value will be larger than a ceramic capacitor that would give similar ripple performance. One benefit is that the larger capacitance may give better transient response for large changes in load current. Table 2 lists several capacitor vendors. U Table 2. Low-ESR Surface Mount Capacitors VENDOR Taiyo-Yuden AVX Kemet TYPE Ceramic Ceramic Tantalum Tantalum Tantalum Organic Aluminum Organic Tantalum or Aluminum Organic Aluminum Organic Ceramic TPS T491, T494, T495, T520 A700 SERIES Sanyo Panasonic TDK POSCAP SP CAP APPLICATIO S I FOR ATIO W U U Catch Diode The catch diode (D1 in Figure 2) must have a reverse voltage rating greater than the maximum input voltage. The average current of the catch diode is given by: IDAVE=IOUT(1-DCMIN) A Schottky diode with a 1A average forward current rating will suffice for most applications. The ON Semiconductor MBRM120LT3 (20V) and MBRM130LT3 (30V) are good choices; they have a tiny package with good thermal properties. Many vendors have suitable surface mount versions of the 1N5817 (20V) and 1N5818 (30V) 1A Schottky diodes such as the Microsemi UPS120. Applications with large step down ratios and high output currents may have more than 1A of average diode current. The ON Semiconductor MBRS230LT3 or International Rectifier 20BQ030 (both 2A, 30V) would be good choices. BOOST Pin Considerations The capacitor and diode tied to the BOOST pin generate a voltage that is higher than the input voltage. In most cases a 0.1μF capacitor and fast switching diode (such as the CMDSH-3 or FMMD914) will work well. Figure 3 3506afb  LT3506/LT3506A shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.5V above the SW pin for full efficiency. For outputs of 3.3V and higher the standard circuit (Figure 3a) is best. For outputs between 2.8V and 3.3V, use a small Schottky diode (such as the BAT-54). For lower output voltages the boost diode can be tied to the input (Figure 3b). The circuit in Figure 3a is more efficient because the BOOST pin current comes from a lower voltage source. Finally, as shown in Figure 3c, the anode of the boost diode can be tied to another source that is at least 3V. For example, if you are generating 3.3V and 1.8V and the 3.3V is on whenever the 1.8V is on, the 1.8V boost diode can be connected to the 3.3V output. In any case, you must also be sure that the maximum voltage at D2 BOOST LT3506 VIN VIN GND VBOOST – VSW ≅ VOUT MAX VBOOST ≅ VIN + VOUT SW VOUT VIN VIN GND VBOOST – VSW ≅ VIN MAX VBOOST ≅ 2VIN C3 (3a) D2 VINB > 3V BOOST LT3506 VIN VIN GND VBOOST – VSW ≅ VINB MAX VBOOST ≅ VINB + VIN MINIMUM VALUE FOR VINB = 3V SW VOUT VIN VIN GND C3 VINB >VIN + 3V D2 BOOST LT3506 (3c) Figure 3. Generating the Boost Voltage  U the BOOST pin is less than the maximum specified in the Absolute Maximum Ratings section. The boost circuit can also run directly from a DC voltage that is higher than the input voltage by more than 3V, as in Figure 3d. The diode is used to prevent damage to the LT3506 in case VINB is held low while VIN is present. The circuit saves several components (both BOOST pins can be tied to D2). However, efficiency may be lower and dissipation in the LT3506 may be higher. Also, if VINB is absent, the LT3506 will still attempt to regulate the output, but will do so with very low efficiency and high dissipation because the switch will not be able to saturate, dropping 1.5V to 2V in conduction. D2 BOOST LT3506 SW VOUT C3 APPLICATIO S I FOR ATIO W U U (3b) SW VOUT MAX VBOOST – VSW ≅ VINB MAX VBOOST ≅ VINB MINIMUM VALUE FOR VINB = VIN + 3V 3506 F03 (3d) 3506afb LT3506/LT3506A The minimum input voltage of an LT3506 application is limited by the minimum operating voltage (