LT1223

LT1223 100MHz Current Feedback Amplifier FEATURES s s s s s s s s s DESCRIPTIO 100MHz Bandwidth at AV = 1 1000V/µs Slew Rate Wide Supply Range: ±5V to ±15V 1mV Input Offset Voltage 1µA Input Bias Current 5MΩ Input Resistance 75ns Settling Time to 0.1% 50mA Output Current 6mA Quiescent Current The LT1223 is a 100MHz current feedback amplifier with very good DC characteristics. The LT1223’s high slew rate, 1000V/µs, wide supply range, ±15V, and large output drive, ±50mA, make it ideal for driving analog signals over double- terminated cables. The current feedback amplifier has high gain bandwidth at high gains, unlike conventional op amps. The LT1223 comes in the industry standard pinout and can upgrade the performance of many older products. The LT1223 is manufactured on Linear Technology’s proprietary complementary bipolar process. APPLICATI s s s s s S Video Amplifiers Buffers IF and RF Amplification Cable Drivers 8-, 10-, 12-Bit Data Acquisition Systems TYPICAL APPLICATI Video Cable Driver 60 V IN + LT1223 – RF 1k 75Ω CABLE 75Ω Voltage Gain vs Frequency 100MHz GAIN BANDWIDTH RG = 10 RG = 33 RG = 110 RG = 470 RG = ∞ + – 1k RG 50 40 VOLTAGE GAIN (dB) 30 20 10 0 VOUT RG 1k 75Ω –10 –20 100k 1M 10M FREQUENCY (Hz) LT1223 • TPC01 R AV = 1 + F RG AT AMPLIFIER OUTPUT. 6dB LESS AT VOUT . LT1223 • TA02 U 100M 1G UO UO 1 LT1223 ABSOLUTE AXI U RATI GS PACKAGE/ORDER I FOR ATIO TOP VIEW NULL –IN +IN V– 1 2 3 4 8 7 6 5 SHUTDOWN V+ OUT NULL Supply Voltage ...................................................... ±18V Differential Input Voltage ......................................... ±5V Input Voltage ............................ Equal to Supply Voltage Output Short Circuit Duration (Note 1) ......... Continuous Operating Temperature Range LT1223M ........................................ –55°C to 125°C LT1223C ................................................ 0°C to 70°C Storage Temperature Range ................. –65°C to 150°C Junction Temperature Plastic Package ........... 150°C Junction Temperature Ceramic Package ........ 175°C Lead Temperature (Soldering, 10 sec.)................. 300°C ORDER PART NUMBER LT1223MJ8 LT1223CJ8 LT1223CN8 LT1223CS8 S8 PART MARKING 1223 J8 PACKAGE N8 PACKAGE 8-LEAD CERAMIC DIP 8-LEAD PLASTIC DIP S8 PACKAGE 8-LEAD PLASTIC SOIC LT1223 • POI01 TJ MAX = 175°C, θJA = 100°C/W(J8) TJ MAX = 150°C, θJA = 100°C/W(N8) TJ MAX = 150°C, θJA = 150°C/W(S8) ELECTRICAL CHARACTERISTICS VS = ± 15V, TA = 25°C, unless otherwise noted. SYMBOL VOS IIN+ IIN– en in RIN CIN CMRR PSRR PARAMETER Input Offset Voltage Noninverting Input Current Inverting Input Current Input Noise Voltage Density Input Noise Current Density Input Resistance Input Capacitance Input Voltage Range Common-Mode Rejection Ratio Inverting Input Current Common-Mode Rejection Power Supply Rejection Ratio Noninverting Input Current Power Supply Rejection Inverting Input Current Power Supply Rejection AV ROL VOUT IOUT SR BW tr tPD ts Large Signal Voltage Gain Transresistance, ∆VOUT/∆IIN– Maximum Output Voltage Swing Maximum Output Current Slew Rate Bandwidth Rise Time Propagation Delay Overshoot Settling Time, 0.1% Differential Gain Differential Phase ROUT IS Open-Loop Output Resistance Supply Current Supply Current, Shutdown VCM = ±10V VCM = ±10V VS = ± 4.5V to ±18V VS = ± 4.5V to ±18V VS = ± 4.5V to ±18V RLOAD = 400Ω, VOUT = ± 10V RLOAD = 400Ω, VOUT = ± 10V RLOAD = 200Ω RLOAD = 200Ω RF = 1.5k, RG = 1.5k, (Note 2) RF = 1k, RG = 1k, VOUT = 100mV RF = 1.5k, RG = 1.5k, VOUT = 1V RF = 1.5k, RG = 1.5k, VOUT = 1V RF = 1.5k, RG = 1.5k, VOUT = 1V RF = 1k, RG = 1k, VOUT = 10V RF = 1k, RG = 1k, RL = 150Ω RF = 1k, RG = 1k, RL = 150Ω VOUT = 0, IOUT = 0 VIN = 0V Pin 8 Current = 200µA 70 1.5 ± 10 50 800 68 ± 10 56 CONDITIONS VCM = 0V VCM = 0V VCM = 0V f = 1kHz, RF = 1k, RG = 10Ω f = 1kHz, RF = 1k, RG = 10Ω VIN = ±10V 1 MIN LT1223M/C TYP ±1 ±1 ±1 3.3 2.2 10 1.5 ± 12 63 30 80 12 60 89 5 ± 12 60 1300 100 6.0 6.0 5 75 0.02 0.12 35 6 2 10 4 100 500 100 MAX ±3 ±3 ±3 UNITS mV µA µA nV/√Hz pA/√Hz MΩ pF V dB nA/V dB nA/V nA/V dB MΩ V mA V/µs MHz ns ns % ns % Deg Ω mA mA 2 U W U U WW W LT1223 ELECTRICAL CHARACTERISTICS VS = ± 15V, VCM = 0V, 0°C ≤ TA ≤ 70°C, unless otherwise noted. SYMBOL VOS IIN+ IIN– RIN CMRR PSRR PARAMETER Input Offset Voltage Noninverting Input Current Inverting Input Current Input Resistance Input Voltage Range Common-Mode Rejection Ratio Inverting Input Current Common-Mode Rejection Power Supply Rejection Ratio Noninverting Input Current Power Supply Rejection Inverting Input Current Power Supply Rejection AV ROL VOUT IOUT IS Large-Signal Voltage Gain Transresistance, ∆VOUT/∆IIN– Maximum Output Voltage Swing Maximum Output Current Supply Current Supply Current, Shutdown VCM = ±10V VCM = ±10V VS = ±4.5V to ±18V VS = ±4.5V to ±18V VS = ±4.5V to ±18V RLOAD = 400Ω, VOUT = ± 10V RLOAD = 400Ω, VOUT = ± 10V RLOAD = 200Ω RLOAD = 200Ω VIN = 0V Pin 8 Current = 200µA CONDITIONS VCM = 0V VCM = 0V VCM = 0V VIN = ±10V q q q q q q q q q q q q q q q q MIN ±1 ±1 ±1 1 ± 10 56 68 LT1223C TYP ±3 ±3 ±3 10 ±12 63 30 80 12 60 MAX mV µA µA UNITS MΩ V dB 100 100 500 nA/V dB nA/V nA/ V dB MΩ V mA 10 4 mA mA 70 1.5 ± 10 50 89 5 ± 12 60 6 2 ELECTRICAL CHARACTERISTICS VS = ± 15V, VCM = 0V, – 55°C ≤ TA ≤ 125°C, unless otherwise noted. SYMBOL VOS IIN+ IIN– RIN CMRR PSRR PARAMETER Input Offset Voltage Noninverting Input Current Inverting Input Current Input Resistance Input Voltage Range Common-Mode Rejection Ratio Inverting Input Current Common-Mode Rejection Power Supply Rejection Ratio Noninverting Input Current Power Supply Rejection Inverting Input Current Power Supply Rejection AV ROL VOUT IOUT IS Large-Signal Voltage Gain Transresistance, ∆VOUT/∆IIN– Maximum Output Voltage Swing Maximum Output Current Supply Current Supply Current, Shutdown The q denotes the specifications which apply over the full operating temperature range. Note 1: A heat sink may be required. Note 2: Noninverting operation, VOUT = ±10V, measured at ± 5V. VCM = ±10V VCM = ±10V VS = ±4.5V to ±15V VS = ±4.5V to ±15V VS = ±4.5V to ±15V RLOAD = 400Ω, VOUT = ± 10V RLOAD = 400Ω, VOUT = ± 10V RLOAD = 200Ω RLOAD = 200Ω VIN = 0V Pin 8 Current = 200µA CONDITIONS VCM = 0V VCM = 0V VCM = 0V VIN = ±10V q q q q q q q q q q q q q q q q MIN LT1223M TYP ±1 ±1 ±1 MAX ±5 ±5 ± 10 UNITS mV µA µA MΩ V dB 1 ± 10 56 68 10 ± 12 63 30 80 12 60 200 500 100 nA/V dB nA/V nA/V dB MΩ V mA 70 1.5 ±7 35 89 5 ± 12 60 6 2 10 4 mA mA 3 LT1223 TYPICAL PERFOR A CE CHARACTERISTICS Supply Current vs Supply Voltage, VIN = 0 (Operating) 10 125°C 8 SUPPLY CURRENT (mA) 4 PIN 8 = 0V OUTPUT SHORT CIRCUIT CURRENT (mA) 25°C 6 –55°C 4 SUPPLY CURRENT (mA) 2 0 0 2 4 6 8 10 12 14 16 18 20 LT1223 • TPC02 SUPPLY VOLTAGE (±V) Input Common-Mode Limit vs Temperature V+ –1 COMMON MODE RANGE (V) –2 –3 –4 +4 +3 +2 +1 V– –50 –25 0 25 50 VS = –15V VS = 5V V S = 15V –l B (µA) +lB (µA) VS = –5V 75 TEMPERATURE (°C) LT1223 • TPC05 VOS vs Common-Mode Voltage 20 15 10 5 VOS (mV) 20 V S = ±15V OUTPUT VOLTAGE SWING (V) OUTPUT VOLTAGE SWING (V) 125°C 0 –5 –10 –15 –20 –15 –10 –5 0 5 10 15 –55°C 25°C COMMON MODE VOLTAGE (V) LT1223 • TPC08 4 UW 100 125 Supply Current vs Supply Voltage (Shutdown) 100 90 80 70 60 50 40 30 20 10 Output Short Circuit-Current vs Temperature 3 125°C 2 25°C –55°C 1 0 0 2 4 6 8 10 12 14 16 18 20 LT1223 • TPC03 0 –50 –25 0 25 50 75 100 125 SUPPLY VOLTAGE (±V) CASE TEMPERATURE (°C) LT1223 • TPC04 +IB vs Common-Mode Voltage 5 4 3 2 1 0 –1 –2 –3 –4 –5 –15 –10 –5 0 5 10 15 25°C –55°C VS = ±15V –IB vs Common-Mode Voltage 10 8 125°C 6 4 2 0 –2 –4 –6 –8 –10 –15 –10 –5 0 5 10 15 25°C –55°C V S = ±15V 125°C COMMON MODE VOLTAGE (V) LT1223 • TPC06 COMMON MODE VOLTAGE (V) LT1223 • TPC07 Output Voltage Swing vs Load Resistor 20 VS = ±15V 125°C 25°C, –55°C 15 15 10 5 0 –5 –10 –15 –20 100 1000 LOAD RESISTOR (Ω) LT1223 • TPC09 Output Voltage Swing vs Supply Voltage 125°C 10 5 0 –5 –10 –15 –20 25°C 25°C –55°C 25°C, –55°C 125°C 10000 125°C –55°C 0 2 4 6 8 10 12 14 16 18 20 LT1223 • TPC10 SUPPLY VOLTAGE (±V) LT1223 TYPICAL PERFOR A CE CHARACTERISTICS –3dB Bandwidth vs Feedback Resistor 100 90 A V = 2; RF = RG R L = 100 Ω ; VS = ±15V NO CAPACITIVE LOAD 100 90 70 60 50 40 30 20 10 0 0 1 2 3 LT1223 • TPC11 70 60 50 40 30 20 10 0 0 FEEDBACK RESISTOR (Ω) –3dB BANDWIDTH (MHz) 80 –3dB BANDWIDTH (MHz) FEEDBACK RESISTOR (k Ω) Maximum Capacitive Load vs Feedback Resistor 10k 100 OPEN LOOP VOLTAGE GAIN (dB) TRANSIMPEDANCE (MΩ) A V = 2; RF = RG R L = 100; VS = ± 15V PEAKING < 5dB CAPACITIVE LOAD (pF) 1k 100 10 0 1 2 3 LT1223 • TPC14 FEEDBACK RESISTOR (kΩ) Spot Noise Voltage and Current vs Frequency 1000 80 POWER SUPPLY REJECTION (dB) SPOT NOISE (nV/√Hz OR pA/√Hz) VS = ±15V RF = 1k 60 POSITIVE MAGNITUDE OF OUTPUT IMPEDANCE (Ω) 100 –i n 10 en +i n 1 10 100 1k 10k LT1223 • TPC17 FREQUENCY (Hz) UW –3dB Bandwidth vs Supply Voltage 1000 RF = RG AV = 2 RL = 100 Ω TA = 25°C RF = 750 RF = 1k Minimum Feedback Resistor vs Voltage Gain 900 800 700 600 500 400 300 200 100 2dB PEAKING 0dB PEAKING VS = ±15V R L = 100 80 RF = 1.5k RF = 2k 5 10 15 LT1223 • TPC12 0 10 20 30 40 50 60 SUPPLY VOLTAGE (± V) VOLTAGE GAIN (V/V) LT1223 • TPC13 Open-Loop Voltage Gain vs Load Resistor 25°C 10 9 –55°C 80 70 60 50 40 100 VS = ±15V VO = ± 10V 1000 LOAD RESISTOR (Ω) LT1223 • TPC15 Transimpedance vs Load Resistor VS = ± 15V VO = ± 10V 90 8 7 6 5 4 3 2 1 25°C 125°C –55°C 125°C 10000 0 100 1000 LOAD RESISTOR (Ω) 10000 LT1223 • TPC16 Power Supply Rejection vs Frequency 100 Output Impedance vs Frequency VS = ±15V 10 40 NEGATIVE 20 1 RF = RG = 3k RF = RG = 1k 0.1 0 10k 100k 1M FREQUENCY (Hz) 10M 100M 0.01 10k 100k 1M FREQUENCY (Hz) 10M 100M LT1223 • TPC18 LT1223 • TPC19 5 LT1223 TYPICAL PERFOR A CE CHARACTERISTICS Voltage Gain and Phase vs Frequency 20 15 10 GAIN RL = 100Ω PHASE RL = 100Ω RL ≥ 1k RL ≥ 1k VS = ±15V RF = RG = 1k 225 0.1 135 90 45 0 –45 –90 TOTAL HARMONIC DISTORTION (%) VOLTAGE GAIN (dB) 0 –5 –10 –15 –20 –25 –30 1M 10M 100M 1G LT1223 • TPC20 DISTORTION (dBc) 5 FREQUENCY (Hz) Noninverting Amplifier Settling Time to 10mV vs Output Step 10 8 6 OUTPUT STEP (V) 4 2 0 –2 –4 –6 –8 –10 0 20 40 60 80 100 SETTLING TIME (ns) LT1223 • TPC23 A V = +1 RF = 1k VS = ± 15V RL = 1k TO 10mV OUTPUT STEP (V) 2 0 –2 –4 –6 –8 –10 0 1 SETTLING TIME (µs) LT1223 • TPC24 OUTPUT STEP (V) TO 10mV APPLICATI S I FOR ATIO Current Feedback Basics The small-signal bandwidth of the LT1223, like all current feedback amplifiers, isn’t a straight inverse function of the closed-loop gain. This is because the feedback resistors determine the amount of current driving the amplifier’s internal compensation capacitor. In fact, the amplifier’s feedback resistor (RF) from output to inverting input works with internal junction capacitances of the LT1223 to set the closed-loop bandwidth. Even though the gain set resistor (RG) from inverting input to ground works with RF to set the voltage gain just like it does in a voltage feedback op amp, the closed-loop bandwidth does not change. This is because the equivalent gain bandwidth product of the current feedback amplifier is set by the Thevenin equivalent resistance at the inverting input and the internal compensation capacitor. By keeping RF constant and changing the gain with RG, the Thevenin resistance changes by the same amount as the change in gain. As a result, the net closed-loop bandwidth of the LT1223 remains the same for various closed-loop gains. 6 U W UW Total Harmonic Distortion vs Frequency 180 VS = ±15V VO = 7VRMS RL = 400 Ω RF = RG =1k –20 2nd and 3rd Harmonic Distortion vs Frequency VS VO RL RF AV = ± 15V = 2VP-P = 100 = 1k = 10dB 2ND 3RD –30 –135 –180 –225 PHASE SHIFT (DEGREES) –40 0.01 –50 THD –60 0.001 10 100 1k FREQUENCY (Hz) LT1223 • TPC21 10k 100k –70 1 10 FREQUENCY (MHz) LT1223 • TPC22 100 Noninverting Amplifier Settling Time to 1mV vs Output Step 10 8 6 4 A V = +1 R F = 1k VS = ± 15V RL = 1k Inverting Amplifier Settling Time vs Output Step 10 8 A V = –1 RF = 1k VS = ± 15V RL = 1k TO 10mV TO 1mV TO 1mV 6 4 2 0 –2 –4 –6 –8 –10 2 TO 1mV TO 10mV TO 1mV 0 20 40 60 80 100 SETTLING TIME (ns) LT1223 • TPC25 U UO LT1223 APPLICATI S I FOR ATIO The curve on the first page shows the LT1223 voltage gain versus frequency while driving 100Ω, for five gain settings from 1 to 100. The feedback resistor is a constant 1k and the gain resistor is varied from infinity to 10Ω. Shown for comparison is a plot of the fixed 100MHz gain bandwidth limitation that a voltage feedback amplifier would have. It is obvious that for gains greater than one, the LT1223 provides 3 to 20 times more bandwidth. It is also evident that second order effects reduce the bandwidth somewhat at the higher gain settings. Feedback Resistor Selection Because the feedback resistor determines the compensation of the LT1223, bandwidth and transient response can be optimized for almost every application. To increase the bandwidth when using higher gains, the feedback resistor (and gain resistor) can be reduced from the nominal 1k value. The Minimum Feedback Resistor versus Voltage Gain curve shows the values to use for ± 15V supplies. Larger feedback resistors can also be used to slow down the LT1223 as shown in the –3dB Bandwidth versus Feedback Resistor curve. Capacitive Loads The LT1223 can be isolated from capacitive loads with a small resistor (10Ω to 20Ω) or it can drive the capacitive load directly if the feedback resistor is increased. Both techniques lower the amplifier’s bandwidth about the same amount. The advantage of resistive isolation is that the bandwidth is only reduced when the capacitive load is present. The disadvantage of resistor isolation is that resistive loading causes gain errors. Because the DC accuracy is not degraded with resistive loading, the desired way of driving capacitive loads, such as flash converters, is to increase the feedback resistor. The Maximum Capacitive Load versus Feedback Resistor curve shows the value of feedback resistor and capacitive load that gives 5dB of peaking. For less peaking, use a larger feedback resistor. Power Supplies The LT1223 may be operated with single or split supplies as low as ± 4V (8V total) to as high as ±18V (36V total). It U is not necessary to use equal value split supplies, however, the offset voltage will degrade about 350µV per volt of mismatch. The internal compensation capacitor decreases with increasing supply voltage. The –3dB Bandwidth versus Supply Voltage curve shows how this affects the bandwidth for various feedback resistors. Generally, the bandwidth at ±5V supplies is about half the value it is at ±15V supplies for a given feedback resistor. The LT1223 is very stable even with minimal supply bypassing, however, the transient response will suffer if the supply rings. It is recommended for good slew rate and settling time that 4.7µF tantalum capacitors be placed within 0.5 inches of the supply pins. Input Range The noninverting input of the LT1223 looks like a 10M resistor in parallel with a 3pF capacitor until the common mode range is exceeded. The input impedance drops somewhat and the input current rises to about 10µA when the input comes too close to the supplies. Eventually, when the input exceeds the supply by one diode drop, the base collector junction of the input transistor forward biases and the input current rises dramatically. The input current should be limited to 10mA when exceeding the supplies. The amplifier will recover quickly when the input is returned to its normal common mode range unless the input was over 500mV beyond the supplies, then it will take an extra 100ns. Offset Adjust Output offset voltage is equal to the input offset voltage times the gain plus the inverting input bias current times the feedback resistor. For low gain applications (3 or less) a 10kΩ pot connected to pins 1 and 5 with wiper to V+ will trim the inverting input current (±10µA) to null the output; it does not change the offset voltage very much. If the LT1223 is used in a high gain application, where input offset voltage is the dominate error, it can be nulled by pulling approximately 100µA from pin 1 or 5. The easy way to do this is to use a 10kΩ pot between pin 1 and 5 with a 150k resistor from the wiper to ground for 15V supply applications. Use a 47k resistor when operating on a 5V supply. W U UO 7 LT1223 APPLICATI Shutdown Pin 8 activates a shutdown control function. Pulling more than 200µA from pin 8 drops the supply current to less than 3mA, and puts the output into a high impedance state. The easy way to force shutdown is to ground pin 8, using an open collector (drain) logic stage. An internal resistor limits current, allowing direct interfacing with no additional parts. When pin 8 is open, the LT1223 operates normally. Slew Rate The slew rate of a current feedback amplifier is not independent of the amplifier gain configuration the way it is in a traditional op amp. This is because the input stage and the output stage both have slew rate limitations. Inverting amplifiers do not slew the input and are therefore limited only by the output stage. High gain, noninverting amplifiers are similar. The input stage slew rate of the LT1223 is about 350V/µs before it becomes nonlinear and is enhanced by the normally reverse-biased emitters on the input transistors. The output slew rate depends on the size of the feedback resistors. The peak output slew rate is about 2000V/µs with a 1k feedback resistor and drops proportionally for larger values. At an output slew rate of 1000V/µs or more, the transistors in the “mirror circuits” will begin to saturate due to the large feedback currents. This causes the output to have slew induced overshoot and is somewhat unusual looking; it is in no way harmful or dangerous to the device. The photos show the LT1223 in a noninverting gain of three (RF = 1k, RG = 500Ω) with a 20V peak-to-peak output slewing at 500V/µs, 1000V/µs and 2000V/µs. Settling Time The Inverting Amplifier Settling Time versus Output Step curve shows that the LT1223 will settle to within 1mV of final value in less than 100ns for all output changes of 10V or less. When operated as an inverting amplifier there is less than 500µV of thermal settling in the amplifier. However, when operating the LT1223 as a noninverting amplifier, there is an additional thermal settling component that is about 200µV for every volt of input common mode change. So a noninverting gain of one amplifier will S I FOR ATIO 8 U Output Slew Rate of 500V/µs Output Slew Rate of 1000V/µs Output Slew Rate at 2000V/µs Shows Aberrations (See Text) W U UO LT1223 APPLICATI S I FOR ATIO have about 2.5mV thermal tail on a 10V step. Unfortunately, reducing the input signal and increasing the gain always results in a thermal tail of about the same amount for a given output step. For this reason we show separate graphs of 10mV and 1mV non-inverting amplifier settling times. Just as the bandwidth of the LT1223 is fairly constant for various closed-loop gains, the settling time remains constant as well. Adjustable Gain Amplifier To make a variable gain amplifier with the LT1223, vary the value of RG. The implementation of RG can be a pot, a light controlled resistor, a FET, or any other low capacitance variable resistor. The value of RF should not be varied to change the gain. If RF is changed, then the bandwidth will be reduced at maximum gain and the circuit will oscillate when RF is very small. V IN + LT1223 – RF VOUT RG LT1223 • TA03 Adjustable Bandwidth Amplifier Because the resistance at the inverting input determines the bandwidth of the LT1223, an adjustable bandwidth circuit can be made easily. The gain is set as before with RF and RG; the bandwidth is maximum when the variable resistor is at a minimum. V IN + LT1223 – 5k VOUT RG RF LT1223 • TA04 U Accurate Bandwidth Limiting The LT1223 It is very common to limit the bandwidth of an op amp by putting a small capacitor in parallel with RF. DO NOT PUT A SMALL CAPACITOR FROM THE INVERTING INPUT OF A CURRENT FEEDBACK AMPLIFIER TO ANYWHERE ELSE, ESPECIALLY NOT TO THE OUTPUT. The capacitor on the inverting input will cause peaking or oscillations. If you need to limit the bandwidth of a current feedback amplifier, use a resistor and capacitor at the noninverting input (R1 & C1). This technique will also cancel (to a degree) the peaking caused by stray capacitance at the inverting input. Unfortunately, this will not limit the output noise the way it does for the op amp. R1 + C1 – LT1223 VOUT V IN RF R1 = 300Ω C1 = 100pF BW = 5MHz RG LT1223 • TA05 W U UO Current Feedback Amplifier Integrator Since we remember that the inverting input wants to see a resistor, we can add one to the standard integrator circuit. This generates a new summing node where we can apply capacitive feedback. The LT1223 integrator has excellent large signal capability and accurate phase shift at high frequencies. + LT1223 VOUT =1 sCI RI VIN RI RF 1k – CI VOUT VIN LT1223 • TA06 9 LT1223 APPLICATI S I FOR ATIO Summing Amplifier (DC Accurate) The summing amplifier is easily made by adding additional inputs to the basic inverting amplifier configuration. The LT1223 has no IOS spec because there is no correlation between the two input bias currents. Therefore, we will not improve the DC accuracy of the inverting amplifier by putting in the extra resistor in the noninverting input. + R1 G V I1 R2 G V I2 • • • Rn G LT1223 – VOUT R F V (R I1 + VI2 + VIn R Gn G1 R G2 VOUT = –R F ) VIn LT1223 • TA07 Difference Amplifier The LT1223 difference amplifier delivers excellent performance if the source impedance is very low. This is because the common mode input resistance is only equal to RF + RG. RG V1 (RF –50) 100 + OPTIONAL TRIM FOR CMRR RG V2 VOUT = RF (V1 – V2 ) RG – LT1223 VOUT RF LT1223 • TA08 Video Instrumentation Amplifier This instrumentation amplifier uses two LT1223s to increase the input resistance to well over 1M. This makes an excellent “loop through” or cable sensing amplifier if the 10 U inverting input (A1) senses the shield and the non-inverting input (A2) senses the center conductor. Since this amplifier does not load the cable (take care to minimize stray capacitance) and it rejects common mode hum and noise, several amplifiers can sense the signal with only one termination at the end of the cable. The design equations are simple. Just select the gain you need (it should be two or more) and the value of the feedback resistor (typically 1k) and calculate RG1 and RG2. The gain can be tweaked with RG2 and the CMRR with RG1 if needed. The bandwidth of the noninverting input signal is not reduced by the presence of the other amplifier, however, the inverting input signal bandwidth is reduced since it passes two amplifiers. The CMRR is good at high frequencies because the bandwidth of the amplifiers are about the same even though they do not necessarily operate at the same gain. RG1 1k RF1 1k RG2 1k RF2 1k – A1 LT1223 + + – A2 LT1223 VOUT VIN – VOUT = G (VIN+ – VIN–) R RF1 = RF2; RG1 = (G – 1) RF2; RG2 = F2 G–1 TRIM GAIN (G) WITH RG2; TRIM CMRR WITH RG1 VIN + LT1223 • TA09 W U UO Cable Driver The cable driver circuit is shown on the front page. When driving a cable it is important to properly terminate both ends if even modest high frequency performance is required. The additional advantage of this is that it isolates the capacitive load of the cable from the amplifier so it can operate at maximum bandwidth. LT1223 TYPICAL APPLICATI V IN 2k 75Ω 75Ω R f = 2k TO STABILIZE CIRCUIT DIFFERENTIAL GAIN = 1% DIFFERENTIAL PHASE = 1° SI PLIFIED SCHE ATIC 7 15k 5 BIAS 1 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. UO 150mA Output Current Video Amp V+ V+ + LT1223 – V– 2k V– IN LT1010 75Ω 75Ω 20 Ω BIAS OUT 75Ω 75Ω 75Ω 75Ω 75Ω 75Ω LT1223 • TA10 W W 10k 8 3 2 6 BIAS 4 LT1223 • TA01 11 LT1223 PACKAGE DESCRIPTIO 0.290 – 0.320 (7.366 – 8.128) 0.008 – 0.018 (0.203 – 0.460) 0.385 ± 0.025 (9.779 ± 0.635) 0° – 15° 1 0.038 – 0.068 (0.965 – 1.727) 0.014 – 0.026 (0.360 – 0.660) 0.125 3.175 0.100 ± 0.010 MIN (2.540 ± 0.254) 0.055 (1.397) MAX 2 3 4 0.300 – 0.320 (7.620 – 8.128) 0.009 – 0.015 (0.229 – 0.381) 0.065 (1.651) TYP 0.125 (3.175) MIN 0.020 (0.508) MIN ( +0.025 0.325 –0.015 +0.635 8.255 –0.381 ) 0.045 ± 0.015 (1.143 ± 0.381) 0.100 ± 0.010 (2.540 ± 0.254) 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.016 – 0.050 0.406 – 1.270 0.053 – 0.069 (1.346 – 1.752) 0.004 – 0.010 (0.101 – 0.254) 0.228 – 0.244 (5.791 – 6.197) 0°– 8° TYP 12 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7487 (408) 432-1900 q FAX: (408) 434-0507 q TELEX: 499-3977 U Dimensions in inches (millimeters) unless otherwise noted. J8 Package 8-Lead Ceramic DIP 0.005 (0.127) MIN 0.405 (10.287) MAX 8 7 6 5 0.200 (5.080) MAX 0.015 – 0.060 (0.381 – 1.524) 0.025 (0.635) RAD TYP 0.220 – 0.310 (5.588 – 7.874) J8 0392 N8 Package 8-Lead Plastic DIP 0.400 (10.160) MAX 8 7 6 5 0.045 – 0.065 (1.143 – 1.651) 0.130 ± 0.005 (3.302 ± 0.127) 0.250 ± 0.010 (6.350 ± 0.254) 1 2 3 4 0.018 ± 0.003 (0.457 ± 0.076) N8 0392 S8 Package 8-Lead Plastic SOIC 0.189 – 0.197 (4.801 – 5.004) 8 7 6 5 0.014 – 0.019 (0.355 – 0.483) 0.050 (1.270) BSC 0.150 – 0.157 (3.810 – 3.988) 1 2 3 4 SO8 0392 LT/GP 1092 5K REV A © LINEAR TECHNOLOGY CORPORATION 1992