LT1228CJ8

LT1228 100MHz Current Feedback Amplifier with DC Gain Control FEATURES ■ DESCRIPTIO ■ ■ ■ ■ ■ ■ Very Fast Transconductance Amplifier Bandwidth: 75MHz gm = 10 × ISET Low THD: 0.2% at 30mVRMS Input Wide ISET Range: 1µA to 1mA Very Fast Current Feedback Amplifier Bandwidth: 100MHz Slew Rate: 1000V/µs Output Drive Current: 30mA Differential Gain: 0.04% Differential Phase: 0.1° High Input Impedance: 25MΩ, 6pF Wide Supply Range: ±2V to ±15V Inputs Common Mode to Within 1.5V of Supplies Outputs Swing Within 0.8V of Supplies Supply Current: 7mA Available in 8-Lead PDIP and SOIC Packages The LT®1228 makes it easy to electronically control the gain of signals from DC to video frequencies. The LT1228 implements gain control with a transconductance amplifier (voltage to current) whose gain is proportional to an externally controlled current. A resistor is typically used to convert the output current to a voltage, which is then amplified with a current feedback amplifier. The LT1228 combines both amplifiers into an 8-pin package, and operates on any supply voltage from 4V (± 2V) to 30V (±15V). A complete differential input, gain controlled amplifier can be implemented with the LT1228 and just a few resistors. The LT1228 transconductance amplifier has a high impedance differential input and a current source output with wide output voltage compliance. The transconductance, gm, is set by the current that flows into Pin 5, ISET. The small signal gm is equal to ten times the value of ISET and this relationship holds over several decades of set current. The voltage at Pin 5 is two diode drops above the negative supply, Pin 4. The LT1228 current feedback amplifier has very high input impedance and therefore it is an excellent buffer for the output of the transconductance amplifier. The current feedback amplifier maintains its wide bandwidth over a wide range of voltage gains making it easy to interface the transconductance amplifier output to other circuitry. The current feedback amplifier is designed to drive low impedance loads, such as cables, with excellent linearity at high frequencies. Frequency Response 6 3 0 –3 ISET = 1mA VS = ±15V RL = 100Ω APPLICATIO S ■ ■ ■ ■ ■ ■ Video DC Restore (Clamp) Circuits Video Differential Input Amplifiers Video Keyer/Fader Amplifiers AGC Amplifiers Tunable Filters Oscillators , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. TYPICAL APPLICATIO 15V R3A 10k R2A 10k Differential Input Variable Gain Amp + 3 4.7µF VIN gm 2 1 4 5 8 ISET R5 10k GAIN (dB) + – + – 7 –6 –9 –12 –15 ISET = 300µA + CFA 6 RF 470Ω HIGH INPUT RESISTANCE EVEN WHEN POWER IS OFF –18dB < GAIN < 2dB VIN ≤ 3VRMS LT1228 • TA01 VOUT –15V R3 100Ω R2 100Ω 4.7µF R4 1.24k R6 6.19k – –18 –21 –24 100k R1 270Ω RG 10Ω U ISET = 100µA 1M 10M 100M LT1228 • TA02 U U + FREQUENCY (Hz) 1228fc 1 LT1228 ABSOLUTE (Note 1) AXI U RATI GS PACKAGE/ORDER I FOR ATIO TOP VIEW IOUT –IN +IN V– 1 2 3 4 gm +– 8 7 6 5 GAIN V+ VOUT ISET Supply Voltage ...................................................... ± 18V Input Current, Pins 1, 2, 3, 5, 8 (Note 8) ............ ± 15mA Output Short Circuit Duration (Note 2) ......... Continuous Operating Temperature Range LT1228C ................................................ 0°C to 70°C LT1228I ............................................. –40°C to 85°C LT1228M (OBSOLETE) .............. –55°C to 125°C Storage Temperature Range ..................–65°C to 150°C Junction Temperature Plastic Package .............................................. 150°C Ceramic Package (OBSOLETE) ................ 175°C Lead Temperature (Soldering, 10 sec).................. 300°C N8 PACKAGE 8-LEAD PLASTIC DIP S8 PACKAGE 8-LEAD PLASTIC SOIC TJ MAX = 150°C, θJA = 100°C/W (N) TJ MAX = 150°C, θJA = 150°C/W (S) ORDER PART NUMBER LT1228CN8 LT1228CS8 LT1228IN8 LT1228IS8 S8 PART MARKING 1228 1228I ORDER PART NUMBER LT1228MJ8 LT1228CJ8 J8 PACKAGE 8-LEAD CERAMIC DIP TJ MAX = 175°C, θJA = 100°C/W (J) OBSOLETE PACKAGE Consider the N8 or S8 Packages for Alternate Source. Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: http://www.linear.com/leadfree/ ELECTRICAL CHARACTERISTICS SYMBOL VOS PARAMETER Input Offset Voltage Input Offset Voltage Drift IIN + The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Current Feedback Amplifier, Pins 1, 6, 8. ± 5V ≤ VS ≤ ± 15V, ISET = 0µA, VCM = 0V unless otherwise noted. CONDITIONS TA = 25°C ● ● MIN TYP ±3 10 ±0.3 MAX ±10 ±15 ±3 ±10 ±65 ±100 UNITS mV mV µV/°C µA µA µA µA nV/√Hz pV/√Hz MΩ MΩ pF V V V V dB dB dB dB 1228fc Noninverting Input Current Inverting Input Current Input Noise Voltage Density Input Noise Current Density Input Resistance Input Capacitance (Note 3) Input Voltage Range TA = 25°C ● IIN– en in RIN CIN TA = 25°C ● ±10 6 1.4 ● ● f = 1kHz, RF = 1k, RG = 10Ω, RS = 0Ω f = 1kHz, RF = 1k, RG = 10Ω, RS = 10k VIN = ±13V, VS = ±15V VIN = ± 3V, VS = ± 5V VS = ±5V VS = ±15V, TA = 25°C VS = ± 5V, TA = 25°C ● ● 2 2 ±13 ±12 ±3 ±2 55 55 55 55 25 25 6 ±13.5 ±3.5 69 69 CMRR Common Mode Rejection Ratio VS = ±15V, VCM = ±13V, TA = 25°C VS = ± 15V, VCM = ± 12V VS = ±5V, VCM = ±3V, TA = 25°C VS = ± 5V, VCM = ± 2V ● ● 2 U W U U WW W LT1228 The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Current Feedback Amplifier, Pins 1, 6, 8. ± 5V ≤ VS ≤ ± 15V, ISET = 0µA, VCM = 0V unless otherwise noted. SYMBOL PARAMETER Inverting Input Current Common Mode Rejection CONDITIONS VS = ±15V, VCM = ±13V, TA = 25°C VS = ± 15V, VCM = ± 12V VS = ± 5V, VCM = ± 3V, TA = 25°C VS = ± 5V, VCM = ± 2V VS = ± 2V to ±15V, TA = 25°C VS = ± 3V to ±15V VS = ± 2V to ±15V, TA = 25°C VS = ± 3V to ±15V VS = ± 2V to ±15V, TA = 25°C VS = ± 3V to ±15V VS = ±15V, VOUT = ±10V, RLOAD = 1k VS = ± 5V, VOUT = ± 2V, RLOAD = 150Ω VS = ±15V, VOUT = ±10V, RLOAD = 1k VS = ± 5V, VOUT = ± 2V, RLOAD = 150Ω VS = ±15V, RLOAD = 400Ω, TA = 25°C VS = ± 5V, RLOAD = 150Ω, TA = 25°C IOUT Is SR SR tr BW tr Maximum Output Current Supply Current Slew Rate (Notes 4 and 6) Slew Rate Rise Time (Notes 5 and 6) Small-Signal Bandwidth Small-Signal Rise Time Propagation Delay Small-Signal Overshoot ts Settling Time Differential Gain (Note 7) Differential Phase (Note 7) Differential Gain (Note 7) Differential Phase (Note 7) RLOAD = 0Ω, TA = 25°C ● ● ● ● ELECTRICAL CHARACTERISTICS MIN TYP 2.5 2.5 MAX 10 10 10 10 UNITS µA/V µA/V µA/V µA/V dB dB ● ● ● PSRR Power Supply Rejection Ratio Noninverting Input Current Power Supply Rejection Inverting Input Current Power Supply Rejection 60 60 80 10 0.1 50 50 5 5 nA/V nA/V µA/V µA/V dB dB kΩ kΩ V V V V ● ● ● ● ● AV ROL VOUT Large-Signal Voltage Gain Transresistance, ∆VOUT/∆IIN– Maximum Output Voltage Swing 55 55 100 100 ±12 ±10 ±3 ±2.5 30 25 300 65 65 200 200 ±13.5 ±3.7 65 6 500 3500 10 100 3.5 3.5 15 45 0.01 0.01 0.04 0.1 20 125 125 11 mA mA mA V/µs V/µs ns MHz ns ns % ns % DEG % DEG VOUT = 0V, ISET = 0V TA = 25°C VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 400Ω TA = 25°C VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 100Ω 0.1%, VOUT = 10V, RF =1k, RG= 1k, RL =1k VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 1k VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 1k VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 150Ω VS = ±15V, RF = 750Ω, RG= 750Ω, RL = 150Ω ● ELECTRICAL CHARACTERISTICS SYMBOL VOS PARAMETER Input Offset Voltage Input Offset Voltage Drift IOS Input Offset Current The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Transconductance Amplifier, Pins 1, 2, 3, 5. ± 5V ≤ VS ≤ ± 15V, ISET = 100µA, VCM = 0V unless otherwise noted. CONDITIONS ISET = 1mA, TA = 25°C ● ● MIN TYP ±0.5 10 40 MAX ±5 ±10 200 500 UNITS mV mV µV/°C nA nA TA = 25°C ● 1228fc 3 LT1228 The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. Transconductance Amplifier, Pins 1, 2, 3, 5. ± 5V ≤ VS ≤ ± 15V, ISET = 100µA, VCM = 0V unless otherwise noted. SYMBOL IB en RIN PARAMETER Input Bias Current Input Noise Voltage Density Input Resistance-Differential Mode Input Resistance-Common Mode CIN Input Capacitance Input Voltage Range VS = ±15V, TA = 25°C VS = ±15V VS = ±5V, TA = 25°C VS = ±5V VS = ±15V, VCM = ±13V, TA = 25°C VS = ± 15V, VCM = ± 12V VS = ±5V, VCM = ±3V, TA = 25°C VS = ± 5V, VCM = ± 2V VS = ± 2V to ±15V, TA = 25°C VS = ± 3V to ±15V ISET = 100µA, IOUT = ±30µA, TA = 25°C ● ● ● ● ● ● ELECTRICAL CHARACTERISTICS CONDITIONS TA = 25°C ● MIN TYP 0.4 20 MAX 1 5 UNITS µA µA nV/√Hz kΩ MΩ MΩ pF V V V V dB dB dB dB dB dB f = 1kHz VIN ≈ ±30mV VS = ±15V, VCM = ±12V VS = ±5V, VCM = ± 2V ● ● ● 30 50 50 ±13 ±12 ±3 ±2 60 60 60 60 60 60 0.75 200 1000 1000 3 ±14 ±4 100 100 100 1.00 – 0.33 100 0.3 1.25 130 3 10 CMRR Common Mode Rejection Ratio PSRR gm IOUT IOL VOUT RO Power Supply Rejection Ratio Transconductance Transconductance Drift Maximum Output Current Output Leakage Current Maximum Output Voltage Swing Output Resistance Output Capacitance (Note 3) µA/mV %/°C µA µA µA V V MΩ MΩ pF ISET = 100µA ISET = 0µA (+IIN of CFA), TA = 25°C ● ● 70 VS = ±15V , R1 = ∞ VS = ±5V , R1 = ∞ VS = ±15V, VOUT = ±13V VS = ±5V, VOUT = ±3V VS = ±5V ISET = 1mA VIN = 30mVRMS at 1kHz, R1 = 100k R1 = 50Ω, ISET = 500µA R1 = 50Ω, ISET = 500µA, 10% to 90% R1 = 50Ω, ISET = 500µA, 50% to 50% ● ● ● ● ● ± 13 ±3 2 2 ± 14 ±4 8 8 6 9 0.2 80 5 5 15 IS THD BW tr Supply Current, Both Amps Total Harmonic Distortion Small-Signal Bandwidth Small-Signal Rise Time Propagation Delay mA % MHz ns ns Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: A heat sink may be required depending on the power supply voltage. Note 3: This is the total capacitance at Pin 1. It includes the input capacitance of the current feedback amplifier and the output capacitance of the transconductance amplifier. Note 4: Slew rate is measured at ± 5V on a ±10V output signal while operating on ±15V supplies with RF = 1k, RG = 110Ω and RL = 400Ω. The slew rate is much higher when the input is overdriven, see the applications section. Note 5: Rise time is measured from 10% to 90% on a ± 500mV output signal while operating on ±15V supplies with RF = 1k, RG = 110Ω and RL = 100Ω. This condition is not the fastest possible, however, it does guarantee the internal capacitances are correct and it makes automatic testing practical. Note 6: AC parameters are 100% tested on the ceramic and plastic DIP packaged parts (J and N suffix) and are sample tested on every lot of the SO packaged parts (S suffix). Note 7: NTSC composite video with an output level of 2V. Note 8: Back to back 6V Zener diodes are connected between Pins 2 and 3 for ESD protection. 1228fc 4 LT1228 TYPICAL PERFOR A CE CHARACTERISTICS Transconductance Amplifier, Pins 1, 2, 3 & 5 Small-Signal Bandwidth vs Set Current 100 VS = ±15V R1 = 100Ω TRANSCONDUCTANCE (µA/mV) –3dB BANDWIDTH (MHz) 10 1000 SET CURRENT (µA) TRANSCONDUCTANCE (µA/mV) R1 = 1k 10 R1 = 10k 1 R1 = 100k 0.1 10 100 SET CURRENT (µA) LT1228 • TPC01 Total Harmonic Distortion vs Input Voltage 10 VS = ±15V 1000 COMMON MODE RANGE (V) OUTPUT DISTORTION (%) SPOT NOISE (pA/√Hz) 1 ISET = 100µA 0.1 ISET = 1mA 0.01 1 10 100 1000 LT1228 • TPC04 INPUT VOLTAGE (mVP–P) Small-Signal Control Path Bandwidth vs Set Current 100 VS = ±2V TO ±15V VIN = 200mV (PIN 2 TO 3) 1.0 0.9 CONTROL PATH GAIN (µA/µA) OUTPUT SATURATION VOLTAGE (V) –3dB BANDWIDTH (MHz) 10 ∆IOUT ∆ISET 1 10 100 SET CURRENT (µA) LT1228 • TPC07 UW Small-Signal Transconductance and Set Current vs Bias Voltage 100 VS = ±2V TO ±15V TA = 25°C 10000 2.0 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 1000 Small-Signal Transconductance vs DC Input Voltage V S = ±2V TO ±15V ISET = 100µA –55°C 25°C 125°C 1 100 0.1 10 0.01 1.0 0.001 0.9 1.0 1.1 1.2 1.3 1.4 0.1 1.5 0 –200 –150 –100 –50 0 50 100 150 200 LT1228 • TPC03 BIAS VOLTAGE, PIN 5 TO 4, (V) LT1228 • TPC02 INPUT VOLTAGE (mVDC) Spot Output Noise Current vs Frequency VS = ±2V TO ±15V TA = 25°C Input Common Mode Limit vs Temperature V+ –0.5 –1.0 –1.5 –2.0 V + = 2V TO 15V ISET = 1mA 100 2.0 1.5 1.0 0.5 V – = –2V TO –15V ISET = 100µA 10 10 100 1k FREQUENCY (Hz) LT1228 • TPC05 10k 100k V– –50 –25 0 25 50 75 100 125 TEMPERATURE (°C) LT1228 • TPC06 Small-Signal Control Path Gain vs Input Voltage V+ –0.5 –1.0 Output Saturation Voltage vs Temperature 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 ∆IOUT ∆ISET ± 2V ≤ VS ≤ ±15V R1 = ∞ +1.0 +0.5 V– –50 1000 0 40 80 120 160 200 –25 0 25 50 75 100 125 INPUT VOLTAGE, PIN 2 TO 3, (mVDC) LT1228 • TPC08 TEMPERATURE (°C) LT1228 • TPC09 1228fc 5 LT1228 TYPICAL PERFOR A CE CHARACTERISTICSCurrent Feedback Amplifier, Pins 1, 6, 8 Voltage Gain and Phase vs Frequency, Gain = 6dB 8 7 6 GAIN PHASE 0 45 90 180 160 PEAKING ≤ 0.5dB PEAKING ≤ 5dB RF = 500Ω RF = 750Ω RF = 1k –3dB BANDWIDTH (MHz) –3dB BANDWIDTH (MHz) VOLTAGE GAIN (dB) 5 4 3 2 1 0 –1 –2 0.1 1 10 100 LT1228 • TPC10 VS = ±15V RL = 100Ω RF = 750Ω FREQUENCY (MHz) Voltage Gain and Phase vs Frequency, Gain = 20dB 22 21 20 GAIN PHASE 0 45 90 135 180 225 180 160 19 18 17 16 15 14 13 12 0.1 1 10 100 LT1228 • TPC13 –3dB BANDWIDTH (MHz) –3dB BANDWIDTH (MHz) VOLTAGE GAIN (dB) VS = ±15V RL = 100Ω RF = 750Ω FREQUENCY (MHz) Voltage Gain and Phase vs Frequency, Gain = 40dB 42 41 40 PHASE 0 45 18 16 –3dB BANDWIDTH (MHz) VOLTAGE GAIN (dB) 39 38 37 36 35 34 33 32 0.1 1 10 100 LT1228 • TPC16 135 180 225 12 10 8 6 4 2 0 0 2 4 6 8 10 –3dB BANDWIDTH (MHz) GAIN VS = ±15V RL = 100Ω RF = 750Ω FREQUENCY (MHz) 6 UW –3dB Bandwidth vs Supply Voltage, Gain = 2, RL = 100Ω 180 160 140 120 100 80 60 40 20 0 0 2 4 6 8 10 12 14 16 18 0 –3dB Bandwidth vs Supply Voltage, Gain = 2, RL = 1k 140 120 100 80 60 40 20 0 SUPPLY VOLTAGE (±V) LT1228 • TPC11 RF = 500Ω RF = 750Ω 135 180 225 PHASE SHIFT (DEG) PHASE SHIFT (DEG) PHASE SHIFT (DEG) PEAKING ≤ 0.5dB PEAKING ≤ 5dB RF = 1k RF = 2k RF = 2k 2 4 6 8 10 12 14 16 18 SUPPLY VOLTAGE (±V) LT1228 • TPC12 –3dB Bandwidth vs Supply Voltage, Gain = 10, RL = 100Ω 180 PEAKING ≤ 0.5dB PEAKING ≤ 5dB 160 140 120 100 80 60 40 20 0 0 2 4 6 8 10 12 14 16 18 0 –3dB Bandwidth vs Supply Voltage, Gain = 10, RL = 1kΩ PEAKING ≤ 0.5dB PEAKING ≤ 5dB 140 120 100 80 60 40 20 0 SUPPLY VOLTAGE (±V) LT1228 • TPC14 RF = 250Ω RF = 250Ω RF = 500Ω RF = 750Ω RF = 1k RF = 2k RF = 500Ω RF = 750Ω RF = 1k RF = 2k 2 4 6 8 10 12 14 16 18 SUPPLY VOLTAGE (±V) LT1228 • TPC15 –3dB Bandwidth vs Supply Voltage, Gain = 100, RL = 100Ω 18 16 14 12 10 8 6 4 2 0 12 14 16 18 0 –3dB Bandwidth vs Supply Voltage, Gain = 100, RL = 1kΩ 90 14 RF = 500Ω RF = 1k RF = 2k RF = 500Ω RF = 1k RF = 2k 2 4 6 8 10 12 14 16 18 SUPPLY VOLTAGE (±V) LT1228 • TPC17 SUPPLY VOLTAGE (±V) LT1228 • TPC18 1228fc LT1228 TYPICAL PERFOR A CE CHARACTERISTICS Current Feedback Amplifier, Pins 1, 6, 8 Maximum Capacitive Load vs Feedback Resistor 10k VS = ± 5V TOTAL HARMONIC DISTORTION (%) 0.10 VS = ±15V RL = 400Ω RF = RG = 750Ω CAPACITIVE LOAD (pF) 1k DISTORTION (dBc) 100 VS = ±15V 10 RL = 1k PEAKING ≤ 5dB GAIN = 2 1 0 1 2 3 LT1228 • TPC19 FEEDBACK RESISTOR (kΩ) Input Common Mode Limit vs Temperature V+ V+ –0.5 –1.0 COMMON MODE RANGE (V) –1.0 –1.5 –2.0 V + = 2V TO 15V OUTPUT SATURATION VOLTAGE (V) –0.5 OUTPUT SHORT-CIRCUIT CURRENT (mA) 2.0 1.5 1.0 0.5 V– –50 –25 0 25 50 75 100 125 V – = –2V TO –15V TEMPERATURE (°C) LT1228 • TPC22 Spot Noise Voltage and Current vs Frequency 100 80 POWER SUPPLY REJECTION (dB) SPOT NOISE (nV/√Hz OR pA/√Hz) 60 POSITIVE 40 OUTPUT IMPEDANCE (Ω) –in 10 en +in 1 10 100 1k FREQUENCY (Hz) LT1228 • TPC25 10k UW Total Harmonic Distortion vs Frequency –20 2nd and 3rd Harmonic Distortion vs Frequency VS = ±15V VO = 2VP–P RL = 100Ω RF = 750Ω AV = 10dB –30 2nd –40 3rd –50 0.01 VO = 7VRMS VO = 1VRMS –60 0.001 10 100 1k FREQUENCY (Hz) LT1228 • TPC20 10k 100k –70 1 10 FREQUENCY (MHz) LT1228 • TPC21 100 Output Saturation Voltage vs Temperature 70 Output Short-Circuit Current vs Temperature 60 RL = ∞ ± 2V ≤ VS ≤ ±15V 50 1.0 0.5 V– –50 –25 40 0 25 50 75 100 125 30 –50 –25 0 25 50 75 100 125 150 175 LT1228 • TPC24 TEMPERATURE (°C) LT1228 • TPC23 TEMPERATURE (°C) Power Supply Rejection vs Frequency 100 VS = ±15V RL = 100Ω RF = RG = 750Ω Output Impedance vs Frequency VS = ±15V 10 1.0 RF = RG = 2k 0.1 RF = RG = 750Ω NEGATIVE 20 0.01 100k 0 10k 100k 1M FREQUENCY (Hz) 10M 100M 0.001 10k 100k 1M FREQUENCY (Hz) 10M 100M LT1228 • TPC26 LT1228 • TPC27 1228fc 7 LT1228 TYPICAL PERFOR A CE CHARACTERISTICS Current Feedback Amplifier, Pins 1, 6 & 8 Settling Time to 10mV vs Output Step 10 8 6 OUTPUT STEP (V) NONINVERTING INVERTING OUTPUT STEP (V) 4 2 0 –2 –4 –6 –8 –10 0 20 40 60 80 100 SETTLING TIME (ns) LT1228 • TPC28 4 2 0 –2 –4 –6 INVERTING NONINVERTING 0 4 8 12 16 20 VS = ±15V RF = RG = 1k SUPPLY CURRENT (mA) VS = ±15V RF = RG = 1k NONINVERTING INVERTING SI PLIFIED SCHE ATIC 7 V+ +IN 3 ISET 5 8 UW W –IN 2 Settling Time to 1mV vs Output Step 10 8 6 NONINVERTING INVERTING 10 9 8 7 6 5 4 3 2 1 0 Supply Current vs Supply Voltage –55°C 25°C 125°C 175°C –8 –10 0 2 4 6 8 10 12 14 16 18 SETTLING TIME (µs) LT1228 • TPC29 SUPPLY VOLTAGE (±V) LT1228 • TPC30 W BIAS IOUT 1 8 GAIN 6 VOUT 4 V– LT1228 • TA03 1228fc LT1228 APPLICATI S I FOR ATIO The LT1228 contains two amplifiers, a transconductance amplifier (voltage-to-current) and a current feedback amplifier (voltage-to-voltage). The gain of the transconductance amplifier is proportional to the current that is externally programmed into Pin 5. Both amplifiers are designed to operate on almost any available supply voltage from 4V (± 2V) to 30V (±15V). The output of the transconductance amplifier is connected to the noninverting input of the current feedback amplifier so that both fit into an eight pin package. TRANSCONDUCTANCE AMPLIFIER The LT1228 transconductance amplifier has a high impedance differential input (Pins 2 and 3) and a current source output (Pin 1) with wide output voltage compliance. The voltage to current gain or transconductance (gm) is set by the current that flows into Pin 5, ISET. The voltage at Pin 5 is two forward biased diode drops above the negative supply, Pin 4. Therefore the voltage at Pin 5 (with respect to V –) is about 1.2V and changes with the log of the set current (120mV/decade), see the characteristic curves. The temperature coefficient of this voltage is about –4mV/°C (–3300ppm/°C) and the temperature coefficient of the logging characteristic is 3300ppm/°C. It is important that the current into Pin 5 be limited to less than 15mA. THE LT1228 WILL BE DESTROYED IF PIN 5 IS SHORTED TO GROUND OR TO THE POSITIVE SUPPLY. A limiting resistor (2k or so) should be used to prevent more than 15mA from flowing into Pin 5. The small-signal transconductance (gm) is equal to ten times the value of ISET (in mA/mV) and this relationship holds over many decades of set current (see the characteristic curves). The transconductance is inversely proportional to absolute temperature (–3300ppm/°C). The input stage of the transconductance amplifier has been designed to operate with much larger signals than is possible with an ordinary diff-amp. The transconductance of the input stage varies much less than 1% for differential input signals over a ±30 mV range (see the characteristic curve Small-Signal Transconductance vs DC Input Voltage). gm 4 R 5 ISET 2.5V 2Eg Vbe U Resistance Controlled Gain If the set current is to be set or varied with a resistor or potentiometer it is possible to use the negative temperature coefficient at Pin 5 (with respect to Pin 4) to compensate for the negative temperature coefficient of the transconductance. The easiest way is to use an LT1004-2.5, a 2.5V reference diode, as shown below: Temperature Compensation of gm with a 2.5V Reference R ISET Vbe LT1004-2.5 V– LT1228 ¥ TA04 W U UO The current flowing into Pin 5 has a positive temperature coefficient that cancels the negative coefficient of the transconductance. The following derivation shows why a 2.5V reference results in zero gain change with temperature: Since g m = and V be q ISET × = 10 • ISET kT 3.87 ⎛ cT n ⎞ akT = Eg – where a = In ⎜ ⎟ q ⎝ Ic ⎠ , ≈ 19.4 at 27°C c = 0.001 n = 3, Ic = 100µA ( ) Eg is about 1.25V so the 2.5V reference is 2Eg. Solving the loop for the set current gives: ISET = ⎛ akT ⎞ 2E g – 2 ⎜ E g – ⎟ q⎠ ⎝ R or ISET = 2akT Rq 1228fc 9 LT1228 APPLICATI S I FOR ATIO Substituting into the equation for transconductance gives: gm = a 10 = 1.94R R The temperature variation in the term “a” can be ignored since it is much less than that of the term “T” in the equation for Vbe. Using a 2.5V source this way will maintain the gain constant within 1% over the full temperature range of –55°C to 125°C. If the 2.5V source is off by 10%, the gain will vary only about ±6% over the same temperature range. We can also temperature compensate the transconductance without using a 2.5V reference if the negative power supply is regulated. A Thevenin equivalent of 2.5V is generated from two resistors to replace the reference. The two resistors also determine the maximum set current, approximately 1.1V/RTH. By rearranging the Thevenin equations to solve for R4 and R6 we get the following equations in terms of RTH and the negative supply, VEE. R4 = R TH ⎛ 2.5V ⎞ ⎜1 – V ⎟ ⎝ EE ⎠ and R6 = R THV EE 2.5V Temperature Compensation of gm with a Thevenin Voltage 1.03k ISET gm R6 6.19kΩ R' R4 1.24kΩ –15V LT1228 ¥ TA05 R' V1 4 5 ISET VTH = 2.5V Vbe R1 = R2 R3 = R4 IOUT = (V1 – V2) R3 • = 1mA/V R5 R1 50pF Voltage Controlled Gain To use a voltage to control the gain of the transconduc- 10 U tance amplifier requires converting the voltage into a current that flows into Pin 5. Because the voltage at Pin 5 is two diode drops above the negative supply, a single resistor from the control voltage source to Pin 5 will suffice in many applications. The control voltage is referenced to the negative supply and has an offset of about 900mV. The conversion will be monotonic, but the linearity is determined by the change in the voltage at Pin 5 (120mV per decade of current). The characteristic is very repeatable since the voltage at Pin 5 will vary less than ± 5% from part to part. The voltage at Pin 5 also has a negative temperature coefficient as described in the previous section. When the gain of several LT1228s are to be varied together, the current can be split equally by using equal value resistors to each Pin 5. For more accurate (and linear) control, a voltage-tocurrent converter circuit using one op amp can be used. The following circuit has several advantages. The input no longer has to be referenced to the negative supply and the input can be either polarity (or differential). This circuit works on both single and split supplies since the input voltage and the Pin 5 voltage are independent of each other. The temperature coefficient of the output current is set by R5. R3 1M R1 1M R2 1M V2 W U UO + LT1006 R5 1k Vbe – R4 1M IOUT TO PIN 5 OF LT1228 LT1228 ¥ TA19 Digital control of the transconductance amplifier gain is done by converting the output of a DAC to a current flowing into Pin 5. Unfortunately most current output DACs sink rather than source current and do not have output 1228fc LT1228 APPLICATI S I FOR ATIO compliance compatible with Pin 5 of the LT1228. Therefore, the easiest way to digitally control the set current is to use a voltage output DAC and a voltage-to-current circuit. The previous voltage-to-current converter will take the output of any voltage output DAC and drive Pin 5 with a proportional current. The R, 2R CMOS multiplying DACs operating in the voltage switching mode work well on both single and split supplies with the above circuit. Logarithmic control is often easier to use than linear control. A simple circuit that doubles the set current for each additional volt of input is shown in the voltage controlled state variable filter application near the end of this data sheet. Transconductance Amplifier Frequency Response The bandwidth of the transconductance amplifier is a function of the set current as shown in the characteristic curves. At set currents below 100µA, the bandwidth is approximately: –3dB bandwidth = 3 • 1011 ISET The peak bandwidth is about 80MHz at 500 µA. When a resistor is used to convert the output current to a voltage, the capacitance at the output forms a pole with the resistor. The best case output capacitance is about 5pF with ±15V supplies and 6pF with ±5V supplies. You must add any PC board or socket capacitance to these values to get the total output capacitance. When using a 1k resistor at the output of the transconductance amp, the output capacitance limits the bandwidth to about 25MHz. The output slew rate of the transconductance amplifier is the set current divided by the output capacitance, which is 6pF plus board and socket capacitance. For example with the set current at 1mA, the slew rate would be over 100V/µs. CURRENT FEEDBACK AMPLIFIER The LT1228 current feedback amplifier has very high noninverting input impedance and is therefore an excellent buffer for the output of the transconductance amplifier. The noninverting input is at Pin 1, the inverting input at Pin 8 and the output at Pin 6. The current feedback amplifier maintains its wide bandwidth for almost all voltage gains making it easy to interface the output levels of the transconductance amplifier to other circuitry. The current feedback amplifier is designed to drive low impedance loads such as cables with excellent linearity at high frequencies. Feedback Resistor Selection The small-signal bandwidth of the LT1228 current feedback amplifier is set by the external feedback resistors and the internal junction capacitors. As a result, the bandwidth is a function of the supply voltage, the value of the feedback resistor, the closed-loop gain and load resistor. The characteristic curves of bandwidth versus supply voltage are done with a heavy load (100Ω) and a light load (1k) to show the effect of loading. These graphs also show U Transconductance Amp Small-Signal Response ISET = 500µA, R1 = 50Ω 1228fc W U UO 11 LT1228 APPLICATIO S I FOR ATIO the family of curves that result from various values of the feedback resistor. These curves use a solid line when the response has less than 0.5dB of peaking and a dashed line for the response with 0.5dB to 5dB of peaking. The curves stop where the response has more than 5dB of peaking. Current Feedback Amp Small-Signal Response VS = ±15V, RF = RG = 750Ω, RL = 100Ω At a gain of two, on ±15V supplies with a 750Ω feedback resistor, the bandwidth into a light load is over 160MHz without peaking, but into a heavy load the bandwidth reduces to 100MHz. The loading has so much effect because there is a mild resonance in the output stage that enhances the bandwidth at light loads but has its Q reduced by the heavy load. This enhancement is only useful at low gain settings, at a gain of ten it does not boost the bandwidth. At unity gain, the enhancement is so effective the value of the feedback resistor has very little effect on the bandwidth. At very high closed-loop gains, the bandwidth is limited by the gain-bandwidth product of about 1GHz. The curves show that the bandwidth at a closed-loop gain of 100 is 10MHz, only one tenth what it is at a gain of two. GAIN (dB) 12 U Capacitance on the Inverting Input Current feedback amplifiers want resistive feedback from the output to the inverting input for stable operation. Take care to minimize the stray capacitance between the output and the inverting input. Capacitance on the inverting input to ground will cause peaking in the frequency response (and overshoot in the transient response), but it does not degrade the stability of the amplifier. The amount of capacitance that is necessary to cause peaking is a function of the closed-loop gain taken. The higher the gain, the more capacitance is required to cause peaking. For example, in a gain of 100 application, the bandwidth can be increased from 10MHz to 17MHz by adding a 2200pF capacitor, as shown below. CG must have very low series resistance, such as silver mica. VIN 1 W U U + CFA 6 VOUT 8 – RF 510Ω CG RG 5.1Ω LT1228 • TA08 Boosting Bandwidth of High Gain Amplifier with Capacitance On Inverting Input 49 46 43 40 37 34 31 28 25 22 19 1 10 FREQUENCY (MHz) LT1228 • TA09 CG = 4700pF CG = 2200pF CG = 0 100 1228fc LT1228 APPLICATI S I FOR ATIO Capacitive Loads The LT1228 current feedback amplifier can drive capacitive loads directly when the proper value of feedback resistor is used. The graph of Maximum Capacitive Load vs Feedback Resistor should be used to select the appropriate value. The value shown is for 5dB peaking when driving a 1k load, at a gain of 2. This is a worst case condition, the amplifier is more stable at higher gains, and driving heavier loads. Alternatively, a small resistor (10Ω to 20Ω) can be put in series with the output to isolate the capacitive load from the amplifier output. This has the advantage that the amplifier bandwidth is only reduced when the capacitive load is present and the disadvantage that the gain is a function of the load resistance. Slew Rate The slew rate of the current feedback amplifier is not independent of the amplifier gain configuration the way it is in a traditional op amp. This is because the input stage and the output stage both have slew rate limitations. The input stage of the LT1228 current feedback amplifier slews at about 100V/µs before it becomes nonlinear. Faster input signals will turn on the normally reverse biased emitters on the input transistors and enhance the slew rate significantly. This enhanced slew rate can be as much as 3500V/µs! Current Feedback Amp Large-Signal Response VS = ±15V, RF = RG = 750Ω Slew Rate Enhanced U The output slew rate is set by the value of the feedback resistors and the internal capacitance. At a gain of ten with a 1k feedback resistor and ±15V supplies, the output slew rate is typically 500V/µs and –850V/µs. There is no input stage enhancement because of the high gain. Larger feedback resistors will reduce the slew rate as will lower supply voltages, similar to the way the bandwidth is reduced. Current Feedback Amp Large-Signal Response VS = ±15V, RF = 1k, RG = 110Ω, RL = 400Ω W U UO Settling Time The characteristic curves show that the LT1228 current feedback amplifier settles to within 10mV of final value in 40ns to 55ns for any output step less than 10V. The curve of settling to 1mV of final value shows that there is a slower thermal contribution up to 20µs. The thermal settling component comes from the output and the input stage. The output contributes just under 1mV/V of output change and the input contributes 300µV/V of input change. Fortunately the input thermal tends to cancel the output thermal. For this reason the noninverting gain of two configuration settles faster than the inverting gain of one. 1228fc 13 LT1228 APPLICATIO S I FOR ATIO Power Supplies The LT1228 amplifiers will operate from single or split supplies from ± 2V (4V total) to ±18V (36V total). It is not necessary to use equal value split supplies, however the offset voltage and inverting input bias current of the current feedback amplifier will degrade. The offset voltage changes about 350µV/V of supply mismatch, the inverting bias current changes about 2.5µA/V of supply mismatch. Power Dissipation The worst case amplifier power dissipation is the total of the quiescent current times the total power supply voltage plus the power in the IC due to the load. The quiescent supply current of the LT1228 transconductance amplifier is equal to 3.5 times the set current at all temperatures. The quiescent supply current of the LT1228 current feedback amplifier has a strong negative temperature coefficient and at 150°C is less than 7mA, typically only 4.5mA. The power in the IC due to the load is a function of the output voltage, the supply voltage and load resistance. The worst case occurs when the output voltage is at half supply, if it can go that far, or its maximum value if it cannot reach half supply. TYPICAL APPLICATIO S Basic Gain Control The basic gain controlled amplifier is shown on the front page of the data sheet. The gain is directly proportional to the set current. The signal passes through three stages from the input to the output. First the input signal is attenuated to match the dynamic range of the transconductance amplifier. The attenuator should reduce the signal down to less than 100mV peak. The characteristic curves can be used to estimate how much distortion there will be at maximum input signal. For single ended inputs eliminate R2A or R3A. The signal is then amplified by the transconductance amplifier (gm) and referred to ground. The voltage gain of the transconductance amplifier is: g m • R1 = 10 • ISET • R1 Lastly the signal is buffered and amplified by the current feedback amplifier (CFA). The voltage gain of the current feedback amplifier is: 1+ RF RG 14 U For example, let’s calculate the worst case power dissipation in a variable gain video cable driver operating on ±12V supplies that delivers a maximum of 2V into 150Ω. The maximum set current is 1mA. W U U U P D = 2VS ISMAX + 3.5ISET + VS – V OMAX ( )( P D = 2 • 12V • 7mA + 3.5 • 1mA + 12V – 2V = 0.252 + 0.133 = 0.385W [ ( )] ( ) VOMAX RL 2 ) 150VΩ The total power dissipation times the thermal resistance of the package gives the temperature rise of the die above ambient. The above example in SO-8 surface mount package (thermal resistance is 150°C/W) gives: Temperature Rise = PDθJA = 0.385W • 150°C/W = 57.75°C Therefore the maximum junction temperature is 70°C +57.75°C or 127.75°C, well under the absolute maximum junction temperature for plastic packages of 150°C. The overall gain of the gain controlled amplifier is the product of all three stages: ⎛ R⎞ ⎛ R3 ⎞ • 10 • ISET • R1 • ⎜ 1 + F ⎟ AV = ⎜ ⎟ ⎝ RG ⎠ ⎝ R3 + R3A ⎠ More than one output can be summed into R1 because the output of the transconductance amplifier is a current. This is the simplest way to make a video mixer. 1228fc LT1228 TYPICAL APPLICATIO S Video Fader 1k 3 VIN1 + gm 1 2 – 5 100Ω 1k 10k 5.1k V– 5V 3k –5V 10k 1k VIN2 1k 100Ω 3 10k 5.1k VS = ±5V + gm 5 1 LOGIC INPUT LT1228 • TA12 2 – The video fader uses the transconductance amplifiers from two LT1228s in the feedback loop of another current feedback amplifier, the LT1223. The amount of signal from each input at the output is set by the ratio of the set currents of the two LT1228s, not by their absolute value. The bandwidth of the current feedback amplifier is inversely proportional to the set current in this configuration. Therefore, the set currents remain high over most of the pot’s range, keeping the bandwidth over 15MHz even when the signal is attenuated 20dB. The pot is set up to completely turn off one LT1228 at each end of the rotation. U Video DC Restore (Clamp) Circuit NOT NECESSARY IF THE SOURCE RESISTANCE IS LESS THAN 50Ω 200Ω 3 + LT1223 CFA VOUT 1000pF V+ 7 gm 1 5 10k RF 0.01µF + – 4 – 2 + CFA 6 VOUT 8 – RG VIDEO INPUT 2N3906 3k RESTORE LT1228 • TA13 The video restore (clamp) circuit restores the black level of the composite video to zero volts at the beginning of every line. This is necessary because AC coupled video changes DC level as a function of the average brightness of the picture. DC restoration also rejects low frequency noise such as hum. The circuit has two inputs: composite video and a logic signal. The logic signal is high except during the back porch time right after the horizontal sync pulse. While the logic is high, the PNP is off and ISET is zero. With ISET equal to zero the feedback to Pin 2 has no affect. The video input drives the noninverting input of the current feedback amplifier whose gain is set by RF and RG. When the logic signal is low, the PNP turns on and ISET goes to about 1mA. Then the transconductance amplifier charges the capacitor to force the output to match the voltage at Pin 3, in this case zero volts. This circuit can be modified so that the video is DC coupled by operating the amplifier in an inverting configuration. Just ground the video input shown and connect RG to the video input instead of to ground. 1228fc 15 LT1228 TYPICAL APPLICATIO S Single Supply Wien Bridge Oscillator 100Ω 2N3906 V+ 10kΩ 3 10kΩ 6V TO 30V V+ 7 470Ω + 10µF + gm 5 1 + CFA 6 2 – 4 8 – RF 680Ω 1.8k 160Ω 1000pF RG 20Ω + 10µF + 10µF 1000pF 160Ω 50Ω 4.7µH 30pF f = 1MHz VO = 6dBm (450mVRMS) 2nd HARMONIC = –38dBc 3rd HARMONIC = –54 dBc FOR 5V OPERATION SHORT OUT 100Ω RESISTOR In this application the LT1228 is biased for operation from a single supply. An artificial signal ground at half supply voltage is generated with two 10k resistors and bypassed with a capacitor. A capacitor is used in series with R G to set the DC gain of the current feedback amplifier to unity. The transconductance amplifier is used as a variable resistor to control gain. A variable resistor is formed by driving the inverting input and connecting the output back to it. The equivalent resistor value is the inverse of the g m. This works with the 1.8k resistor to make a variable attenuator. The 1MHz oscillation frequency is set by the Wien bridge network made up of two 1000pF capacitors and two 160Ω resistors. For clean sine wave oscillation, the circuit needs a net gain of one around the loop. The current feedback amplifier has a gain of 34 to keep the voltage at the transconductance amplifier input low. The Wien bridge has an attenuation of 3 at resonance; therefore the attenuation of the 1.8k 16 U resistor and the transconductance amplifier must be about 11, resulting in a set current of about 600µA at oscillation. At start-up there is no set current and therefore no attenuation for a net gain of about 11 around the loop. As the output oscillation builds up it turns on the PNP transistor which generates the set current to regulate the output voltage. 0.1µF 12MHz Negative Resistance LC Oscillator V+ 9.1k 3 1k 2 51Ω VO + gm 7 1 5 8 V– + CFA 6 – 4 51Ω VO – 1k 750Ω 50Ω 2N3906 4.3k 330Ω 2N3904 LT1228 • TA14 10k 0.1µF VO = 10dB V– AT VS = ±5V ALL HARMONICS 40dB DOWN AT VS = ±12V ALL HARMONICS 50dB DOWN LT1228 • TA15 This oscillator uses the transconductance amplifier as a negative resistor to cause oscillation. A negative resistor results when the positive input of the transconductance amplifier is driven and the output is returned to it. In this example a voltage divider is used to lower the signal level at the positive input for less distortion. The negative resistor will not DC bias correctly unless the output of the transconductance amplifier drives a very low resistance. Here it sees an inductor to ground so the gain at DC is zero. The oscillator needs negative resistance to start and that is provided by the 4.3k resistor to Pin 5. As the output level rises it turns on the PNP transistor and in turn the NPN which steals current from the transconductance amplifier bias input. 1228fc LT1228 TYPICAL APPLICATIO S Filters Single Pole Low/High/Allpass Filter VIN LOWPASS INPUT R3A 1k R3 120Ω VIN HIGHPASS INPUT PHASE SHIFT (DEGREES) Using the variable transconductance of the LT1228 to make variable filters is easy and predictable. The most straight forward way is to make an integrator by putting a capacitor at the output of the transconductance amp and buffering it with the current feedback amplifier. Because the input bias current of the current feedback amplifier must be supplied by the transconductance amplifier, the set current should not be operated below 10 µA. This limits the filters to about a 100:1 tuning range. The Single Pole circuit realizes a single pole filter with a corner frequency (fC) proportional to the set current. The U 3 + gm 1 5 C 330pF + CFA 6 VOUT 2 – 8 – RF 1k ISET RG 1k R2A 1k R2 120Ω fC = R +1 R2 10 I × SET × F × C RG R2 + R2A 2π LT1228 • TA16 fC = 109 ISET FOR THE VALUES SHOWN Allpass Filter Phase Response 0 1mA SET CURRENT –45 –90 –135 100µA SET CURRENT –180 10k 100k 1M 10M LT1228 • TA17 FREQUENCY (Hz) values shown give a 100kHz corner frequency for 100 µA set current. The circuit has two inputs, a lowpass filter input and a highpass filter input. To make a lowpass filter, ground the highpass input and drive the lowpass input. Conversely for a highpass filter, ground the lowpass input and drive the highpass input. If both inputs are driven, the result is an allpass filter or phase shifter. The allpass has flat amplitude response and 0° phase shift at low frequencies, going to –180° at high frequencies. The allpass filter has –90° phase shift at the corner frequency. 1228fc 17 LT1228 TYPICAL APPLICATIO S Voltage Controlled State Variable Filter 1k LT1006 10k VC 180Ω 2N3906 100pF VIN 100Ω fO = 100kHz AT VC = 0V fO = 200kHz AT VC = 1V fO = 400kHz AT VC = 2V fO = 800kHz AT VC = 3V fO = 1.6MHz AT VC = 4V The state variable filter has both lowpass and bandpass outputs. Each LT1228 is configured as a variable integrator whose frequency is set by the attenuators, the capacitors and the set current. Because the integrators have both positive and negative inputs, the additional op amp normally required is not needed. The input attenuators set the circuit up to handle 3VP–P signals. The set current is generated with a simple circuit that gives logarithmic voltage to current control. The two PNP transistors should be a matched pair in the same package for 18 U + – 51k 3k –5V 5V 3.3k 3 7 3k 5 gm 2 + – 4 1 + CFA 6 BANDPASS OUTPUT 18pF –5V 3.3k 100Ω 5V 3 100Ω 2 7 3.3k 8 – 1k + gm 5 1 + CFA 6 LOWPASS OUTPUT – 4 18pF –5V 3.3k 8 – 1k LT1228 • TA18 best accuracy. If discrete transistors are used, the 51k resistor should be trimmed to give proper frequency response with VC equal zero. The circuit generates 100 µA for VC equal zero volts and doubles the current for every additional volt. The two 3k resistors divide the current between the two LT1228s. Therefore the set current of each amplifier goes from 50µA to 800µA for a control voltage of 0V to 4V. The resulting filter is at 100kHz for VC equal zero, and changes it one octave/V of control input. 1228fc LT1228 PACKAGE DESCRIPTIO CORNER LEADS OPTION (4 PLCS) .300 BSC (7.62 BSC) .023 – .045 (0.584 – 1.143) HALF LEAD OPTION .008 – .018 (0.203 – 0.457) .045 – .068 (1.143 – 1.650) FULL LEAD OPTION 0° – 15° NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP/PLATE OR TIN PLATE LEADS .300 – .325 (7.620 – 8.255) .008 – .015 (0.203 – 0.381) .255 ± .015* (6.477 ± 0.381) ( +.035 .325 –.015 +0.889 8.255 –0.381 ) INCHES MILLIMETERS *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010 INCH (0.254mm) NOTE: 1. DIMENSIONS ARE .010 – .020 × 45° (0.254 – 0.508) .008 – .010 (0.203 – 0.254) 0°– 8° TYP .050 BSC .016 – .050 (0.406 – 1.270) NOTE: 1. DIMENSIONS IN .245 MIN INCHES (MILLIMETERS) 2. DRAWING NOT TO SCALE 3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .006" (0.15mm) .030 ±.005 TYP RECOMMENDED SOLDER PAD LAYOUT Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U J8 Package 8-Lead CERDIP (Narrow .300 Inch, Hermetic) (Reference LTC DWG # 05-08-1110) .405 (10.287) MAX 8 7 6 5 .005 (0.127) MIN .200 (5.080) MAX .015 – .060 (0.381 – 1.524) .025 (0.635) RAD TYP 1 2 3 .220 – .310 (5.588 – 7.874) 4 .045 – .065 (1.143 – 1.651) .014 – .026 (0.360 – 0.660) .100 (2.54) BSC .125 3.175 MIN J8 0801 OBSOLETE PACKAGE N8 Package 8-Lead PDIP (Narrow .300 Inch) (Reference LTC DWG # 05-08-1510) .400* (10.160) MAX 8 7 6 5 .045 – .065 (1.143 – 1.651) .130 ± .005 (3.302 ± 0.127) .065 (1.651) TYP .120 (3.048) .020 MIN (0.508) MIN .018 ± .003 (0.457 ± 0.076) N8 1002 1 2 3 4 .100 (2.54) BSC S8 Package 8-Lead Plastic Small Outline (Narrow .150 Inch) (Reference LTC DWG # 05-08-1610) .189 – .197 (4.801 – 5.004) NOTE 3 .045 ±.005 8 7 6 5 .053 – .069 (1.346 – 1.752) .004 – .010 (0.101 – 0.254) .160 ±.005 .228 – .244 (5.791 – 6.197) .150 – .157 (3.810 – 3.988) .014 – .019 NOTE 3 (0.355 – 0.483) TYP .050 (1.270) BSC 1 2 3 4 SO8 0303 1228fc 19 LT1228 TYPICAL APPLICATIO S RF AGC Amplifier (Leveling Loop) 15V RF INPUT 0.6VRMS to 1.3VRMS 25MHz 10k 100 Ω 2 3 7 gm 1 300Ω 8 Inverting Amplifier with DC Output Less Than 5mV 5V V+ 2 – gm 7 1 5 4 V– R5 3 + + + CFA 6 VO 2 CARRIER INPUT 30mV 4.7µF 100 µF 8 – RF 1k LT1228 • TA21 RELATED PARTS PART NUMBER LT1227 LT1251/LT1256 LT1399 DESCRIPTION 140MHz Current Feedback Amplifier 40MHz Video Fader 400MHz Current Feedback Amplifier COMMENTS 1100V/µs Slew Rate, 0.01% Differential Gain, 0.03% Differential Phase Accurate Linear Gain Control: ±1% Typ, ±3% Max 800V/µs Slew Rate, 80mA Output Current 20 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com + VS = ±5V, R5 = 3.6k RG VS = ±15V, R5 = 13.6k 1k VOUT MUST BE LESS THAN 200mVP–P FOR LOW OUTPUT OFFSET VIN BW = 30Hz TO 20MHz INCLUDES DC U + – + CFA 5 – OUTPUT 2VP–P 4 470Ω 0.01µF 10k –15V 10k 4pF 0.01µF 10Ω 10k 15V – A3 LT1006 10k 100k AMPLITUDE ADJUST 4.7k –15V 1N4148’s COUPLE THERMALLY LT1004 1.2V LT1228 • TA20 + –15V Amplitude Modulator + 3 4.7µF + gm 7 1 5 4 10k 1k 8 + CFA 6 – – RF 750Ω VOUT 0dBm(230mV) AT MODULATION = 0V –5V MODULATION INPUT ≤ 8VP–P RG 750Ω LT1228 • TA22 1228fc LT 0107 REV C • PRINTED IN USA © LINEAR TECHNOLOGY CORPORATION 1994