LT1578C

LT1578/LT1578-2.5 1.5A, 200kHz Step-Down Switching Regulator FEATURES s s s s s s s s s s s s DESCRIPTIO 1.5A Switch Current High Efficiency—Low Loss 0.2Ω Switch Constant 200kHz Switching Frequency 4V to 15V Input VoltageRange Minimum Output: 1.21V Low Supply Current: 1.9mA Low Shutdown Current: 20µA Easily Synchronizable Up to 400kHz Cycle-by-Cycle Current Limit Reduced EMI Generation Low Thermal Resistance SO-8 Package Uses Small Low Value Inductors The LT ®1578 is a 200kHz monolithic buck mode switching regulator. A 1.5A switch is included on the die along with all the necessary oscillator, control and logic circuitry. The topology is current mode for fast transient response and good loop stability. The LT1578 is a modified version of the LT1507 that has been optimized for noise sensitive applications. It will operate over a 4V to 15V input range. In addition, the reference voltage has been lowered to allow the device to produce output voltages down to 1.2V. Quiescent current has been reduced by a factor of two. Switch on resistance has been reduced by 30%. Switch transition times have been slowed to reduce EMI generation. The oscillator frequency has been reduced to 200kHz to maintain high efficiency over a wide output current range. The pinout has been changed to improve PC layout by allowing the high current, high frequency switching circuitry to be easily isolated from low current, noise sensitive control circuitry. The new SO-8 package includes a fused ground lead that significantly reduces the thermal resistance of the device to extend the ambient operating temperature range. Standard surface mount external parts can be used including the inductor and capacitors. , LTC and LT are registered trademarks of Linear Technology Corporation. APPLICATIO S s s s s Portable Computers Battery-Powered Systems Battery Chargers Distributed Power Systems TYPICAL APPLICATION Efficiency vs Load Current 3.3V Buck Converter INPUT 5V TO 15V C3* 10µF TO 50µF 90 85 + VIN BOOST C2 0.33µF VSW EFFICIENCY (%) L1** 15µH D2 1N914 OUTPUT** 3.3V, 1.25A 80 75 70 65 60 55 50 0 0.25 0.50 0.75 1.00 LOAD CURRENT (A) 1.25 1.50 VOUT = 3.3V VIN = 5V L = 25µH LT1578 OPEN = ON SHDN GND FB VC CC 100pF D1 1N5818 R1 8.66k R2 4.99k * RIPPLE CURRENT RATING ≥ IOUT/2 ** INCREASE L1 TO 30µH FOR LOAD CURRENTS ABOVE 0.6A AND TO 60µH ABOVE 1A SEE APPLICATIONS INFORMATION + C1 100µF, 10V SOLID TANTALUM 1578 TA01 U 1578 TA02 U U 1 LT1578/LT1578-2.5 ABSOLUTE MAXIMUM RATINGS (Note 1) PACKAGE/ORDER INFORMATION TOP VIEW VSW 1 VIN 2 BOOST 3 GND 4 8 SYNC 7 SHDN 6 FB/SENSE 5 VC Input Voltage .......................................................... 16V BOOST Pin Above Input Voltage ............................. 10V SHDN Pin Voltage ..................................................... 7V SENSE Pin Voltage .................................................... 4V FB Pin Voltage (Adjustable Part) ............................ 3.5V FB Pin Current (Adjustable Part) ............................ 1mA SYNC Pin Voltage ..................................................... 7V Operating Junction Temperature Range LT1578C ............................................... 0°C to 125° C LT1578I ........................................... – 40°C to 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C ORDER PART NUMBER LT1578CS8 LT1578IS8 LT1578CS8-2.5 LT1578IS8-2.5 S8 PART MARKING 1578 1578I 157825 578I25 S8 PACKAGE 8-LEAD PLASTIC SO θJA = 80°C/ W WITH FUSED GROUND PIN CONNECTED TO GROUND PLANE OR LARGE LANDS Consult factory for Military grade parts. The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TJ = 25°C. VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted. PARAMETER Feedback Voltage Sense Voltage (Fixed 2.5) All Conditions Sense Pin Resistance Reference Voltage Line Regulation Feedback Input Bias Current Error Amplifier Voltage Gain (Notes 2, 10) Error Amplifier Transconductance (Note 10) VC Pin to Switch Current Transconductance Error Amplifier Source Current Error Amplifier Sink Current VC Pin Switching Threshold VC Pin High Clamp Switch Current Limit Slope Compensation (Note 8) Switch On Resistance (Note 7) Maximum Switch Duty Cycle Minimum Switch Duty Cycle (Note 9) Switch Frequency Switch Frequency Line Regulation Frequency Shifting Threshold on FB Pin Minimum Input Voltage (Note 3) Minimum Boost Voltage (Note 4) 4.3V ≤ VIN ≤ 15V q q q ELECTRICAL CHARACTERISTICS CONDITIONS All Conditions q MIN 1.195 1.18 2.46 2.44 5.7 TYP 1.21 2.5 9.5 0.01 0.5 400 1050 1.5 110 130 0.8 2.1 2 0.3 0.2 94 94 8 200 0 0.74 4.0 2.3 MAX 1.225 1.24 2.54 2.56 13.7 0.03 2 1300 1700 190 200 UNITS V V V V kΩ %/V µA µMho µMho A/ V µA µA V V A A Ω Ω % % % kHz kHz %/ V V V V ∆I (VC) = ± 10µA q 200 800 400 40 50 VFB = 1.1V VFB = 1.4V Duty Cycle = 0 VC Open, VFB = 1.1V, DC ≤ 50% DC = 80% ISW = 1.5A VFB = 1.1V q q q 1.5 3.5 0.35 0.45 q q 90 86 180 160 0.4 VC Set to Give 50% Duty Cycle q 4.3V ≤ VIN ≤ 15V ∆f = 10kHz ISW ≤ 1.5A q q q q 220 240 0.15 1.0 4.3 3.0 2 U W U U WW W LT1578/LT1578-2.5 The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TJ = 25°C. VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted. PARAMETER Boost Current (Note 5) VIN Supply Current (Note 6) Shutdown Supply Current Lockout Threshold Shutdown Thresholds Synchronization Threshold Synchronizing Range SYNC Pin Input Resistance Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: Gain is measured with a VC swing equal to 200mV above the switching threshold level to 200mV below the upper clamp level. Note 3: Minimum input voltage is not measured directly, but is guaranteed by other tests. It is defined as the voltage where internal bias lines are still regulated so that the reference voltage and oscillator frequency remain constant. Actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. See Applications Information. Note 4: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. Note 5: Boost current is the current flowing into the boost pin with the pin held 5V above input voltage. It flows only during switch on time. Note 6: Input supply current is the bias current drawn by the input pin with switching disabled. CONDITIONS ISW = 0.5A ISW = 1.5A VSHDN = 0V, VIN ≤ 15V, VSW = 0V, VC Open q ELECTRICAL CHARACTERISTICS MIN q q q TYP 9 27 1.9 20 2.42 0.37 0.45 1.5 40 VC Open VC Open Device Shutting Down Device Starting Up q q q 2.34 0.13 0.25 250 MAX 18 50 2.7 50 75 2.50 0.60 0.7 2.2 400 UNITS mA mA mA µA µA V V V V kHz kΩ Note 7: Switch on resistance is calculated by dividing VIN to VSW voltage by the forced current (1.5A). See Typical Performance Characteristics for the graph of switch voltage at other currents. Note 8: Slope compensation is the current subtracted from the switch current limit at 80% duty cycle. See Maximum Output Load Current in the Applications Information section for further details. Note 9: Minimum on-time is 400ns typical. For a 200kHz operating frequency this means the minimum duty cycle is 8%. In frequency foldback mode, the effective duty cycle will be less than 8%. Note 10: Transconductance and voltage gain refer to the internal amplifier exclusive of the voltage divider. To calculate gain and transconductance referred to the sense pin on the fixed voltage parts, divide values shown by the ratio 2.5/1.21. TYPICAL PERFORMANCE CHARACTERISTICS Switch Voltage Drop 0.5 125°C 25°C –20°C SWITCH PEAK CURRENT (A) 0.4 SWITCH VOLTAGE (V) 0.3 1.5 MINIMUM FEEDBACK VOLTAGE (V) 0.2 0.1 0 0 0.25 0.50 0.75 1.00 SWITCH CURRENT (A) 1.25 1.50 UW 1576 G01 Switch Peak Current Limit 2.5 TYPICAL 2.0 1.22 1.23 Feedback Pin Voltage 1.21 1.0 0.5 1.20 0 0 20 60 40 DUTY CYCLE (%) 80 100 1576 G02 1.19 0 25 50 75 100 –50 –25 JUNCTION TEMPERATURE (°C) 125 1576 G03 3 LT1578/LT1578-2.5 TYPICAL PERFORMANCE CHARACTERISTICS Shutdown Pin Bias Current (VSHDN = Lockout Threshold) 4 AT 2.44V STANDBY THRESHOLD (CURRENT FLOWS OUT OF PIN) SHDN PIN CURRENT (µA) SHDN PIN CURRENT (µA) 3 140 120 100 80 60 40 20 CURRENT REQUIRED TO FORCE SHUTDOWN (FLOWS OUT OF PIN). AFTER SHUTDOWN, CURRENT DROPS TO A FEW µA 125 SHUTDOWN PIN VOLTAGE (V) 2 1 0 0 25 50 75 100 –50 –25 JUNCTION TEMPERATURE (°C) Standby Thresholds 2.46 INPUT SUPPLY CURRENT (µA) SHUTDOWN PIN VOLTAGE (V) 20 VSHDN = 0V 15 INPUT SUPPLY CURRENT (µA) 2.45 2.44 ON 2.43 STANDBY 2.42 2.41 2.40 50 100 –50 –25 25 75 0 JUNCTION TEMPERATURE (°C) Error Amplifier Transconductance 2000 PHASE 1500 GAIN (µMho) TRANSCONDUCTANCE (µMho) 150 GAIN 100 VC 1200 1000 800 600 400 200 1000 500 VFB 1 × 10–3 ( ) ROUT 570k COUT 2.4pF 50 0 ERROR AMPLIFIER EQUIVALENT CIRCUIT 0 SWITCHING FREQUENCY (kHz) OR CURRENT (µA) RLOAD = 50Ω –500 10 100 1k 10k FREQUENCY (Hz) 100k –50 1M 1576 G09 4 UW 1576 G04 Shutdown Pin Bias Current (VSHDN = Shutdown Threshold) 180 160 0.8 0.7 0.6 0.5 0.4 Shutdown Thresholds START-UP SHUTDOWN 0.3 0.2 0.1 0 0 50 25 75 100 –50 –25 JUNCTION TEMPERATURE (°C) 125 125 0 0 25 50 75 100 –50 –25 JUNCTION TEMPERATURE (°C) 1576 G05 1576 G06 Shutdown Supply Current 25 30 25 20 15 10 5 0 Shutdown Supply Current VIN = 10V 10 5 0 125 0 5 10 INPUT VOLTAGE (V) 15 1576 G08 0 0.1 0.2 0.3 SHUTDOWN VOLTAGE (V) 0.4 1576 G010 1576 G07 Error Amplifier Transconductance 200 1600 1400 200 250 Frequency Foldback SWITCHING FREQUENCY PHASE (DEG) 150 100 50 FEEDBACK PIN CURRENT 0 25 75 100 0 50 –50 –25 JUNCTION TEMPERATURE (°C) 125 0 0 1.0 0.5 1.5 FEEDBACK VOLTAGE (V) 2.0 1576 G12 1576 G11 LT1578/LT1578-2.5 TYPICAL PERFORMANCE CHARACTERISTICS Switching Frequency 240 SWITCH CURRENT LIMIT (A) 2.0 1.5 1.0 0.5 INPUT VOLTAGE (V) 220 FREQUENCY (kHz) 200 180 160 0 25 50 75 100 –50 –25 JUNCTION TEMPERATURE (°C) Maximum Output Current at VOUT = 5V 1.6 1.4 L = 60µH L = 30µH L = 15µH 1.6 1.4 OUTPUT CURRENT (A) OUTPUT CURRENT (A) OUTPUT CURRENT (A) 1.2 1.0 0.8 0.6 0.4 0.2 0 6 9 12 INPUT VOLTAGE (V) 1578 G15 BOOST Pin Current 30 25 BOOST PIN CURRENT (mA) 20 15 10 5 0 1.0 THRESHOLD VOLTAGE (V) 0 0.25 Kool Mµ is a registered trademark of Magnetics, Inc. Metglas is a registered trademark of AlliedSignal, Inc. UW 1576 G13 Switch Current Limit Foldback 3.0 2.5 4.25 4.50 Minimum Input Voltage to Start with 3.3V Output 4.00 3.75 0 125 0 1.0 0.4 0.6 0.8 0.2 FEEDBACK PIN VOLTAGE (V) 1.2 1578 G19 3.50 1 10 100 LOAD CURRENT (mA) 1000 1576 G14 Maximum Output Current at VOUT = 3.3V 1.6 L = 60µH 1.4 L = 30µH L = 15µH 1.2 1.0 0.8 0.6 0.4 0.2 0 4 6 8 10 12 14 1578 G16 Maximum Output Current at VOUT = 2.5V L = 60µH L = 30µH L = 15µH 1.2 1.0 0.8 0.6 0.4 0.2 0 15 4 6 8 10 12 14 1578 G17 INPUT VOLTAGE (V) INPUT VOLTAGE (V) VC Pin Shutdown Threshold 0.8 0.6 0.4 0.2 0.50 0.75 1.00 SWITCH CURRENT (A) 1.25 1.50 0 0 25 50 75 100 –50 –25 JUNCTION TEMPERATURE (°C) 125 1576 G20 1576 G21 5 LT1578/LT1578-2.5 PIN FUNCTIONS VSW (Pin 1): The switch pin is the emitter of the on-chip power NPN switch. This pin is driven up to the input pin voltage during switch on time. Inductor current drives the switch pin negative during switch off time. Negative voltage is clamped with the external catch diode. Maximum negative switch voltage allowed is – 0.8V. VIN (Pin 2): This is the collector of the on-chip power NPN switch. This pin powers the internal circuitry and internal regulator. At NPN switch on and off, high dI/dt edges occur through this pin. Keep the external bypass and catch diode close to this pin. Trace inductance in this path will create a voltage spike at switch off, adding to the VCE voltage across the internal NPN. BOOST (Pin 3): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. Without this added voltage, the typical switch voltage loss would be about 1.5V. The additional boost voltage allows the switch to saturate with its voltage drop approximating that of a 0.2Ω FET structure. Efficiency improves from 75% for conventional bipolar designs to > 88% for the LT1578. GND (Pin 4): The GND pin connection needs consideration for two reasons. First, it acts as the reference for the regulated output, so load regulation will suffer if the “ground” end of the load is not at the same voltage as the GND pin of the IC. This condition will occur when load current or other currents flow through metal paths between the GND pin and the load ground point. Keep the ground path short between the GND pin and the load and use a ground plane when possible. The second consideration is EMI caused by GND pin current spikes. Internal capacitance between the VSW pin and the GND pin creates very narrow (100MHz oscilloscope must be used, and waveforms should be observed on the leads of the package. This switch off spike will also cause the SW node to go below ground. The LT1578 has special circuitry inside which mitigates this problem, but negative voltages over 1V lasting longer than 10ns should be avoided. Note that 100MHz oscilloscopes are barely fast enough to see the details of the falling edge overshoot in Figure 7. A second, much lower frequency ringing is seen during switch off time if load current is low enough to allow the inductor current to fall to zero during part of the switch off time (see Figure 8). Switch and diode capacitance resonate with the inductor to form damped ringing at 1MHz to 10 MHz. This ringing is not harmful to the regulator and it has not been shown to contribute significantly to EMI. Any attempt to damp it with an RC snubber will slightly degrade efficiency. INPUT BYPASSING AND VOLTAGE RANGE RISE AND FALL WAVEFORMS ARE SUPERIMPOSED (PULSE WIDTH IS NOT 350ns) 5V/DIV 50ns/DIV 1578 F07 Figure 7. Switch Node Response 5V/DIV SWITCH NODE VOLTAGE 50mA/DIV INDUCTOR CURRENT 1µs/DIV 1578 F08 Figure 8. Discontinuous Mode Ringing U W U U Input Bypass Capacitor Step-down converters draw current from the input supply in pulses. The average height of these pulses is equal to load current, and the duty cycle is equal to VOUT/ VIN. Rise and fall times of the current are very fast. A local bypass capacitor across the input supply is necessary to ensure proper operation of the regulator and minimize the ripple current fed back into the input supply. The capacitor also forces switching current to flow in a tight local loop, minimizing EMI. Do not cheat on the ripple current rating of the input bypass capacitor, but also do not be overly concerned with the value in microfarads. The input capacitor is intended to absorb all the switching current ripple, which can have an RMS value as high as one half of the load current. Ripple current ratings on the capacitor must be observed to ensure reliable operation. In many cases it is necessary to parallel two capacitors to obtain the required ripple rating. Both capacitors must be of the same value and manufacturer to guarantee power sharing. The actual value of the capacitor in microfarads is not particularly important 17 LT1578/LT1578-2.5 APPLICATIONS INFORMATION because at 200kHz, any value above 15µF is essentially resistive. RMS ripple current rating is the critical parameter. Actual RMS current can be calculated from: series for instance, see Table 3), but even these units may fail if the input voltage surge approaches the maximum voltage rating of the capacitor. AVX recommends derating capacitor voltage by 2:1 for high surge applications. The highest voltage rating is 50V, so 25V may be a practical input voltage upper limit when using solid tantalum capacitors for input bypassing. Larger capacitors may be necessary when the input voltage is very close to the minimum specified on the data sheet. Small voltage dips during switch on time are not normally a problem, but at very low input voltage they may cause erratic operation because the input voltage drops below the minimum specification. Problems can also occur if the input-to-output voltage differential is near minimum. The amplitude of these dips is normally a function of capacitor ESR and ESL because the capacitive reactance is small compared to these terms. ESR tends to be the dominate term and is inversely related to physical capacitor size within a given capacitor type. SYNCHRONIZING The SYNC pin is used to synchronize the internal oscillator to an external signal. The SYNC input must pass from a logic level low, through the maximum synchronization threshold with a duty cycle between 10% and 90%. The input can be driven directly from a logic level output. The synchronizing range is equal to initial operating frequency up to 400kHz. This means that minimum practical sync frequency is equal to the worst-case high self-oscillating frequency (250kHz), not the typical operating frequency of 200kHz. Caution should be used when synchronizing above 280kHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. This type of subharmonic switching only occurs at input voltages less than twice output voltage. Higher inductor values will tend to eliminate this problem. See Frequency Compensation section for a discussion of an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensation. Application Note 19 has more details on the theory of slope compensation. IRIPPLE RMS = IOUT VOUT VIN − VOUT / VIN () ( ) The term inside the radical has a maximum value of 0.5 when input voltage is twice output, and stays near 0.5 for a relatively wide range of input voltages. It is common practice therefore to simply use the worst-case value and assume that RMS ripple current is one half of load current. At maximum output current of 1.5A for the LT1578, the input bypass capacitor should be rated at 0.75A ripple current. Note however, that there are many secondary considerations in choosing the final ripple current rating. These include ambient temperature, average versus peak load current, equipment operating schedule, and required product lifetime. For more details, see Application Notes 19 and 46, and Design Note 95. Input Capacitor Type Some caution must be used when selecting the type of capacitor used at the input to regulators. Aluminum electrolytics are lowest cost, but are physically large to achieve adequate ripple current rating, and size constraints (especially height) may preclude their use. Ceramic capacitors are now available in larger values, and their high ripple current and voltage rating make them ideal for input bypassing. Cost is fairly high and footprint may also be somewhat large. Solid tantalum capacitors would be a good choice, except that they have a history of occasional spectacular failures when they are subjected to large current surges during power-up. The capacitors can short and then burn with a brilliant white light and lots of nasty smoke. This phenomenon occurs in only a small percentage of units, but it has led some OEMs to forbid their use in high surge applications. The input bypass capacitors of regulators can see these high surges when a battery or high capacitance source is connected. Several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (AVX TPS 18 U 2 W U U LT1578/LT1578-2.5 APPLICATIONS INFORMATION At power-up, when VC is being clamped by the FB pin (see Figure 2, Q2), the sync function is disabled. This allows the frequency foldback to operate in the shorted output condition. During normal operation, switching frequency is controlled by the internal oscillator until the FB pin reaches 0.7V, after which the SYNC pin becomes operational. If no synchronization is required, this pin should be connected to ground. THERMAL CALCULATIONS Power dissipation in the LT1578 chip comes from four sources: switch DC loss, switch AC loss, boost circuit current, and input quiescent current. The following formulas show how to calculate each of these losses. These formulas assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. Switch loss: PSW = RSW IOUT ( ) (VOUT) + 60ns(IOUT)(VIN)(f) 2 VIN Boost current loss: PBOOST = VOUT IOUT / 50 VIN 2 ( ) Quiescent current loss: PQ = VIN  0.55 • 10−3  + VOUT 1.6 • 10−3      2   VOUT  0.004   + VIN ( ) RSW = Switch resistance (≈ 0.2Ω) 60ns = Equivalent switch current/voltage overlap time f = Switch frequency Example: with VIN = 10V, VOUT = 5V and IOUT = 1A: U W U U (0.2)(1) (5) +  60 • 10−9 (1)(10)2 00 • 10 3 PSW = 2  10 = 0.1 + 012 = 0.22W . 2    (5) (1/ 50) = 0.05W PBOOST = (5) (0.004) PQ = 10 0.55 • 10−3  + 5 1.6 • 10−3  + 2 10     10 = 0.02W Total power dissipation is 0.22 + 0.05 + 0.02 = 0.29W. Thermal resistance for LT1578 package is influenced by the presence of internal or backside planes. With a full plane under the SO package, thermal resistance will be about 80°C/W. No plane will increase resistance to about 120°C/W. To calculate die temperature, add in worst-case ambient temperature: TJ = TA + θJA (PTOT) With the SO-8 package (θJA = 80°C/W), at an ambient temperature of 50°C, TJ = 50 + 80 (0.29) = 73.2°C Die temperature is highest at low input voltage, so use lowest continuous input operating voltage for thermal calculations. FREQUENCY COMPENSATION Loop frequency compensation of switching regulators can be a rather complicated problem because the reactive components used to achieve high efficiency also introduce multiple poles into the feedback loop. The inductor and output capacitor on a conventional step-down converter actually form a resonant tank circuit that can exhibit peaking and a rapid 180° phase shift at the resonant frequency. By contrast, the LT1578 uses a “current mode” architecture to help alleviate the phase shift created by the inductor. The basic connections are shown in Figure 9. Figure 10 shows a Bode plot of the phase and gain of the power section of the LT1578, measured from the VC pin to 19 LT1578/LT1578-2.5 APPLICATIONS INFORMATION the output. Gain is set by the 1.5A/V transconductance of the LT1578 power section and the effective complex impedance from output to ground. Gain rolls off smoothly above the 160Hz pole frequency set by the 100µF output capacitor. Phase drop is limited to about 85°. Phase recovers and gain levels off at the zero frequency (≈16kHz) set by capacitor ESR (0.1Ω). Error amplifier transconductance phase and gain are shown in Figure 11. The error amplifier can be modeled as a transconductance of 1000µMho, with an output impedance of 570kΩ in parallel with 2.4pF. In all practical applications, the compensation network from the VC pin to ground has a much lower impedance than the output impedance of the amplifier at frequencies above 200Hz. LT1578 CURRENT MODE POWER STAGE gm = 1.5A/V VSW ERROR AMPLIFIER FB ESR R1 GAIN (µMho) OUTPUT 1.21V GND VC + C1 R2 CF RC CC Figure 9. Model for Loop Response 40 VIN = 10V VOUT = 5V IOUT = 500mA 20 GAIN PHASE (DEG) GAIN (dB) 40 0 LOOP GAIN (dB) PHASE 0 –40 –20 –80 –40 10 100 1k 10k FREQUENCY (Hz) –120 100k 1578 F07 Figure 10. Response from VC Pin to Output 20 U W U – + U This means that the error amplifier characteristics themselves do not contribute excess phase shift to the loop, and the phase/gain characteristics of the error amplifier section are completely controlled by the external compensation network. In Figure 12, full loop phase/gain characteristics are shown with a compensation capacitor of 100pF, giving the error amplifier a pole at 2.8kHz, with phase rolling off to 90° and staying there. The overall loop has a gain of 66dB at low frequency, rolling off to unity-gain at 58kHz. The phase plot shows a two-pole characteristic until the ESR of the output capacitor brings it back to single pole above 16kHz. Phase margin is about 77° at unity-gain. 2000 PHASE 1500 GAIN 200 150 PHASE (DEG) 1000 100 500 VFB 1 × 10–3 ( ) ROUT 570k VC COUT 2.4pF 50 0 ERROR AMPLIFIER EQUIVALENT CIRCUIT 0 RLOAD = 50Ω –500 10 100 1k 10k FREQUENCY (Hz) 100k –50 1M 1578 F11 1578 F09 Figure 11. Error Amplifier Gain and Phase 80 180 60 PHASE 135 LOOP PHASE (DEG) 40 VIN = 10V VOUT = 5V IOUT = 500mA COUT = 100µF 10V, AVX TPS CC = 100pF L = 30µH 10 100 1k 10k FREQUENCY (Hz) 90 20 45 GAIN 0 0 –20 100k –45 1M 1578 F12 Figure 12. Overall Loop Characteristics LT1578/LT1578-2.5 APPLICATIONS INFORMATION Analog experts will note that around 7kHz, phase dips close to the zero phase margin line. This is typical of switching regulators, especially those that operate over a wide range of loads. This region of low phase is not a problem as long as it does not occur near unity-gain. In practice, the variability of output capacitor ESR tends to dominate all other effects with respect to loop response. Variations in ESR will cause unity-gain to move around, but at the same time phase moves with it so that adequate phase margin is maintained over a very wide range of ESR (≥ ± 3:1). What About a Resistor in the Compensation Network? It is common practice in switching regulator design to add a “zero” to the error amplifier compensation to increase loop phase margin. This zero is created in the external network in the form of a resistor (RC) in series with the compensation capacitor. Increasing the size of this resistor generally creates better and better loop stability, but there are two limitations on its value. First, the combination of output capacitor ESR and a large value for RC may cause loop gain to stop rolling off altogether, creating a gain margin problem. An approximate formula for RC where gain margin falls to zero is: R C Loop Gain = 1 = ( ) (GMP)(GMA)(ESR)(1.21) VOUT GMP = Transconductance of power stage = 1.5A/V GMA = Error amplifier transconductance = 1(10–3) ESR = Output capacitor ESR 1.21 = Reference voltage With VOUT = 5V and ESR = 0.1Ω, a value of 27.5k for RC would yield zero gain margin, so this represents an upper limit. There is a second limitation however which has nothing to do with theoretical small signal dynamics. This resistor sets high frequency gain of the error amplifier, including the gain at the switching frequency. If the switching frequency gain is high enough, an excessive amout of output ripple voltage will appear at the VC pin resulting in improper operation of the regulator. In a marginal case, subharmonic switching occurs, as U W U U evidenced by alternating pulse widths seen at the switch node. In more severe cases, the regulator squeals or hisses audibly even though the output voltage is still roughly correct. None of this will show on a Bode plot since this is an amplitude insensitive measurement. Tests have shown that if ripple voltage on the VC is held to less than 100mVP-P, the LT1578 will generally be well behaved. The formula below will give an estimate of VC ripple voltage when RC is added to the loop, assuming that RC is large compared to the reactance of CC at 200kHz. VC RIPPLE = ( (RC)(GMA)(VIN − VOUT)(ESR)(1.21) ) (VIN)(L)(f) GMA = Error amplifier transconductance (1000µMho) If a series compensation resistor of 15k gave the best overall loop response, with adequate gain margin, the resulting VC pin ripple voltage with VIN = 10V, VOUT = 5V, ESR = 0.1Ω, L = 30µH, would be: (15k)(1• 10−3 )(10 − 5)(0.1)(1.21) = 0.151V VC(RIPPLE ) = (10)(30 • 10−6 )(200 • 103 ) This ripple voltage is high enough to possibly create subharmonic switching. In most situations a compromise value (< 10k in this case) for the resistor gives acceptable phase margin and no subharmonic problems. In other cases, the resistor may have to be larger to get acceptable phase response, and some means must be used to control ripple voltage at the VC pin. The suggested way to do this is to add a capacitor (CF) in parallel with the RC /CC network on the VC pin. The pole frequency for this capacitor is typically set at one-fifth of the switching frequency so that it provides significant attenuation of the switching ripple, but does not add unacceptable phase shift at the loop unity-gain frequency. With RC = 15k, CF = (2π)(f)(RC ) 5 = 2π 200 • 103 15k ( 5 )( ) = 265pF 21 LT1578/LT1578-2.5 APPLICATIONS INFORMATION How Do I Test Loop Stability? The “standard” compensation for LT1578 is a 100pF capacitor for CC, with RC = 0. While this compensation will work for most applications, the “optimum” value for loop compensation components depends, to various extents, on parameters which are not well controlled. These include inductor value (± 30% due to production tolerance, load current and ripple current variations), output capacitance (± 20% to ± 50% due to production tolerance, temperature, aging and changes at the load), output capacitor ESR (± 200% due to production tolerance, temperature and aging), and finally, DC input voltage and output load current . This makes it important for the designer to check out the final design to ensure that it is “robust” and tolerant of all these variations. One way to check switching regulator loop stability is by pulse loading the regulator output while observing the transient response at the output, using the circuit shown in Figure 13. The regulator loop is “hit” with a small transient AC load current at a relatively low frequency, 50Hz to 1kHz. This causes the output to jump a few millivolts, then settle back to the original value, as shown in Figure 14. A well behaved loop will settle back cleanly, whereas a loop with poor phase or gain margin will “ring” as it settles. The number of rings indicates the degree of stability, and the frequency of the ringing shows the approximate unity-gain frequency of the loop. Amplitude of the signal is not particularly important, as long as the amplitude is not so high that the loop behaves nonlinearly. SWITCHING REGULATOR ADJUSTABLE INPUT SUPPLY ADJUSTABLE DC LOAD Figure 13. Loop Stability Test Circuit 10mV/DIV 5A/DIV 0.2ms/DIV Figure 14. Loop Stability Check 22 U W U U RIPPLE FILTER 470Ω 4.7k 330pF TO X1 OSCILLOSCOPE PROBE + 100µF TO 1000µF 50Ω 3300pF TO OSCILLOSCOPE SYNC 100Hz TO 1kHz 100mV TO 1VP-P 1578 F13 VOUT AT IOUT = 500mA BEFORE FILTER VOUT AT IOUT = 500mA AFTER FILTER VOUT AT IOUT = 50mA AFTER FILTER LOAD PULSE THROUGH 50Ω f ≈ 780Hz 1578 F14 LT1578/LT1578-2.5 APPLICATIONS INFORMATION The output of the regulator contains both the desired low frequency transient information and a reasonable amount of high frequency (200kHz) ripple. The ripple makes it difficult to observe the small transient, so a two-pole, 100kHz filter has been added. This filter is not particularly critical; even if it attenuated the transient signal slightly, this wouldn’t matter because amplitude is not critical. After verifying that the setup is working correctly, start varying load current and input voltage to see if you can find any combination that makes the transient response look suspiciously “ringy.” This procedure may lead to an adjustment for best loop stability or faster loop transient response. Nearly always you will find that loop response looks better if you add in several kΩ for RC. Do this only if necessary, because as explained before, RC above 1k may require the addition of CF to control VC pin ripple. If everything looks OK, use a heat gun and cold spray on the circuit (especially the output capacitor) to bring out any temperature-dependent characteristics. Keep in mind that this procedure does not take initial component tolerance into account. You should see fairly clean response under all load and line conditions to ensure that component variations will not cause problems. One note here: according to Murphy, the component most likely to be changed in production is the output capacitor, because that is the component most likely to have manufacturer variations (in ESR) large enough to cause problems. It would be a wise move to lock down the sources of the output capacitor in production. Also, try varying component values by a factor of 2 and see if the behavior is still acceptable. Double and halve the values of RC and CC and output capacitors. If the regulator still works correctly, it will likely be good in production. A possible exception to the “clean response” rule is at very light loads, as evidenced in Figure 14 with ILOAD = 50mA. Switching regulators tend to have dramatic shifts in loop response at very light loads, mostly because the inductor current becomes discontinuous. One common result is very slow but stable characteristics. A second possibility is low phase margin, as evidenced by ringing at the output with transients. The good news is that the low phase margin at light loads is not particularly sensitive to component variation, so if it looks reasonable under a transient test, it will probably not be a problem in production. Note that frequency of the light load ringing may vary with component tolerance but phase margin generally hangs in there. POSITIVE-TO-NEGATIVE CONVERTER The circuit in Figure 15 is a classic positive-to-negative topology using a grounded inductor. It differs from the standard approach in the way the IC chip derives its feedback signal. Because the LT1578 accepts only positive feedback signals, the ground pin must be tied to the regulated negative output. A resistor divider to ground or, in this case, the sense pin, then provides the proper feedback voltage for the chip. D1 1N4148 C2 L1* 0.33µF 15µH VSW R1 15.8k FB GND VC CC RC R2 4.99k D2 1N5818 U W U U INPUT 5.5V TO 15V C3 10µF TO 50µF BOOST VIN LT1578 + + C1 100µF 10V TANT ×2 OUTPUT** – 5V, 0.5A * INCREASE L1 TO 30µH OR 60µH FOR HIGHER CURRENT APPLICATIONS. SEE APPLICATIONS INFORMATION ** MAXIMUM LOAD CURRENT DEPENDS ON MINIMUM INPUT VOLTAGE AND INDUCTOR SIZE. SEE APPLICATIONS INFORMATION 1578 F15 Figure 15. Positive-to-Negative Converter Inverting regulators differ from buck regulators in the basic switching network. Current is delivered to the output as square waves with a peak-to-peak amplitude much greater than load current. This means that maximum load current will be significantly less than the LT1578’s 1.5A maximum switch current, even with large inductor values. The buck converter in comparison, delivers current to the output as a triangular wave superimposed on a DC level equal to load current, and load current can approach 1.5A 23 LT1578/LT1578-2.5 APPLICATIONS INFORMATION with large inductors. Output ripple voltage for the positiveto-negative converter will be much higher than a buck converter. Ripple current in the output capacitor will also be much higher. The following equations can be used to calculate operating conditions for the positive-to-negative converter. Maximum load current:   VIN VOUT  VOUT VIN − 0.35 IP −  2 VOUT + VIN f L    IMAX = VOUT + VIN − 0.35 VOUT + VF This duty cycle is close enough to 50% that IP can be assumed to be 1.5A. OUTPUT DIVIDER If the adjustable part is used, the resistor connected to VOUT (R2) should be set to approximately 5k. R1 is calculated from: ( ( ( )( )( )( )( ) )( )( IP = Maximum rated switch current VIN = Minimum input voltage VOUT = Output voltage VF = Catch diode forward voltage 0.35 = Switch voltage drop at 1.5A Example: with VIN(MIN) = 5.5V, VOUT = 5V, L = 30µH, VF = 0.5V, IP = 1.5A: IMAX = 0.6A. Note that this equation does not take into account that maximum rated switch current (IP) on the LT1578 is reduced slightly for duty cycles above 50%. If duty cycle is expected to exceed 50% (input voltage less than output voltage), use the actual IP value from the Electrical Characteristics table. Operating duty cycle: OUTPUT RIPPLE VOLTAGE (mVP-P) DC = VOUT + VF VIN − 0.3 + VOUT + VF (This formula uses an average value for switch loss, so it may be several percent in error.) With the conditions above: DC = 5 + 0.5 = 51% 5.5 − 0.3 + 5 + 0.5 24 U W U U ) R1 = R2 VOUT − 1.21 1.21 ( ) ) INDUCTOR VALUE Unlike buck converters, positive-to-negative converters cannot use large inductor values to reduce output ripple voltage. At 200kHz, values larger than 75µH make almost no change in output ripple. The graph in Figure 16 shows peak-to-peak output ripple voltage for a 5V to – 5V converter versus inductor value. The criteria for choosing the 150 5V TO –5V CONVERTER OUTPUT CAPACITOR’S ESR = 0.1Ω DISCONTINUOUS ILOAD = 0.1A DISCONTINUOUS ILOAD = 0.25A 120 90 60 30 CONTINUOUS ILOAD > 0.38A 0 0 15 45 60 30 INDUCTOR SIZE (µH) 75 1578 F16 Figure 16. Ripple Voltage on Positive-to-Negative Converter LT1578/LT1578-2.5 APPLICATIONS INFORMATION inductor is therefore typically based on ensuring that peak switch current rating is not exceeded. This gives the lowest value of inductance that can be used, but in some cases (lower output load currents) it may give a value that creates unnecessarily high output ripple voltage. A compromise value is often chosen that reduces output ripple. As you can see from the graph, large inductors will not give arbitrarily low ripple, but small inductors can give high ripple. The difficulty in calculating the minimum inductor size needed is that you must first know whether the switcher will be in continuous or discontinuous mode at the critical point where switch current is 1.5A. The first step is to use the following formula to calculate the load current where the switcher must use continuous mode. If your load current is less than this, use the discontinuous mode formula to calculate the minimum inductor value needed. If the load current is higher, use the continuous mode formula. Output current where continuous mode is needed: ICONT = For the example above, with maximum load current of 0.25A: ICONT = (V ) (I ) 4(V + V )(V + V 2 2 IN P IN OUT IN OUT + VF ) Minimum inductor discontinuous mode: L MIN = 2 VOUT IOUT 2 P ( )( ) (f)(I ) (V )(V ) IN OUT Minimum inductor continuous mode: L MIN =   VOUT + VF 2 f VIN + VOUT IP − IOUT  1 +   VIN   ( )( ) ( U W U U (5.5) (1.5) = 0.38A 4(5.5 + 5)(5.5 + 5 + 0.5) 2 2 This says that discontinuous mode can be used and the minimum inductor needed is found from: L MIN = ( )( ) = 5.6µH  200 • 103  1.5 2  ( ) 2 5 0.25 In practice, the inductor should be increased by about 30% over the calculated minimum to handle losses and variations in value. This suggests a minimum inductor of 7.3µH for this application, but looking at the ripple voltage chart shows that output ripple voltage could be reduced by a factor of two by using a 30µH inductor. There is no rule of thumb here to make a final decision. If modest ripple is needed and the larger inductor does the trick, this is probably the best solution. If ripple is noncritical use the smaller inductor. If ripple is extremely critical, a second stage filter may have to be added in any case, and the lower value of inductance can be used. Keep in mind that the output capacitor is the other critical factor in determining output ripple voltage. Ripple shown on the graph (Figure 16) is with a capacitor’s ESR of 0.1Ω. This is reasonable for AVX type TPS “D” or “E” size surface mount solid tantalum capacitors, but the final capacitor chosen must be looked at carefully for ESR characteristics. )   25 LT1578/LT1578-2.5 APPLICATIONS INFORMATION Ripple Current in the Input and Output Capacitors Positive-to-negative converters have high ripple current in both the input and output capacitors. For long capacitor lifetime, the RMS value of this current must be less than the high frequency ripple current rating of the capacitor. The following formula will give an approximate value for RMS ripple current. This formula assumes continuous conduction mode and a large inductor value. Small inductors will give somewhat higher ripple current, especially in discontinuous mode. The exact formulas are very complex and appear in Application Note 44, pages 30 and 31. For our purposes here, a simple fudge factor (ff) is added. The value for ff is about 1.2 for load currents above 0.38A (in continuous conduction mode) and L ≥10µH. It increases to about 2.0 for smaller inductors at lower load currents (in discontinuous conduction mode). Capacitor IRMS = ff IOUT ff = Fudge factor (1.2 to 2.0) Diode Current ( )( ) VOUT VIN 26 U W U U Average diode current is equal to load current. Peak diode current will be considerably higher. Peak diode current: Continuous Mode = IOUT (VIN + VOUT ) + (VIN)(VOUT ) VIN 2(L)( f)( VIN + VOUT ) 2 IOUT VOUT Discontinuous Mode = ( )( ) (L)(f) Keep in mind that during start-up and output overloads, the average diode current may be much higher than with normal loads. Care should be used if diodes rated less than 1A are used, especially if continuous overload conditions must be tolerated. LT1578/LT1578-2.5 PACKAGE DESCRIPTION 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0°– 8° TYP 0.014 – 0.019 (0.355 – 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.016 – 0.050 (0.406 – 1.270) Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U Dimensions in inches (millimeters) unless otherwise noted. S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 8 7 6 5 0.228 – 0.244 (5.791 – 6.197) 0.150 – 0.157** (3.810 – 3.988) 1 2 3 4 0.053 – 0.069 (1.346 – 1.752) 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) BSC SO8 1298 27 LT1578/LT1578-2.5 TYPICAL APPLICATION Dual Output SEPIC Converter The circuit in Figure 17 generates both positive and negative 5V outputs with a single piece of magnetics. The inductor L1 is a 33µH surface mount inductor from Coiltronics. It is manufactured with two identical windings that can be connected in series or parallel. The topology for the 5V output is a standard buck converter. The – 5V topology would be a simple flyback winding coupled to the buck converter if C4 were not present. C4 creates the SEPIC (Single-Ended Primary Inductance Converter) topology which improves regulation and reduces ripple current in L1. Without C4, the voltage swing on L1B compared to L1A would vary due to relative loading and coupling losses. C4 provides a low impedance path to maintain an equal voltage swing in L1B, improving regulation. In a flyback converter, during switch on time, all the converter’s energy is stored in L1A only, since no current flows in L1B. At switch off, energy is transferred by magnetic coupling into L1B, powering the – 5V rail. C4 pulls L1B positive during switch on time, causing current to flow, and energy to build in L1B and C4. At switch off, the energy stored in both L1B and C4 supply the – 5V rail. This reduces the current in L1A and changes L1B current waveform from square to triangular. For details on this circuit see Design Note 100. INPUT 6V TO 15V GND * L1 IS A SINGLE CORE WITH TWO WINDINGS COILTRONICS CTX33-2 ** AVX TSPD107M010 † IF LOAD CAN GO TO ZERO, AN OPTIONAL PRELOAD OF 1k TO 5k MAY BE USED TO IMPROVE LOAD REGULATION C4** 100µF RELATED PARTS PART NUMBER LT1074/LT1076 LTC1174 LT1370 LT1371 LT1372/LT1377 LT1376 LT1507 LT1676/LT1776 LTC1772 LTC1735 LT1777 DESCRIPTION Step-Down Switching Regulators High Efficiency Step-Down and Inverting DC/DC Converter High Efficiency DC/DC Converter High Efficiency DC/DC Converter High Efficiency Step-Down Switching Regulator High Efficiency Step-Down Switching Regulator High Efficiency Step-Down Switching Regulators SOT-23 Low Voltage Step-Down DC/DC Controller High Efficiency Step-Down Converter Low Noise Step-Down Switching Regulator COMMENTS 40V Input, 100kHz, 5A and 2A 0.5A, 150kHz Burst ModeTM Operation 42V, 6A, 500kHz Switch 35V, 3A, 500kHz Switch 25V, 1.5A, 500kHz Switch 15V, 1.5A, 500kHz Switch 7.4V to 60V Input, 100kHz/200kHz 550kHz, Drives PFET, 6-Lead SOT-23 Package; up to 4.5A Output Current Synchronous Buck Controller Drives External MOSFETs 48V Input, Internally Limited dV/dt, Programmable di/dt 500kHz and 1MHz High Efficiency 1.5A Switching Regulators Boost Topology Burst Mode is a trademark of Linear Technology Corporation. 28 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 q FAX: (408) 434-0507 q www.linear-tech.com U + VIN BOOST LT1578 C2 0.33µF VSW D2 1N914 L1A* 33µH OUTPUT 5V R1 15.8k SHDN GND C3 22µF 35V TANT FB VC CC 100pF + D1 1N5818 R2 4.99k C1** 100µF 10V TANT + L1B* D3 1N5818 + C5** 100µF 10V TANT OUTPUT –5V† 1578 F17 Figure 17. Dual Output SEPIC Converter 1578f LT/TP 0100 4K • PRINTED IN USA © LINEAR TECHNOLOGY CORPORATION 1999