LT1767EMS8E-3.3

LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 Monolithic 1.5A, 1.25MHz Step-Down Switching Regulators FEATURES s s s s s s s s s s s s s DESCRIPTIO 1.5A Switch in a Small MSOP Package Constant 1.25MHz Switching Frequency High Power Exposed Pad (MS8E) Package Wide Operating Voltage Range: 3V to 25V High Efficiency 0.22Ω Switch 1.2V Feedback Reference Voltage Fixed Output Voltages of 1.8V, 2.5V, 3.3V, 5V 2% Overall Output Tolerance Uses Low Profile Surface Mount Components Low Shutdown Current: 6µA Synchronizable to 2MHz Current Mode Loop Control Constant Maximum Switch Current Rating at All Duty Cycles* The LT®1767 is a 1.25MHz monolithic buck switching regulator. A high efficiency 1.5A, 0.22Ω switch is included on the die together with all the control circuitry required to complete a high frequency, current mode switching regulator. Current mode control provides fast transient response and excellent loop stability. New design techniques achieve high efficiency at high switching frequencies over a wide operating range. A low dropout internal regulator maintains consistent performance over a wide range of inputs from 24V systems to LiIon batteries. An operating supply current of 1mA improves efficiency, especially at lower output currents. Shutdown reduces quiescent current to 6µA. Maximum switch current remains constant at all duty cycles. Synchronization allows an external logic level signal to increase the internal oscillator from 1.4MHz to 2MHz. The LT1767 is available in an 8-pin MSOP fused leadframe package and a low thermal resistance exposed pad package. Full cycle-by-cycle short-circuit protection and thermal shutdown are provided. High frequency operation allows the reduction of input and output filtering components and permits the use of chip inductors. , LTC and LT are registered trademarks of Linear Technology Corporation. *Patent Pending APPLICATIO S s s s s s DSL Modems Portable Computers Wall Adapters Battery-Powered Systems Distributed Power TYPICAL APPLICATIO 12V to 3.3V Step-Down Converter D2 CMDSH-3 C2 0.1µF VIN 12V BOOST VIN OPEN OR HIGH = ON LT1767-3.3 SHDN SYNC FB GND VC CC 1.5nF RC 4.7k *MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING. 1767 TA01 L1 5µH EFFICIENCY (%) C3 2.2µF CERAMIC VSW OUTPUT 3.3V 1.2A* D1 UPS120 C1 10µF CERAMIC U Efficiency vs Load Current 95 VIN = 10V VOUT = 5V 90 85 VIN = 5V VOUT = 3.3V 80 75 70 0 0.2 0.4 0.6 0.8 1 LOAD CURRENT (A) 1.2 1.4 1767 TA01a U U sn1767 1767fas 1 LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 ABSOLUTE MAXIMUM RATINGS Input Voltage .......................................................... 25V BOOST Pin Above SW ............................................ 20V Max BOOST Pin Voltage .......................................... 35V SHDN Pin ............................................................... 25V FB Pin Voltage .......................................................... 6V FB Pin Current ....................................................... 1mA PACKAGE/ORDER INFORMATION ORDER PART NUMBER TOP VIEW BOOST VIN SW GND 1 2 3 4 8 7 6 5 SYNC VC FB SHDN MS8 PACKAGE 8-LEAD PLASTIC MSOP TJMAX = 125°C, θJA = 110°C/W GROUND PIN CONNECTED TO LARGE COPPER AREA LT1767EMS8 LT1767EMS8-1.8 LT1767EMS8-2.5 LT1767EMS8-3.3 LT1767EMS8-5 MS8 PART MARKING LTLS LTWG LTWD LTWE LTWF Consult LTC Marketing for parts specified with wider operating temperature ranges. The q denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VC = 0.8V, Boost = VIN + 5V, SHDN, SYNC and switch open unless otherwise noted. PARAMETER Maximum Switch Current Limit Oscillator Frequency Switch On Voltage Drop VIN Undervoltage Lockout VIN Supply Current Shutdown Supply Current Feedback Voltage CONDITION TA = 0°C to 125°C TA = < 0°C 3.3V < VIN < 25V q ELECTRICAL CHARACTERISTICS ISW = – 1.5A, 0°C ≤ TA ≤ 125°C and –1.3A, TA < 0°C q (Note 3) VFB = VNOM + 17% VSHDN = 0V, VIN = 25V, VSW = 0V 3V < VIN < 25V, 0.4V < VC < 0.9V (Note 3) FB Input Current LT1767 (Adj) 2 U U W WW U W (Note 1) SYNC Pin Current .................................................. 1mA Operating Junction Temperature Range (Note 2) LT1767E .......................................... – 40°C to 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C ORDER PART NUMBER TOP VIEW BOOST VIN SW GND 1 2 3 4 8 7 6 5 SYNC VC FB SHDN MS8E PACKAGE 8-LEAD PLASTIC MSOP TJMAX = 125°C, θJA = 40°C/W EXPOSED GND PAD CONNECTED TO LARGE COPPER AREA ON PCB LT1767EMS8E LT1767EMS8E-1.8 LT1767EMS8E-2.5 LT1767EMS8E-3.3 LT1767EMS8E-5 MS8E PART MARKING LTZG LTZH LTZJ LTZK LTZL MIN 1.5 1.3 1.1 1.1 TYP 2 1.25 330 MAX 3 3 1.4 1.5 400 500 2.73 1.3 20 45 1.218 1.224 1.836 2.55 3.366 5.1 – 0.5 UNITS A A MHz MHz mV mV V mA µA µA V V V V V V µA q q q 2.47 2.6 1 6 LT1767 (Adj) q 1.182 1.176 1.764 2.45 3.234 4.9 1.2 1.8 2.5 3.3 5 – 0.25 LT1767-1.8 LT1767-2.5 LT1767-3.3 LT1767-5 q q q q q sn1767 1767fas LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 The q denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VC = 0.8V, Boost = VIN + 5V, SHDN, SYNC and switch open unless otherwise noted. PARAMETER FB Input Resistance CONDITION LT1767-1.8 LT1767-2.5 LT1767-3.3 LT1767-5 0.4V < VC < 0.9V ∆IVC = ± 10µA VFB = VNOM – 17% VFB = VNOM + 17% Duty Cycle = 0% VC = 1.2V, ISW = 400mA q q q q q q q q ELECTRICAL CHARACTERISTICS MIN 10.5 14.7 19 29 150 500 80 70 TYP 15 21 27.5 42 350 850 120 110 2.5 0.35 0.9 MAX 21 30 39 60 1300 160 180 UNITS kΩ kΩ kΩ kΩ µMho µA µA A/V V V % % Error Amp Voltage Gain Error Amp Transconductance VC Pin Source Current VC Pin Sink Current VC Pin to Switch Current Transconductance VC Pin Minimum Switching Threshold VC Pin 1.5A ISW Threshold Maximum Switch Duty Cycle Minimum Boost Voltage Above Switch Boost Current SHDN Threshold Voltage SHDN Input Current (Shutting Down) SHDN Threshold Current Hysteresis SYNC Threshold Voltage SYNC Input Frequency SYNC Pin Resistance 85 80 90 1.8 10 30 2.7 15 45 1.40 –13 10 2.2 2 20 ISW = – 1.5A, 0°C ≤ TA ≤ 125°C and –1.3A, TA < 0°C ISW = – 0.5A (Note 4) ISW = – 1.5A, 0°C ≤ TA ≤ 125°C and –1.3A, TA < 0°C (Note 4) SHDN = 60mV Above Threshold SHDN = 100mV Below Threshold q q q q q q V mA mA V µA µA V MHz kΩ 1.27 –7 4 1.5 1.33 –10 7 1.5 ISYNC = 1mA Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LT1767E is guaranteed to meet performance specifications from 0°C to 125°C. Specifications over the – 40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: Minimum input voltage is defined as the voltage where the internal regulator enters lockout. Actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. See Applications Information. Note 4: Current flows into the BOOST pin only during the on period of the switch cycle. TYPICAL PERFORMANCE CHARACTERISTICS FB vs Temperature (Adj) 1.22 400 125°C 350 1.21 300 250 –40°C 200 150 100 50 1.18 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 1767 G01 SWITCH VOLTAGE (mV) FREQUENCY (MHz) FB VOLTAGE (V) 1.20 1.19 UW Switch On Voltage Drop 1.50 1.45 1.40 1.35 1.30 1.25 1.20 1.15 Oscillator Frequency 25°C 0 0 0.5 1 SWITCH CURRENT (A) 1.5 1767 G02 1.10 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 1767 G03 sn1767 1767fas 3 LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 TYPICAL PERFOR A CE CHARACTERISTICS SHDN Threshold vs Temperature 1.40 1.38 SHDN THRESHOLD (V) VIN CURRENT (µA) SHDN INPUT (µA) 1.36 1.34 1.32 1.30 –50 –25 0 25 50 75 TEMPERATURE (°C) Minimum Input Voltage for 2.5V Out 3.5 300 3.3 INPUT VOLTAGE (V) VIN CURRENT (µA) VIN CURRENT (µA) 3.1 2.9 2.7 2.5 0.001 0.1 0.01 LOAD CURRENT (A) Current Limit Foldback 2.0 40 1.5 SWITCH PEAK CURRENT (A) OUTPUT CURRENT (A) SWITCH CURRENT 1.0 20 1.1 OUTPUT CURRENT (A) 1.5 0.5 FB CURRENT 0 0 0.2 0.4 0.6 0.8 FEEDBACK VOLTAGE (V) 4 UW 100 1767 G04 SHDN Supply Current vs VIN 7 SHDN = 0V 6 5 4 3 2 1 0 0 5 10 15 VIN (V) 20 25 30 1767 G05 SHDN IP Current vs Temperature –12 –10 SHUTTING DOWN –8 –6 –4 –2 0 –50 STARTING UP 125 –25 0 25 50 75 TEMPERATURE (°C) 100 125 1767 G06 SHDN Supply Current 1200 VIN = 15V 250 200 150 100 50 0 1000 800 600 400 200 0 Input Supply Current MINIMUM INPUT VOLTAGE 1 1767 G07 0 0.2 0.4 0.6 0.8 1 1.2 SHUTDOWN VOLTAGE (V) 1.4 0 5 10 15 20 INPUT VOLTAGE (V) 25 30 1767 G09 1767 G08 Maximum Load Current, VOUT = 5V 1.5 Maximum Load Current, VOUT = 2.5V 1.3 30 L = 4.7µH L = 4.7µH 1.3 L = 2.2µH 1.1 L = 1.5µH 0.9 FB INPUT CURRENT (µA) 0.9 L = 2.2µH 10 0.7 L = 1.5µH 1 0 1.2 1767 G10 0.5 0 5 10 15 INPUT VOLTAGE (V) 20 25 1767 G11 0.7 0 5 10 15 INPUT VOLTAGE (V) 20 25 1767 G12 sn1767 1767fas LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 PIN FUNCTIONS FB: The feedback pin is used to set output voltage using an external voltage divider that generates 1.2V at the pin with the desired output voltage. The fixed voltage 1.8V, 2.5V, 3.3V and 5V versions have the divider network included internally and the FB pin is connected directly to the output. If required, the current limit can be reduced during start up or short-circuit when the FB pin is below 0.5V (see the Current Limit Foldback graph in the Typical Performance Characteristics section). An impedance of less than 5kΩ (adjustable part only) at the FB pin is needed for this feature to operate. BOOST: The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. Without this added voltage, the typical switch voltage loss would be about 1.5V. The additional boost voltage allows the switch to saturate and voltage loss approximates that of a 0.22Ω FET structure. VIN: This is the collector of the on-chip power NPN switch. This pin powers the internal circuitry and internal regulator. At NPN switch on and off, high dI/dt edges occur on this pin. Keep the external bypass capacitor and catch diode close to this pin. All trace inductance in this path will create a voltage spike at switch off, adding to the VCE voltage across the internal NPN. GND: The GND pin acts as the reference for the regulated output, so load regulation will suffer if the “ground” end of the load is not at the same voltage as the GND pin of the IC. This condition will occur when load current or other currents flow through metal paths between the GND pin and the load ground point. Keep the ground path short between the GND pin and the load and use a ground plane when possible. Keep the path between the input bypass and the GND pin short. The GND pin of the MS8 package is directly attached to the internal tab. This pin should be attached to a large copper area to improve thermal resistance. The exposed pad of the MS8E package is also connected to GND. This should be soldered to a large copper area to improve its thermal resistance. VSW: The switch pin is the emitter of the on-chip power NPN switch. This pin is driven up to the input pin voltage during switch on time. Inductor current drives the switch pin negative during switch off time. Negative voltage must be clamped with an external catch diode with a VBR < 0.8V. SYNC: The sync pin is used to synchronize the internal oscillator to an external signal. It is directly logic compatible and can be driven with any signal between 20% and 80% duty cycle. The synchronizing range is equal to initial operating frequency, up to 2MHz. See Synchronization section in Applications Information for details. When not in use, this pin should be grounded. SHDN: The shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes. The 1.33V threshold can function as an accurate undervoltage lockout (UVLO), preventing the regulator from operating until the input voltage has reached a predetermined level. Float or pull high to put the regulator in the operating mode. VC: The VC pin is the output of the error amplifier and the input of the peak switch current comparator. It is normally used for frequency compensation, but can do double duty as a current clamp or control loop override. This pin sits at about 0.35V for very light loads and 0.9V at maximum load. It can be driven to ground to shut off the output. U U U sn1767 1767fas 5 LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 BLOCK DIAGRAM The LT1767 is a constant frequency, current mode buck converter. This means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. In addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. A switch cycle starts with an oscillator pulse which sets the RS flip-flop to turn the switch on. When switch current reaches a level set by the inverting input of the comparator, the flip-flop is reset and the switch turns off. Output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. This technique means that the error amplifier commands current to be delivered to the output rather than voltage. A voltage fed system will have low phase shift up to the resonant frequency of the inductor 0.01Ω VIN 2 2.5V BIAS REGULATOR SYNC 8 SHUTDOWN COMPARATOR 1.33V 3µA ERROR AMPLIFIER gm = 850µMho 7 VC Figure 1. Block Diagram sn1767 1767fas 6 – SHDN 5 + W and output capacitor, then an abrupt 180° shift will occur. The current fed system will have 90° phase shift at a much lower frequency, but will not have the additional 90° shift until well beyond the LC resonant frequency. This makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response. High switch efficiency is attained by using the BOOST pin to provide a voltage to the switch driver which is higher than the input voltage, allowing switch to be saturated. This boosted voltage is generated with an external capacitor and diode. A comparator connected to the shutdown pin disables the internal regulator, reducing supply current. + INTERNAL VCC – CURRENT SENSE AMPLIFIER VOLTAGE GAIN = 40 SLOPE COMP Σ 0.35V 1 BOOST 1.25MHz OSCILLATOR S CURRENT COMPARATOR RS FLIP-FLOP DRIVER CIRCUITRY + – 7µA R Q1 POWER SWITCH 3 VSW – + PARASITIC DIODES DO NOT FORWARD BIAS 6 FB 1.2V 4 GND 1767 F01 LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 APPLICATIONS INFORMATION FB RESISTOR NETWORK If an output voltage of 1.8V, 2.5V, 3.3V or 5V is required, the respective fixed option part, -1.8, -2.5, -3.3 or -5, should be used. The FB pin is tied directly to the output; the necessary resistive divider is already included on the part. For other voltage outputs, the adjustable part should be used and an external resistor divider added. The suggested resistor (R2) from FB to ground is 10k. This reduces the contribution of FB input bias current to output voltage to less than 0.25%. The formula for the resistor (R1) from VOUT to FB is: of capacitance is less important and has no significant effect on loop stability. If operation is required close to the minimum input required by the output of the LT1767, a larger value may be required. This is to prevent excessive ripple causing dips below the minimum operating voltage, resulting in erratic operation. If tantalum capacitors are used, values in the 22µF to 470µF range are generally needed to minimize ESR and meet ripple current and surge ratings. Care should be taken to ensure the ripple and surge ratings are not exceeded. The AVX TPS and Kemet T495 series are surge rated. AVX recommends derating capacitor operating voltage by 2:1 for high surge applications. OUTPUT CAPACITOR OUTPUT R1 = R2 VOUT − 1. 2 1.2 − R2(0.25µA) VSW ERROR AMPLIFIER ( ) LT1767 + – 1.2V FB R1 + R2 10k 1767 F02 VC GND Figure 2. Feedback Network INPUT CAPACITOR Step-down regulators draw current from the input supply in pulses. The rise and fall times of these pulses are very fast. The input capacitor is required to reduce the voltage ripple this causes at the input of LT1767 and force the switching current into a tight local loop, thereby minimizing EMI. The RMS ripple current can be calculated from: IRIPPLE(RMS) = IOUT VOUT VIN − VOUT ( ) / VIN2 Higher value, lower cost ceramic capacitors are now available in smaller case sizes. These are ideal for input bypassing since their high frequency capacitive nature removes most ripple current rating and turn-on surge problems. At higher switching frequency, the energy storage requirement of the input capacitor is reduced so values in the range of 1µF to 4.7µF are suitable for most applications. Y5V or similar type ceramics can be used since the absolute value U W U U Unlike the input capacitor, RMS ripple current in the output capacitor is normally low enough that ripple current rating is not an issue. The current waveform is triangular, with an RMS value given by: IRIPPLE(RMS) = 0.29 VOUT VIN − VOUT ( )( (L)(f)(VIN) ) The LT1767 will operate with both ceramic and tantalum output capacitors. Ceramic capacitors are generally chosen for their small size, very low ESR (effective series resistance), and good high frequency operation, reducing output ripple voltage. Their low ESR removes a useful zero in the loop frequency response, common to tantalum capacitors. To compensate for this, the VC loop compensation pole frequency must typically be reduced by a factor of 10. Typical ceramic output capacitors are in the 1µF to 10µF range. Since the absolute value of capacitance defines the pole frequency of the output stage, an X7R or X5R type ceramic, which have good temperature stability, is recommended. Tantalum capacitors are usually chosen for their bulk capacitance properties, useful in high transient load applications. ESR rather than capacitive value defines output ripple at 1.25MHz. Typical LT1767 applications require a tantalum capacitor with less than 0.3Ω ESR at 22µF to 500µF, see Table 2. sn1767 1767fas 7 LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 APPLICATIONS INFORMATION Table 2. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current E Case Size ESR (Max, Ω ) Ripple Current (A) AVX TPS, Sprague 593D AVX TAJ D Case Size AVX TPS, Sprague 593D C Case Size AVX TPS 0.2 (typ) 0.5 (typ) 0.1 to 0.3 0.7 to 1.1 0.1 to 0.3 0.7 to 0.9 0.7 to 1.1 0.4 Figure 3 shows a comparison of output ripple for a ceramic and tantalum capacitor at 200mA ripple current. VOUT USING 47µF, 0.1Ω TANTALUM CAPACITOR (10mV/DIV) VOUT USING 2.2µF CERAMIC CAPACITOR (10mV/DIV) VSW (5V/DIV) 0.2µs/DIV Figure 3. Output Ripple Voltage Waveform INDUCTOR CHOICE AND MAXIMUM OUTPUT CURRENT Maximum output current for a buck converter is equal to the maximum switch rating (IP) minus one half peak to peak inductor current. In past designs, the maximum switch current has been reduced by the introduction of slope compensation. Slope compensation is required at duty cycles above 50% to prevent an affect called subharmonic oscillation (see Application Note 19 for details). The LT1767 has a new circuit technique that maintains a constant switch current rating at all duty cycles. (Patent Pending) For most applications, the output inductor will be in the 1µH to 10µH range. Lower values are chosen to reduce the physical size of the inductor, higher values allow higher output currents due to reduced peak to peak ripple current, 8 U W U U and reduces the current at which discontinuous operation occurs. The following formula gives maximum output current for continuous mode operation, implying that the peak to peak ripple (2x the term on the right) is less than the maximum switch current. Continuous Mode IOUT (MAX) = IP − (VOUT )(VIN − VOUT ) 2(L)( f)(VIN ) Discontinuous operation occurs when IOUT (DIS) = (VOUT ) 2(L)(f) For VIN = 8V, VOUT = 5V and L = 3.3µH, IOUT (MAX) = 1.5 − 2 3.3 • 10− 6 1.25 • 106 8 = 1.5 − 0.23 = 1.27 A Note that the worst case (minimum output current available) condition is at the maximum input voltage. For the same circuit at 15V, maximum output current would be only 1.1A. 1767 F03 ( (5)(8 − 5) )( )( ) When choosing an inductor, consider maximum load current, core and copper losses, allowable component height, output voltage ripple, EMI, fault current in the inductor, saturation, and of course, cost. The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. Choose a value in microhenries from the graphs of maximum load current. Choosing a small inductor with lighter loads may result in discontinuous mode of operation, but the LT1767 is designed to work well in either mode. Assume that the average inductor current is equal to load current and decide whether or not the inductor must withstand continuous fault conditions. If maximum load current is 0.5A, for instance, a 0.5A inductor may not survive a continuous 2A overload condition. Also, the instantaneous application of input or release from shutdown, at high input voltages, may cause sn1767 1767fas LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 APPLICATIONS INFORMATION saturation of the inductor. In these applications, the soft-start circuit shown in Figure 10 should be used. 2. Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t omit this step. Powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core materials fall somewhere in between. IPEAK = IOUT + VOUT VIN − VOUT 2 L f VIN 4. After making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. Use the experts in the Linear Technology’s applications department if you feel uncertain about the final choice. They have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. CATCH DIODE The suggested catch diode (D1) is a UPS120 Schottky, or its Motorola equivalent, MBRM120LTI/MBRM130LTI. It is rated at 2A average forward current and 20V/30V reverse voltage. Typical forward voltage is 0.5V at 1A. The diode conducts current only during switch off time. Peak reverse voltage is equal to regulator input voltage. Average forward current in normal operation can be calculated from: ID ( AVG) = IOUT VIN − VOUT VIN ( )( )( ) ( ) VIN = Maximum input voltage f = Switching frequency, 1.25MHz 3. Decide if the design can tolerate an “open” core geometry like a rod or barrel, which have high magnetic field radiation, or whether it needs a closed core like a toroid to prevent EMI problems. This is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. Table 3 PART NUMBER Coiltronics TP1-2R2 TP2-2R2 TP3-4R7 TP4- 100 Murata LQH1C1R0M04 LQH3C1R0M24 LQH3C2R2M24 LQH4C1R5M04 Sumida CD73- 100 CDRH4D18-2R2 CDRH5D18-6R2 CDRH5D28-100 10 2.2 6.2 10 1.44 1.32 1.4 1.3 0.080 0.058 0.071 0.048 3.5 1.8 1.8 2.8 1.0 1.0 2.2 1.5 0.51 1.0 0.79 1.0 0.28 0.06 0.1 0.09 1.8 2.0 2.0 2.6 2.2 2.2 4.7 10 1.3 1.5 1.5 1.5 0.188 0.111 0.181 0.146 1.8 2.2 2.2 3.0 VALUE (uH) ISAT(Amps) DCR (Ω) HEIGHT (mm) U W U U ( ) BOOST PIN For most applications, the boost components are a 0.1µF capacitor and a CMDSH-3 diode. The anode is typically connected to the regulated output voltage to generate a voltage approximately VOUT above VIN to drive the output stage. The output driver requires at least 2.7V of headroom throughout the on period to keep the switch fully saturated. However, the output stage discharges the boost capacitor during the on time. If the output voltage is less than 3.3V, it is recommended that an alternate boost supply is used. The boost diode can be connected to the input, although, care must be taken to prevent the 2x VIN boost voltage from exceeding the BOOST pin absolute maximum rating. The additional voltage across the switch driver also increases power loss, reducing efficiency. If available, an independent supply can be used with a local bypass capacitor. A 0.1µF boost capacitor is recommended for most applications. Almost any type of film or ceramic capacitor is sn1767 1767fas 9 LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 APPLICATIONS INFORMATION suitable, but the ESR should be 10mA Figure 9. Dual Source Supply with 6µA Reverse Leakage C2 0.1µF D2 CMDSH-3 C2 0.1µF VIN 12V BOOST VIN LT1767-5 SHDN SYNC FB GND VC CC 330pF Q1 D1: UPS120 Q1: 2N3904 VSW D1 L1 5µH + C3 2.2µF 100µF + C1 CSS R3 15nF 2k 1767 F10 R4 47k Figure 10. Buck Converter with Adjustable Soft-Start PACKAGE DESCRIPTION MS8 Package 8-Lead Plastic MSOP (Reference LTC DWG # 05-08-1660) 0.043 (1.10) MAX 0.007 (0.18) 0.021 ± 0.006 (0.53 ± 0.015) 0° – 6° TYP SEATING PLANE 0.193 ± 0.006 (4.90 ± 0.15) 0.118 ± 0.004** (3.00 ± 0.102) 0.034 (0.86) REF 0.118 ± 0.004* (3.00 ± 0.102) 0.009 – 0.015 (0.22 – 0.38) * DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U U W U U BOOST VSW 5µH 3.3V, 1A ALTERNATE SUPPLY UPS120 2.2µF D2 CMDSH-3 VIN 6V TO 15V BOOST VIN LT1767-5 SHDN SYNC GND C3 2.2µF 16V CERAMIC VSW FB VC CC 330pF L1A* 9µH OUTPUT 5V OUTPUT 5V 1A + D1 C1 100µF 10V TANT GND C4 2.2µF * L1 IS A SINGLE CORE WITH TWO WINDINGS 16V BH ELECTRONICS #511-1013 † IF LOAD CAN GO TO ZERO, CERAMIC AN OPTIONAL PRELOAD OF 1k TO 5k MAY BE USED TO IMPROVE LOAD REGULATION D1, D3: UPS120 C5 L1B* 100µF 10V TANT + OUTPUT –5V† D3 1767 F11 Figure 11. Dual Output SEPIC Converter 8 76 5 0.0256 (0.65) BSC 0.005 ± 0.002 (0.13 ± 0.05) 1 23 4 MSOP (MS8) 1100 sn1767 1767fas 15 LT1767/LT1767-1.8/ LT1767-2.5/LT1767-3.3/LT1767-5 PACKAGE DESCRIPTION MS8E Package 8-Lead Plastic MSOP (Reference LTC DWG # 05-08-1662) 0.889 ± 0.127 (.035 ± .005) 3.00 ± 0.102 (.118 ± .004) (NOTE 3) BOTTOM VIEW OF EXPOSED PAD OPTION 8 7 65 0.52 (.206) REF 1 2.06 ± 0.102 (.080 ± .004) 1.83 ± 0.102 (.072 ± .004) 2.794 ± 0.102 (.110 ± .004) 5.23 (.206) MIN 2.083 ± 0.102 3.2 – 3.45 (.082 ± .004) (.126 – .136) 0.42 ± 0.04 (.0165 ± .0015) TYP 0.65 (.0256) BSC GAUGE PLANE 0.53 ± 0.015 (.021 ± .006) DETAIL “A” 0.18 (.077) 1 1.10 (.043) MAX 23 4 0.86 (.034) REF 8 RECOMMENDED SOLDER PAD LAYOUT NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX RELATED PARTS PART NUMBER LT1370 LT1371 LT1372/LT1377 LT1374 LT1375/LT1376 LT1507 LT1576 LT1578 LT1616 LT1676/LT1776 LTC1765 LTC1877 LTC1878 LTC3401 LTC3402 LTC3404 DESCRIPTION High Efficiency DC/DC Converter High Efficiency DC/DC Converter 500kHz and 1MHz High Efficiency 1.5A Switching Regulators High Efficiency Step-Down Switching Regulator 1.5A Step-Down Switching Regulators 1.5A Step-Down Switching Regulator 1.5A Step-Down Switching Regulator 1.5A Step-Down Switching Regulator 600mA Step-Down Switching Regulator Wide Input Range Step-Down Switching Regulators 1.25MHz, 3A Wide Input Range Step-Down DC/DC High Efficiency Monolithic Step-Down Regulator High Efficiency Monolithic Step-Down Regulator Single Cell, High Current (1A), Micropower, Synchronous 3MHz Step-Up DC/DC Converter Single Cell, High Current (2A), Micropower, Synchronous 3MHz Step-Up DC/DC Converter 1.4MHz High Efficiency, Monolithic Synchronous Step-Down Regulator COMMENTS 42V, 6A, 500kHz Switch 35V, 3A, 500kHz Switch Boost Topology 25V, 4.5A, 500kHz Switch 500kHz, Synchronizable in SO-8 Package 500kHz, 4V to 16V Input, SO-8 Package 200kHz, Reduced EMI Generation 200kHz, Reduced EMI Generation 1.4MHz, 4V to 25V Input, SOT-23 Package 60V Input, 700mA Internal Switches VTH = 3V to 25V, SO-8 and TSSOP-16E Packages 550kHz, MS8, VIN Up to 10V, IQ =10µA, IOUT to 600mA at VIN = 5V 550kHz, MS8, VIN Up to 6V, IQ = 10µA, IOUT to 600mA at VIN = 3.3V VIN = 0.5V to 5V, Up to 97% Efficiency Synchronizable Oscillator from 100kHz to 3MHz VIN = 0.7V to 5V, Up to 95% Efficiency Synchronizable Oscillator from 100kHz to 3MHz Up to 95% Efficiency, 100% Duty Cycle, IQ = 10µA, VIN = 2.65V to 6V Burst Mode is a trademark of Linear Technology Corporation. sn1767 1767fas 16 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 q FAX: (408) 434-0507 q U 0.254 (.010) DETAIL “A” 0° – 6° TYP 4.88 ± 0.1 (.192 ± .004) 3.00 ± 0.102 (.118 ± .004) NOTE 4 SEATING PLANE 0.22 – 0.38 (.009 – .015) 0.65 (.0256) BCS 0.13 ± 0.05 (.005 ± .002) MSOP (MS8E) 0102 LT/TP 0302 REV A 2K • PRINTED IN USA www.linear.com © LINEAR TECHNOLOGY CORPORATION 1999