LT1912EMSE-TRPBF

LT1912 36V, 2A, 500kHz Step-Down Switching Regulator FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTION The LT®1912 is an adjustable frequency (200kHz to 500kHz) monolithic step-down switching regulator that accepts input voltages up to 36V. A high efficiency 0.25Ω switch is included on the die along with a boost Schottky diode and the necessary oscillator, control, and logic circuitry. Current mode topology is used for fast transient response and good loop stability. The LT1912 allows the use of ceramic capacitors resulting in low output ripple while keeping total solution size to a minimum. The low current shutdown mode reduces input supply current to less than 1μA while a resistor and capacitor on the RUN/SS pin provide a controlled output voltage ramp (soft-start). The LT1912 is available in 10-Pin MSOP and 3mm × 3mm DFN packages with exposed pads for low thermal resistance. , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Wide Input Range: Operation From 3.6V to 36V 2A Maximum Output Current Adjustable Switching Frequency: 200kHz to 500kHz Low Shutdown Current: IQ < 1μA Integrated Boost Diode Synchronizable Between 250kHz to 500kHz Saturating Switch Design: 0.25Ω On-Resistance 0.790V Feedback Reference Voltage Output Voltage: 0.79V to 20V Soft-Start Capability Small 10-Pin Thermally Enhanced MSOP and (3mm × 3mm) DFN Packages APPLICATIONS ■ ■ ■ ■ ■ Automotive Battery Regulation Set Top Box Distributed Supply Regulation Industrial Supplies Wall Transformer Regulation TYPICAL APPLICATION 3.3V Step-Down Converter VIN 4.5V TO 36V VIN OFF ON 20k 4.7μF 470pF 68.1k VC RT 316k GND FB 100k 47μF 50 0 1912 TA01 Efficiency VOUT 3.3V 2A 100 VOUT = 5V 90 EFFICIENCY (%) VOUT = 3.3V BD BOOST 0.47μF LT1912 SW 6.8μH RUN/SS 80 70 60 VIN = 12V L = 6.8μF F = 500kHz 0.5 1.0 1.5 LOAD CURRENT (A) 2 1912 TA01b SYNC 1912f 1 LT1912 ABSOLUTE MAXIMUM RATINGS (Note 1) VIN, RUN/SS Voltage .................................................36V BOOST Pin Voltage ...................................................56V BOOST Pin Above SW Pin.........................................30V FB, RT, VC Voltage .......................................................5V BD, SYNC Voltage .....................................................30V Operating Junction Temperature Range (Note 2) LT1912E ........................................... –40°C to 125°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec) (MSE Only) ........................................................... 300°C PIN CONFIGURATION TOP VIEW BD BOOST SW VIN RUN/SS 1 2 3 4 5 11 10 RT 9 VC 8 FB 7 N/C 6 SYNC TOP VIEW BD BOOST SW VIN RUN/SS 1 2 3 4 5 10 9 8 7 6 RT VC FB N/C SYNC 11 DD PACKAGE 10-LEAD (3mm × 3mm) PLASTIC DFN JA = 45°C/W, JC = 10°C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB MSE PACKAGE 10-LEAD PLASTIC MSOP JA = 45°C/W, JC = 10°C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH LT1912EDD#PBF LT1912EMSE#PBF TAPE AND REEL LT1912EDD#TRPBF LT1912EMSE#TRPBF PART MARKING LDJT LTDJS PACKAGE DESCRIPTION 10-Lead (3mm × 3mm) Plastic DFN 10-Lead Plastic MSOP TEMPERATURE RANGE –40°C to 125°C –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS PARAMETER Minimum Input Voltage Quiescent Current from VIN The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUN/SS = 10V, VBOOST = 15V, VBD = 3.3V unless otherwise noted. (Note 2) CONDITIONS ● MIN TYP 3 0.01 450 1.3 0.01 0.9 1 2.7 MAX 3.6 0.5 600 1.7 0.5 1.3 5 3 UNITS V μA μA μA μA mA μA V VRUN/SS = 0.2V VBD = 3V, Not Switching VBD = 0, Not Switching VRUN/SS = 0.2V VBD = 3V, Not Switching VBD = 0, Not Switching ● Quiescent Current from BD ● Minimum Bias Voltage (BD Pin) 1912f 2 LT1912 ELECTRICAL CHARACTERISTICS PARAMETER Feedback Voltage ● The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUN/SS = 10V VBOOST = 15V, VBD = 3.3V unless otherwise noted. (Note 2) CONDITIONS MIN 780 775 TYP 790 790 7 0.002 25 1000 45 45 3.5 2 RT = 187k ● MAX 800 805 30 0.01 UNITS mV mV nA %/V μMho μA μA A/V V FB Pin Bias Current (Note 3) FB Voltage Line Regulation Error Amp gm Error Amp Gain VC Source Current VC Sink Current VC Pin to Switch Current Gain VC Clamp Voltage Switching Frequency Minimum Switch Off-Time Switch Current Limit Switch VCESAT Boost Schottky Reverse Leakage Minimum Boost Voltage (Note 4) BOOST Pin Current RUN/SS Pin Current RUN/SS Input Voltage High RUN/SS Input Voltage Low SYNC Low Threshold SYNC High Threshold SYNC Pin Bias Current VFB = 0.8V, VC = 0.4V 4V < VIN < 36V ● 160 3.2 200 60 3.7 500 0.02 240 150 4.2 2 2.1 35 10 2.5 kHz nS A mV μA V mA μA V V V Duty Cycle = 5% ISW = 2A VSW = 10V, VBD = 0V ● 1.5 22 5 0.2 0.5 ISW = 1A VRUN/SS = 2.5V 0.7 VSYNC = 0V 0.1 V μA Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT1912E is guaranteed to meet performance specifications from 0°C to 125°C. Specifications over the –40°C to 125°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: Bias current flows out of the FB pin. Note 4: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the switch. 1912f 3 LT1912 TYPICAL PERFOR A CE CHARACTERISTICS Efficiency 100 VIN = 12V 90 85 80 EFFICIENCY (%) VIN = 34V VIN = 24V 70 VIN = 34V 75 70 65 60 L: NEC PLC-0745-5R6 f: 500kHz 55 50 0 VOUT = 3.3V 1.5 L: NEC PLC-0745-5R6 f: 500kHz 1.0 5 10 VOUT = 3.3V L = 4.7μH f = 500kHz 15 20 INPUT VOLTAGE (V) 25 30 1912 G03 90 EFFICIENCY (%) LOAD CURRENT (A) 80 60 VOUT = 5V 0 50 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 LOAD CURRENT (A) 1912 G01 Maximum Load Current 3.5 TYPICAL 3.0 LOAD CURRENT (A) SWITCH CURRENT LIMIT(A) 4.0 3.5 3.0 2.5 2.0 1.5 1.0 SWITCH CURRENT LIMIT (A) 2.5 MINIMUM 2.0 1.5 1.0 5 10 20 15 INPUT VOLTAGE (V) Switch Voltage Drop 700 600 BOOST PIN CURRENT (mA) VOLTAGE DROP (mV) 500 400 300 200 100 0 0 500 1000 2000 1500 SWITCH CURRENT (mA) 2500 1912 G07 4 UW VOUT = 5V L = 4.7μH f = 500kHz 25 TA = 25°C unless otherwise noted. Efficiency VIN = 7V VIN = 12V 4.0 3.5 3.0 2.5 Maximum Load Current TYPICAL VIN = 24V MINIMUM 2.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 LOAD CURRENT (A) 1912 G02 Switch Current Limit 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 20 60 40 DUTY CYCLE (%) 80 100 1912 G05 Switch Current Limit DUTY CYCLE = 10 % DUTY CYCLE = 90 % 30 1912 G04 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1912 G06 Boost Pin Current 80 70 60 50 40 30 20 10 0 0 500 1000 1500 2000 SWITCH CURRENT (mA) 2500 1912 G08 1912f LT1912 TYPICAL PERFOR A CE CHARACTERISTICS Feedback Voltage 840 1.20 1.15 820 FREQUENCY (NORMALIZED) FEEDBACK VOLTAGE (mV) 1.10 1.05 1.00 0.95 0.90 0.85 760 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1912 G09 SWITCHING FREQUENCY (NORMALIZED) 800 780 Minimum Switch On-Time 140 MINIMUM SWITCH ON TIME (ns) 120 SWITCH CURRENT LIMIT (A) 100 80 60 40 20 0 –50 –25 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 0 25 50 75 100 125 150 TEMPERATURE (˚C) 1912 G12 RUN/SS PIN CURRENT (μA) Boost Diode 1.4 1.2 BOOST DIODE Vf (V) VC PIN CURRENT (μA) 1.0 0.8 0.6 0.4 0.2 0 50 40 30 20 10 0 –10 –20 –30 –40 0 0.5 1.0 1.5 BOOST DIODE CURRENT (A) 2.0 1912 G15 UW TA = 25°C unless otherwise noted. Switching Frequency 1.2 1.0 0.8 0.6 0.4 0.2 0 0 25 50 75 100 125 150 TEMPERATURE (°C) 1912 G10 Frequency Foldback 0.80 –50 –25 0 100 200 300 400 500 600 700 800 900 FB PIN VOLTAGE (mV) 1912 G11 Soft-Start 12 10 8 6 4 2 0 RUN/SS Pin Current 0 0.5 2.5 2 1.5 RUN/SS PIN VOLTAGE (V) 1 3 3.5 1912 G13 0 5 20 30 15 25 10 RUN/SS PIN VOLTAGE (V) 35 1912 G14 Error Amp Output Current –50 –200 0 –100 100 FB PIN ERROR VOLTAGE (V) 200 1912 G16 1912f 5 LT1912 TYPICAL PERFOR A CE CHARACTERISTICS VC Voltages 2.50 VSW 5V/DIV 2.00 CURRENT LIMIT CLAMP VC VOLTAGE (V) 1.50 1.00 SWITCHING THRESHOLD VOUT 10mV/DIV VOUT 10mV/DIV 0.50 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1912 G19 6 UW TA = 25°C unless otherwise noted. Switching Waveforms; Continuous Operation Switching Waveforms; Discontinuous Operation VSW 5V/DIV IL 0.5A/DIV IL 1A/DIV 2μs/DIV VIN = 12V; VOUT = 3.3V ILOAD = 110mA 1912 G21 2μs/DIV VIN = 12V; VOUT = 3.3V ILOAD = 1A 1912 G22 1912f LT1912 PI FU CTIO S BD (Pin 1): This pin connects to the anode of the boost Schottky diode. BD also supplies current to the internal regulator. BOOST (Pin 2): This pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. SW (Pin 3): The SW pin is the output of the internal power switch. Connect this pin to the inductor, catch diode and boost capacitor. VIN (Pin 4): The VIN pin supplies current to the LT1912’s internal regulator and to the internal power switch. This pin must be locally bypassed. RUN/SS (Pin 5): The RUN/SS pin is used to put the LT1912 in shutdown mode. Tie to ground to shut down the LT1912. Tie to 2.5V or more for normal operation. If the shutdown feature is not used, tie this pin to the VIN pin. RUN/SS also provides a soft-start function; see the Applications Information section. SYNC (Pin 6): This is the external clock synchronization input. Ground this pin when SYNC function is not used. Tie to a clock source for synchronization. Clock edges should have rise and fall times faster than 1μs. See synchronizing section in Applications Information. N/C (Pin 7): This pin should be tied to ground. FB (Pin 8): The LT1912 regulates the FB pin to 0.790V. Connect the feedback resistor divider tap to this pin. VC (Pin 9): The VC pin is the output of the internal error amplifier. The voltage on this pin controls the peak switch current. Tie an RC network from this pin to ground to compensate the control loop. RT (Pin 10): Oscillator Resistor Input. Connecting a resistor to ground from this pin sets the switching frequency. Exposed Pad (Pin 11): Ground. The Exposed Pad must be soldered to PCB. BLOCK DIAGRA VIN C1 4 VIN INTERNAL 0.79V REF 5 RUN/SS ∑ SLOPE COMP 10 RT 6 RT OSCILLATOR 200kHz–500kHz SYNC SOFT-START ERROR AMP + – GND 11 R2 FB 8 R1 + – W U U U BD 1 SWITCH LATCH R Q S BOOST 2 C3 SW L1 3 D1 C2 VOUT VC CLAMP VC 9 CC CF RC 1912 BD 1912f 7 LT1912 OPERATION The LT1912 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT, enables an RS flip-flop, turning on the internal power switch. An amplifier and comparator monitor the current flowing between the VIN and SW pins, turning the switch off when this current reaches a level determined by the voltage at VC. An error amplifier measures the output voltage through an external resistor divider tied to the FB pin and servos the VC pin. If the error amplifier’s output increases, more current is delivered to the output; if it decreases, less current is delivered. An active clamp on the VC pin provides current limit. The VC pin is also clamped to the voltage on the RUN/SS pin; soft-start is implemented by generating a voltage ramp at the RUN/SS pin using an external resistor and capacitor. An internal regulator provides power to the control circuitry. The bias regulator normally draws power from the VIN pin, but if the BD pin is connected to an external voltage higher than 3V bias power will be drawn from the external source (typically the regulated output voltage). This improves efficiency. The RUN/SS pin is used to place the LT1912 in shutdown, disconnecting the output and reducing the input current to less than 1μA. The switch driver operates from either the input or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to fully saturate the internal bipolar NPN power switch for efficient operation. The oscillator reduces the LT1912’s operating frequency when the voltage at the FB pin is low. This frequency foldback helps to control the output current during startup and overload. 1912f 8 LT1912 APPLICATIONS INFORMATION FB Resistor Network The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the 1% resistors according to: V R1= R2 OUT – 1 0.79 V Reference designators refer to the Block Diagram. Setting the Switching Frequency The LT1912 uses a constant frequency PWM architecture that can be programmed to switch from 200kHz to 500kHz by using a resistor tied from the RT pin to ground. A table showing the necessary RT value for a desired switching frequency is in Figure 1. SWITCHING FREQUENCY (kHz) 200 300 400 500 RT VALUE (kΩ) 187 121 88.7 68.1 minimum on and off times. The switch can turn on for a minimum of ~150ns and turn off for a minimum of ~150ns. Typical minimum on time at 25°C is 80ns. This means that the minimum and maximum duty cycles are: DCMIN = fSW tON(MIN ) DCMAX = 1– fSW tOFF (MIN ) where fSW is the switching frequency, the tON(MIN) is the minimum switch on time (~150ns), and the tOFF(MIN) is the minimum switch off time (~150ns). These equations show that duty cycle range increases when switching frequency is decreased. A good choice of switching frequency should allow adequate input voltage range (see next section) and keep the inductor and capacitor values small. Input Voltage Range The maximum input voltage for LT1912 applications depends on switching frequency, the Absolute Maximum Ratings of the VIN and BOOST pins, and the operating mode. While the output is in start-up, short-circuit, or other overload conditions, the switching frequency should be chosen according to the following equation. VIN(MIN ) = VOUT + VD –V +V 1– fSW tOFF (MIN ) D SW Figure 1. Switching Frequency vs. RT Value Operating Frequency Tradeoffs Selection of the operating frequency is a tradeoff between efficiency, component size, minimum dropout voltage, and maximum input voltage. The advantage of high frequency operation is that smaller inductor and capacitor values may be used. The disadvantages are lower efficiency, lower maximum input voltage, and higher dropout voltage. The highest acceptable switching frequency (fSW(MAX)) for a given application can be calculated as follows: fSW (MAX ) = VD + VOUT tON(MIN ) ( VD + VIN – VSW ) where VIN is the typical input voltage, VOUT is the output voltage, VD is the catch diode drop (~0.5V) and VSW is the internal switch drop (~0.5V at max load). This equation shows that slower switching frequency is necessary to safely accommodate high VIN/VOUT ratio. Also, as shown in the next section, lower frequency allows a lower dropout voltage. The reason input voltage range depends on the switching frequency is because the LT1912 switch has finite where VIN(MAX) is the maximum operating input voltage, VOUT is the output voltage, VD is the catch diode drop (~0.5V), VSW is the internal switch drop (~0.5V at max load), fSW is the switching frequency (set by RT), and tON(MIN) is the minimum switch on time (~150ns). Note that a higher switching frequency will depress the maximum operating input voltage. Conversely, a lower switching frequency will be necessary to achieve safe operation at high input voltages. If the output is in regulation and no short-circuit, startup, or overload events are expected, then input voltage transients of up to 36V are acceptable regardless of the switching frequency. In this mode, the LT1912 may enter pulse skipping operation where some switching pulses 1912f 9 LT1912 APPLICATIONS INFORMATION are skipped to maintain output regulation. In this mode the output voltage ripple and inductor current ripple will be higher than in normal operation. The minimum input voltage is determined by either the LT1912’s minimum operating voltage of ~3.6V or by its maximum duty cycle (see equation in previous section). The minimum input voltage due to duty cycle is: VIN(MAX ) = VOUT + VD –V +V fSW tON(MIN ) D SW The largest inductor ripple current occurs at the highest VIN. To guarantee that the ripple current stays below the specified maximum, the inductor value should be chosen according to the following equation: L= VOUT + VD fSW IL 1– VOUT + VD VIN(MAX ) where VIN(MIN) is the minimum input voltage, and tOFF(MIN) is the minimum switch off time (150ns). Note that higher switching frequency will increase the minimum input voltage. If a lower dropout voltage is desired, a lower switching frequency should be used. Inductor Selection For a given input and output voltage, the inductor value and switching frequency will determine the ripple current. The ripple current ΔIL increases with higher VIN or VOUT and decreases with higher inductance and faster switching frequency. A reasonable starting point for selecting the ripple current is: ΔIL = 0.4(IOUT(MAX)) where IOUT(MAX) is the maximum output load current. To guarantee sufficient output current, peak inductor current must be lower than the LT1912’s switch current limit (ILIM). The peak inductor current is: IL(PEAK) = IOUT(MAX) + ΔIL/2 where IL(PEAK) is the peak inductor current, IOUT(MAX) is the maximum output load current, and ΔIL is the inductor ripple current. The LT1912’s switch current limit (ILIM) is at least 3.5A at low duty cycles and decreases linearly to 2.5A at DC = 0.8. The maximum output current is a function of the inductor ripple current: IOUT(MAX) = ILIM – ΔIL/2 Be sure to pick an inductor ripple current that provides sufficient maximum output current (IOUT(MAX)). where VD is the voltage drop of the catch diode (~0.4V), VIN(MAX) is the maximum input voltage, VOUT is the output voltage, fSW is the switching frequency (set by RT), and L is in the inductor value. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be about 30% higher. For robust operation in fault conditions (start-up or short circuit) and high input voltage (>30V), the saturation current should be above 3.5A. To keep the efficiency high, the series resistance (DCR) should be less than 0.1Ω, and the core material should be intended for high frequency applications. Table 1 lists several vendors and suitable types. Table 1. Inductor Vendors VENDOR Murata TDK Toko URL www.murata.com www.componenttdk.com www.toko.com PART SERIES LQH55D SLF7045 SLF10145 D62CB D63CB D75C D75F Sumida www.sumida.com CR54 CDRH74 CDRH6D38 CR75 TYPE Open Shielded Shielded Shielded Shielded Shielded Open Open Shielded Shielded Open Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger value inductor provides a slightly higher maximum load current and will reduce the output voltage ripple. If your load is lower than 2A, then you can decrease the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one 1912f 10 LT1912 APPLICATIONS INFORMATION with a lower DCR resulting in higher efficiency. There are several graphs in the Typical Performance Characteristics section of this data sheet that show the maximum load current as a function of input voltage and inductor value for several popular output voltages. Low inductance may result in discontinuous mode operation, which is okay but further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), there is a minimum inductance required to avoid subharmonic oscillations. See AN19. Input Capacitor Bypass the input of the LT1912 circuit with a ceramic capacitor of X7R or X5R type. Y5V types have poor performance over temperature and applied voltage, and should not be used. A 4.7μF to 10μF ceramic capacitor is adequate to bypass the LT1912 and will easily handle the ripple current. Note that larger input capacitance is required when a lower switching frequency is used. If the input power source has high impedance, or there is significant inductance due to long wires or cables, additional bulk capacitance may be necessary. This can be provided with a lower performance electrolytic capacitor. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT1912 and to force this very high frequency switching current into a tight local loop, minimizing EMI. A 4.7μF capacitor is capable of this task, but only if it is placed close to the LT1912 and the catch diode (see the PCB Layout section). A second precaution regarding the ceramic input capacitor concerns the maximum input voltage rating of the LT1912. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT1912 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT1912’s voltage rating. This situation is easily avoided (see the Hot Plugging Safely section). Output Capacitor and Output Ripple The output capacitor has two essential functions. Along with the inductor, it filters the square wave generated by the LT1912 to produce the DC output. In this role it determines the output ripple, and low impedance at the switching frequency is important. The second function is to store energy in order to satisfy transient loads and stabilize the LT1912’s control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A good starting value is: COUT = 100 VOUT fSW where fSW is in MHz, and COUT is the recommended output capacitance in μF Use X5R or X7R types. This . choice will provide low output ripple and good transient response. Transient performance can be improved with a higher value capacitor if the compensation network is also adjusted to maintain the loop bandwidth. A lower value of output capacitor can be used to save space and cost but transient performance will suffer. See the Frequency Compensation section to choose an appropriate compensation network. When choosing a capacitor, look carefully through the data sheet to find out what the actual capacitance is under operating conditions (applied voltage and temperature). A physically larger capacitor, or one with a higher voltage rating, may be required. High performance tantalum or electrolytic capacitors can be used for the output capacitor. Low ESR is important, so choose one that is intended for use in switching regulators. The ESR should be specified by the supplier, and should be 0.05Ω or less. Such a capacitor will be larger than a ceramic capacitor and will have a larger capacitance, because the capacitor must be large to achieve low ESR. Table 2 lists several capacitor vendors. 1912f 11 LT1912 APPLICATIONS INFORMATION Table 2. Capacitor Vendors VENDOR Panasonic PHONE (714) 373-7366 URL www.panasonic.com PART SERIES Ceramic, Polymer, Tantalum Kemet Sanyo (864) 963-6300 (408) 749-9714 www.kemet.com www.sanyovideo.com Ceramic, Tantalum Ceramic, Polymer, Tantalum Murata AVX Taiyo Yuden (864) 963-6300 (408) 436-1300 www.murata.com www.avxcorp.com www.taiyo-yuden.com Ceramic Ceramic, Tantalum Ceramic TPS Series POSCAP T494, T495 EEF Series COMMANDS Catch Diode The catch diode conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID(AVG) = IOUT (VIN – VOUT)/VIN where IOUT is the output load current. The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to the typical peak switch current. Peak reverse voltage is equal to the regulator input voltage. Use a Schottky diode with a reverse voltage rating greater than the input voltage. Table 3 lists several Schottky diodes and their manufacturers. Table 3. Diode Vendors PART NUMBER On Semicnductor MBRM120E MBRM140 Diodes Inc. B220 B230 DFLS240L International Rectifier 10BQ030 20BQ030 VR (V) 20 40 20 30 40 30 30 IAVE (A) 1 1 2 2 2 1 2 420 VF AT 1A (mV) 530 550 VF AT 2A (mV) 595 Ceramic Capacitors A precaution regarding ceramic capacitors concerns the maximum input voltage rating of the LT1912. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT1912 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT1912’s rating. This situation is easily avoided (see the Hot Plugging Safely section). Frequency Compensation The LT1912 uses current mode control to regulate the output. This simplifies loop compensation. In particular, the LT1912 does not require the ESR of the output capacitor for stability, so you are free to use ceramic capacitors to achieve low output ripple and small circuit size. Frequency compensation is provided by the components tied to the VC pin, as shown in Figure 2. Generally a capacitor (CC) and a resistor (RC) in series to ground are used. In addition, there may be lower value capacitor in parallel. This capacitor (CF) is not part of the loop compensation but is used to filter noise at the switching frequency, and is required only if a phase-lead capacitor is used or if the output capacitor has high ESR. Loop compensation determines the stability and transient performance. Designing the compensation network is a bit complicated and the best values depend on the application and in particular the type of output capacitor. A 1912f 500 500 500 470 470 12 LT1912 APPLICATIONS INFORMATION practical approach is to start with one of the circuits in this data sheet that is similar to your application and tune the compensation network to optimize the performance. Stability should then be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the stability using a transient load. Figure 2 shows an equivalent circuit for the LT1912 control loop. The error amplifier is a transconductance amplifier with finite output impedance. The power section, consisting of the modulator, power switch and inductor, is modeled as a transconductance amplifier generating an output current proportional to the voltage at the VC pin. Note that the output capacitor integrates this current, and that the capacitor on the VC pin (CC) integrates the error amplifier output current, resulting in two poles in the loop. In most cases a zero is required and comes from either the output capacitor ESR or from a resistor RC in series with CC. This simple model works well as long as the value of the inductor is not too high and the loop crossover frequency is much lower than the switching frequency. A phase lead capacitor (CPL) across the feedback divider may improve the transient response. Figure 3 shows the transient response when the load current is stepped from 500mA to 1500mA and back to 500mA. BOOST and BIAS Pin Considerations Capacitor C3 and the internal boost Schottky diode (see the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases a 0.22μF capacitor will work well. Figure 2 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.3V above the SW pin for best efficiency. For outputs of 3V and above, the standard circuit (Figure 4a) is best. For outputs between 2.8V and 3V, use a 1μF boost capacitor. A 2.5V output presents a special case because it is marginally adequate to support the boosted drive stage while using the internal boost diode. For reliable BOOST pin operation with 2.5V outputs use a good external Schottky diode (such as the ON Semi MBR0540), and a 1μF boost capacitor (see Figure 4b). For lower output voltages the boost diode can be tied to the input (Figure 4c), or to another supply greater than 2.8V. Tying BD to VIN reduces the maximum input voltage to 30V. The circuit in Figure 4a is more efficient because the BOOST pin current and BD pin quiescent current comes from a lower voltage source. You must also be sure that the maximum voltage ratings of the BOOST and BD pins are not exceeded. The minimum operating voltage of an LT1912 application is limited by the minimum input voltage (3.6V) and by the maximum duty cycle as outlined in a previous section. For proper startup, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, LT1912 CURRENT MODE POWER STAGE gm = 3.5mho SW ERROR AMPLIFIER FB ESR gm = 420μmho 3Meg C1 POLYMER OR TANTALUM R2 CERAMIC R1 CPL OUTPUT VC CF RC CC Figure 2. Model for Loop Response VOUT 100mV/DIV IL 0.5A/DIV VIN = 12V; FRONT PAGE APPLICATION 10μs/DIV 1912 F03 Figure 3. Transient Load Response of the LT1912 Front Page Application as the Load Current is Stepped from 500mA to 1500mA. VOUT = 3.3V 1912f – + 0.8V + C1 GND 1912 F02 13 LT1912 APPLICATIONS INFORMATION then the boost capacitor may not be fully charged. Because the boost capacitor is charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. Figure 5 shows a plot of minimum load to start and to run as a function of input voltage. In many cases the discharged output capacitor will present a load to the switcher, which will allow it to start. The plots show the worst-case situation where VIN is ramping very slowly. For lower start-up voltage, the boost diode can be tied to VIN; however, this restricts the input range to one-half of the absolute maximum rating of the BOOST pin. At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This reduces the minimum input voltage to approximately 300mV above VOUT. At higher load currents, the inductor current is continuous and the duty cycle is limited by the maximum duty cycle of the LT1912, requiring a higher input voltage to maintain regulation. VIN VOUT BD BOOST VIN VIN LT1912 SW C3 4.7μF GND (4a) For VOUT > 2.8V VOUT BD BOOST VIN VIN LT1912 SW C3 D2 4.7μF GND (4b) For 2.5V < VOUT < 2.8V VOUT BD BOOST VIN LT1912 SW C3 Soft-Start The RUN/SS pin can be used to soft-start the LT1912, reducing the maximum input current during start-up. The RUN/SS pin is driven through an external RC filter to create a voltage ramp at this pin. Figure 6 shows the startup and shut-down waveforms with the soft-start circuit. By choosing a large RC time constant, the peak start-up current can be reduced to the current that is required to regulate the output, with no overshoot. Choose the value of the resistor so that it can supply 20μA when the RUN/SS pin reaches 2.5V. Synchronization Synchronizing the LT1912 oscillator to an external frequency can be done by connecting a square wave (with 20% to 80% duty cycle) to the SYNC pin. The square wave amplitude should have valleys that are below 0.3V and peaks that are above 0.8V (up to 6V). The LT1912 may be synchronized over a 250kHz to 500kHz range. The RT resistor should be chosen to set the LT1912 4.7μF GND 1912 FO4 (4c) For VOUT < 2.5V; VIN(MAX) = 30V Figure 4. Three Circuits For Generating The Boost Voltage switching frequency 20% below the lowest synchronization input. For example, if the synchronization signal will be 250kHz and higher, the RT should be chosen for 200kHz. To assure reliable and safe operation the LT1912 will only synchronize when the output voltage is near regulation. It is therefore necessary to choose a large enough inductor value to supply the required output current at the frequency set by the RT resistor. See Inductor Selection section. It is also important to note that slope compensation is set by the RT value: When the sync frequency is much higher than the one set by RT, the slope compensation will be significantly reduced which may require a larger inductor value to prevent subharmonic oscillation. 1912f 14 LT1912 APPLICATIONS INFORMATION 6.0 5.5 INPUT VOLTAGE (V) 5.0 4.5 4.0 TO RUN 3.5 3.0 2.5 2.0 1 VOUT = 3.3V TA = 25°C L = 8.2μH f = 500kHz 10 100 1000 LOAD CURRENT (A) 10000 2ms/DIV 1912 F06 TO START (WORST CASE) RUN 15k RUN/SS 0.22μF GND IL 1A/DIV VRUN/SS 2V/DIV VOUT 2V/DIV 8.0 7.0 INPUT VOLTAGE (V) 6.0 VIN Figure 6. To Soft-Start the LT1912, Add a Resisitor and Capacitor to the RUN/SS Pin D4 MBRS140 VIN RUN/SS VC BOOST LT1912 TO START (WORST CASE) 5.0 4.0 3.0 2.0 1 TO RUN SW VOUT VOUT = 5V TA = 25°C L = 8.2μH f = 500kHz 10 100 1000 LOAD CURRENT (A) 10000 GND FB BACKUP 1912 F07 1912 F05 Figure 5. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit Shorted and Reversed Input Protection If the inductor is chosen so that it won’t saturate excessively, an LT1912 buck regulator will tolerate a shorted output. There is another situation to consider in systems where the output will be held high when the input to the LT1912 is absent. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode OR-ed with the LT1912’s output. If the VIN pin is allowed to float and the RUN/SS pin is held high (either by a logic signal or because it is tied to VIN), then the LT1912’s internal circuitry will pull its quiescent current through its SW pin. This is fine if your system can tolerate a few mA in this state. If you ground the RUN/SS pin, the SW pin current will drop to essentially zero. However, if the VIN pin is grounded while the output is held high, then parasitic diodes inside the LT1912 can pull large currents from the output through Figure 7. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output. It Also Protects the Circuit from a Reversed Input. The LT1912 Runs Only When the Input is Present the SW pin and the VIN pin. Figure 7 shows a circuit that will run only when the input voltage is present and that protects against a shorted or reversed input. PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 8 shows the recommended component placement with trace, ground plane and via locations. Note that large, switched currents flow in the LT1912’s VIN and SW pins, the catch diode (D1) and the input capacitor (C1). The loop formed by these components should be as small as possible. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components. 1912f 15 LT1912 APPLICATIONS INFORMATION The SW and BOOST nodes should be as small as possible. Finally, keep the FB and VC nodes small so that the ground traces will shield them from the SW and BOOST nodes. The Exposed Pad on the bottom of the package must be soldered to ground so that the pad acts as a heat sink. To keep thermal resistance low, extend the ground plane as much as possible, and add thermal vias under and near the LT1912 to additional ground planes within the circuit board and on the bottom side. Hot Plugging Safely The small size, robustness and low impedance of ceramic capacitors make them an attractive option for the input bypass capacitor of LT1912 circuits. However, these capacitors can cause problems if the LT1912 is plugged into a live supply (see Linear Technology Application Note 88 for a complete discussion). The low loss ceramic capacitor, combined with stray inductance in series with the power source, forms an under damped tank circuit, and the voltage at the VIN pin of the LT1912 can ring to twice the nominal input voltage, possibly exceeding the LT1912’s rating and damaging the part. If the input supply is poorly controlled or the user will be plugging the LT1912 into an energized supply, the input network should be designed to prevent this overshoot. Figure 9 shows the waveforms that result when an LT1912 circuit is connected to a 24V supply through six feet of 24-gauge twisted pair. The first plot is the response with a 4.7μF ceramic capacitor at the input. The input voltage rings as high as 50V and the input current peaks at 26A. A good solution is shown in Figure 9b. A 0.7Ω resistor is added in series with the input to eliminate the voltage overshoot (it also reduces the peak input current). A 0.1μF capacitor improves high frequency filtering. For high input voltages its impact on efficiency is minor, reducing efficiency by 1.5 percent for a 5V output at full load operating from 24V. R1 L1 VOUT C2 RRT CC RC R2 D1 C1 GND 1912 F08 VIAS TO LOCAL GROUND PLANE VIAS TO VOUT VIAS TO SYNC VIAS TO RUN/SS VIAS TO VIN OUTLINE OF LOCAL GROUND PLANE Figure 8. A Good PCB Layout Ensures Proper, Low EMI Operation High Temperature Considerations The PCB must provide heat sinking to keep the LT1912 cool. The Exposed Pad on the bottom of the package must be soldered to a ground plane. This ground should be tied to large copper layers below with thermal vias; these layers will spread the heat dissipated by the LT1912. Place additional vias can reduce thermal resistance further. With these steps, the thermal resistance from die (or junction) to ambient can be reduced to JA = 35°C/W or less. With 100 LFPM airflow, this resistance can fall by another 25%. Further increases in airflow will lead to lower thermal resistance. Because of the large output current capability of the LT1912, it is possible to dissipate enough heat to raise the junction temperature beyond the absolute maximum of 125°C. When operating at high ambient temperatures, the maximum load current should be derated as the ambient temperature approaches 125°C. 1912f 16 LT1912 APPLICATIONS INFORMATION Power dissipation within the LT1912 can be estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss and inductor loss. The die temperature is calculated by multiplying the LT1912 power dissipation by the thermal resistance from junction to ambient. Other Linear Technology Publications Application Notes 19, 35 and 44 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design Note 100 shows how to generate a bipolar output supply using a buck regulator. 1912f 17 LT1912 APPLICATIONS INFORMATION CLOSING SWITCH SIMULATES HOT PLUG IIN VIN LT1912 DANGER VIN 20V/DIV RINGING VIN MAY EXCEED ABSOLUTE MAXIMUM RATING + 4.7μF LOW IMPEDANCE ENERGIZED 24V SUPPLY STRAY INDUCTANCE DUE TO 6 FEET (2 METERS) OF TWISTED PAIR IIN 10A/DIV 20μs/DIV (9a) 0.7Ω LT1912 VIN 20V/DIV + 0.1μF 4.7μF IIN 10A/DIV (9b) 20μs/DIV LT1912 VIN 20V/DIV + 22μF 35V AI.EI. + 4.7μF IIN 10A/DIV (9c) 20μs/DIV 1912 F09 Figure 9. A Well Chosen Input Network Prevents Input Voltage Overshoot and Ensures Reliable Operation when the LT1912 is Connected to a Live Supply 1912f 18 LT1912 TYPICAL APPLICATIONS 5V Step-Down Converter VIN 6.8V TO 36V VIN ON OFF RUN/SS BD BOOST 0.47μF 4.7μF 16.2k 68.1k 470pF f = 500kHz 1912 TA02 VOUT 5V 2A L 6.8μH VC RT LT1912 SW D SYNC 536k GND FB 100k 47μF D: DIODES INC. DFLS240L L: TAIYO YUDEN NP06DZB6R8M 3.3V Step-Down Converter VIN 4.4V TO 36V VIN ON OFF RUN/SS BD BOOST 0.47μF 4.7μF 20k 68.1k 470pF f = 500kHz 1912 TA03 VOUT 3.3V 2A L 6.8μH VC RT LT1912 SW D SYNC 316k GND FB 100k 47μF D: DIODES INC. DFLS240L L: TAIYO YUDEN NP06DZB4R7M 1912f 19 LT1912 TYPICAL APPLICATIONS 2.5V Step-Down Converter VIN 4V TO 36V VIN ON OFF RUN/SS BD BOOST 1μF 4.7μF 20k 215k 68.1k 330pF f = 500kHz 1912 TA04 D2 L 6.8μH VOUT 2.5V 2A VC RT LT1912 SW D1 SYNC GND FB 100k 47μF D1: DIODES INC. DFLS240L D2: MBR0540 L: TAIYO YUDEN NP06DZB4R7M 1912f 20 LT1912 TYPICAL APPLICATIONS 12V Step-Down Converter VIN 15V TO 36V VIN ON OFF RUN/SS BD BOOST 0.47μF 10μF 26.1k 68.1kHz 330pF f = 500kHz 1912 TA06 VOUT 12V 2A L 10μH VC RT LT1912 SW D SYNC GND 715k FB 50k 22μF D: DIODES INC. DFLS240L L: NEC/TOKIN PLC-0755-100 1.8V Step-Down Converter VIN 3.5V TO 27V VIN ON OFF RUN/SS BD BOOST 0.47μF 4.7μF 18.2k 68.1k 330pF f = 500kHz 1912 TA08 VOUT 1.8V 2A L 3.3μH VC RT LT1912 SW D SYNC 127k GND FB 100k 47μF D: DIODES INC. DFLS240L L: TAIYO YUDEN NP06DZB3R3M 1912f 21 LT1912 PACKAGE DESCRIPTION DD Package 10-Lead Plastic DFN (3mm × 3mm) (Reference LTC DWG # 05-08-1699) R = 0.115 TYP 6 0.675 ± 0.05 0.38 ± 0.10 10 3.50 ± 0.05 1.65 ± 0.05 2.15 ± 0.05 (2 SIDES) PACKAGE OUTLINE 0.25 ± 0.05 0.50 BSC 2.38 ± 0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE PIN 1 TOP MARK (SEE NOTE 6) 3.00 ± 0.10 (4 SIDES) 1.65 ± 0.10 (2 SIDES) (DD) DFN 1103 5 0.200 REF 0.75 ± 0.05 2.38 ± 0.10 (2 SIDES) 1 0.25 ± 0.05 0.50 BSC 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD 1912f 22 LT1912 PACKAGE DESCRIPTION MSE Package 10-Lead Plastic MSOP (Reference LTC DWG # 05-08-1663) BOTTOM VIEW OF EXPOSED PAD OPTION 2.794 ± 0.102 (.110 ± .004) 0.889 ± 0.127 (.035 ± .005) 1 2.06 ± 0.102 (.081 ± .004) 1.83 ± 0.102 (.072 ± .004) 5.23 (.206) MIN 2.083 ± 0.102 3.20 – 3.45 (.082 ± .004) (.126 – .136) 10 0.50 0.305 ± 0.038 (.0197) (.0120 ± .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT 3.00 ± 0.102 (.118 ± .004) (NOTE 3) 10 9 8 7 6 0.497 ± 0.076 (.0196 ± .003) REF 4.90 ± 0.152 (.193 ± .006) 0.254 (.010) GAUGE PLANE 0.53 ± 0.152 (.021 ± .006) DETAIL “A” 0.18 (.007) SEATING PLANE 1.10 (.043) MAX DETAIL “A” 0° – 6° TYP 12345 3.00 ± 0.102 (.118 ± .004) (NOTE 4) 0.86 (.034) REF 0.17 – 0.27 (.007 – .011) TYP NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 0.50 (.0197) BSC 0.127 ± 0.076 (.005 ± .003) MSOP (MSE) 0603 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 1912f 23 LT1912 TYPICAL APPLICATIO VIN 3.6V TO 27V VIN ON OFF RUN/SS BD BOOST 0.47μF 4.7μF 16.2k 68.1k 330pF f = 500kHz 1912 TA09 D: DIODES INC. DFLS240L L: TAIYO YUDEN NP06DZB3R3M RELATED PARTS PART NUMBER LT1933 LT3437 LT1936 LT3493 LT1976/LT1977 LT1767 LT1940 LT1766 LT3434/LT3435 LT3480 LT3481 LT3684 LT3685 DESCRIPTION 500mA (IOUT), 500kHz Step-Down Switching Regulator in SOT-23 60V, 400mA (IOUT), MicroPower Step-Down DC/DC Converter with Burst Mode 36V, 1.4A (IOUT), 500kHz High Efficiency Step-Down DC/DC Converter 36V, 1.2A (IOUT), 750kHz High Efficiency Step-Down DC/DC Converter 60V, 1.2A (IOUT), 200kHz/500kHz, High Efficiency StepDown DC/DC Converter with Burst Mode 25V, 1.2A (IOUT), 1.1MHz, High Efficiency Step-Down DC/DC Converter Dual 25V, 1.4A (IOUT), 1.1MHz, High Efficiency Step-Down DC/DC Converter 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 60V, 2.4A (IOUT), 200/500kHz, High Efficiency Step-Down DC/DC Converter with Burst Mode COMMENTS VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.6mA, ISD