LT1934IS6-1

LT1934/LT1934-1 Micropower Step-Down Switching Regulators in ThinSOT FEATURES s s s s s s s s s DESCRIPTIO Wide Input Voltage Range: 3.2V to 34V Micropower Operation: IQ = 12µA 5V at 250mA from 6.5V to 34V Input (LT1934) 5V at 60mA from 6.5V to 34V Input (LT1934-1) 3.3V at 250mA from 4.5V to 34V Input (LT1934) 3.3V at 60mA from 4.5V to 34V Input (LT1934-1) Low Shutdown Current: 1V VFB = 0V VFB = 1V –40°C ≤ TA ≤ 85°C –40°C ≤ TA ≤ 125°C q q q q VFB Falling q q 1.22 1.21 1.25 1.25 10 2 2 0.007 1.4 85 83 1.8 12 88 88 200 65 ISW = 300mA (LT1934) ISW = 75mA (LT1934-1) LT1934 LT1934-1 ISW = 300mA (LT1934) ISW = 75mA (LT1934-1) ISW = 300mA (LT1934) ISW = 75mA (LT1934-1) 350 90 300 120 490 160 12 10 2.5 2.5 2 400 120 8.5 6.0 1.8 1.7 2 U mV mV mA mA mA mA V V µA 1934f W U U WW W LT1934/LT1934-1 The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VBOOST = 15V, unless otherwise noted. PARAMETER SHDN Pin Current SHDN Input Voltage High SHDN Input Voltage Low Note 1: Absolute Maximum Ratings are those values beyond which the life of the device may be impaired. Note 2: The LT1934E and LT1934E-1 are guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and CONDITIONS VSHDN = 2.3V VSHDN = 34V 2.3 0.25 MIN TYP 0.5 1.5 MAX 5 UNITS µA µA V V ELECTRICAL CHARACTERISTICS correlation with statistical process controls. The LT1934I and LT1934I-1 specifications are guaranteed over the –40°C to 125°C temperature range. Note 3: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. TYPICAL PERFOR A CE CHARACTERISTICS LT1934 Efficiency, VOUT = 5V 100 LT1934 VOUT = 5V L = 47µH 90 TA = 25°C 80 100 VIN = 12V EFFICIENCY (%) EFFICIENCY (%) EFFICIENCY (%) VIN = 24V 70 60 50 0.1 1 10 100 LOAD CURRENT (mA) 1934 G01 LT1934-1 Efficiency, VOUT = 3.3V 100 500 SWITCH CURRENT LIMIT (mA) 90 EFFICIENCY (%) LT1934-1 VOUT = 3.3V L = 100µH TA = 25°C VIN = 12V VIN = 24V 80 300 OFF TIME (µs) 70 60 50 0.1 1 10 100 1934 G04 LOAD CURRENT (mA) UW LT1934 Efficiency, VOUT = 3.3V LT1934 VOUT = 3.3V L = 47µH 90 TA = 25°C 80 VIN = 24V VIN = 12V 70 100 LT1934-1 Efficiency, VOUT = 5V LT1934-1 VOUT = 5V L = 150µH TA = 25°C VIN = 12V 80 VIN = 24V 70 90 VIN = 5V 60 60 50 0.1 50 1 10 100 LOAD CURRENT (mA) 1934 G02 0.1 1 10 100 1934 G03 LOAD CURRENT (mA) Current Limit vs Temperature 3.0 LT1934 400 2.5 2.0 1.5 1.0 0.5 Off Time vs Temperature 200 LT1934-1 100 0 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 0 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 125 1934 G05 1934 G06 1934f 3 LT1934/LT1934-1 TYPICAL PERFOR A CE CHARACTERISTICS Frequency Foldback 16 14 SHDN PIN CURRENT (µA) TA = 25°C 10 8 6 4 2 0 0 0.2 0.8 1.0 0.6 FEEDBACK PIN VOLTAGE (V) 0.4 1.2 1934 G07 FEEDBACK VOLTAGE (V) SWITCH OFF TIME (µs) 12 Quiescent Current vs Temperature 20 4.0 QUIESCENT CURRENT (µA) 15 UVLO (V) 10 5 0 –50 –25 Minimum Input Voltage VOUT = 3.3V 6.0 LT1934 VOUT = 3.3V 5.5 TA = 25°C BOOST DIODE TIED TO OUTPUT INPUT VOLTAGE (V) 5.0 4.5 4.0 INPUT VOLTAGE (V) VIN TO START VIN TO RUN 3.5 3.0 0.1 4 UW VFB vs Temperature 1.27 2.0 SHDN Bias Current vs SHDN Voltage TA = 25°C 1.26 1.5 1.25 1.0 1.24 0.5 1.23 1.22 –50 –25 0 50 25 0 75 TEMPERATURE (°C) 100 125 0 2 4 6 8 SHDN PIN VOLTAGE (V) 10 12 1934 G09 1934 G08 Undervoltage Lockout vs Temperature 3.5 3.0 2.5 75 0 25 50 TEMPERATURE (°C) 100 125 2.0 –50 –25 75 0 25 50 TEMPERATURE (°C) 100 125 1934 G10 1934 G11 Minimum Input Voltage VOUT = 5V 8 LT1934 VOUT = 5V TA = 25°C 7 BOOST DIODE TIED TO OUTPUT VIN TO START 6 VIN TO RUN 5 1 10 100 LOAD CURRENT (mA) 1934 G12 4 0.1 1 10 100 LOAD CURRENT (mA) 1934 G13 1934f LT1934/LT1934-1 PI FU CTIO S BOOST (Pin 1): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. GND (Pin 2): Tie the GND pin to a local ground plane below the LT1934 and the circuit components. Return the feedback divider to this pin. FB (Pin 3): The LT1934 regulates its feedback pin to 1.25V. Connect the feedback resistor divider tap to this pin. Set the output voltage according to VOUT = 1.25V (1 + R1/R2) or R1 = R2 (VOUT/1.25 – 1). SHDN (Pin 4): The SHDN pin is used to put the LT1934 in shutdown mode. Tie to ground to shut down the LT1934. Apply 2.3V or more for normal operation. If the shutdown feature is not used, tie this pin to the VIN pin. VIN (Pin 5): The VIN pin supplies current to the LT1934’s internal regulator and to the internal power switch. This pin must be locally bypassed. SW (Pin 6): The SW pin is the output of the internal power switch. Connect this pin to the inductor, catch diode and boost capacitor. BLOCK DIAGRA VIN VIN C2 5 + + – BOOST ON TIME 12µs DELAY OFF TIME 1.8µs DELAY R Q′ S Q SW L1 6 D1 C1 VOUT C3 D2 1 ON OFF 4 SHDN W U U U VREF 1.25V + ENABLE – 2 GND R2 3 FB R1 FEEDBACK COMPARATOR 1934 BD 1934f 5 LT1934/LT1934-1 OPERATIO The LT1934 uses Burst Mode control, combining both low quiescent current operation and high switching frequency, which result in high efficiency across a wide range of load currents and a small total circuit size. A comparator monitors the voltage at the FB pin of the LT1934. If this voltage is higher than the internal 1.25V reference, the comparator disables the oscillator and power switch. In this state, only the comparator, reference and undervoltage lockout circuits are active, and the current into the VIN pin is just 12µA. As the load current discharges the output capacitor, the voltage at the FB pin falls below 1.25V and the comparator enables the oscillator. The LT1934 begins to switch, delivering current to the output capacitor. The output voltage rises, and when it overcomes the feedback comparator’s hysteresis, the oscillator is disabled and the LT1934 returns to its micropower state. The oscillator consists of two one-shots and a flip-flop. A rising edge from the off-time one-shot sets the flipflop, which turns on the internal NPN power switch. The switch remains on until either the on-time one-shot trips or the current limit is reached. A sense resistor and amplifier monitor the current through the switch and resets 6 U (Refer to Block Diagram) the flip-flop when this current reaches 400mA (120mA for the LT1934-1). After the 1.8µs delay of the off-time one-shot, the cycle repeats. Generally, the LT1934 will reach current limit on every cycle—the off time is fixed and the on time is regulated so that the LT1934 operates at the correct duty cycle. The 1.8µs off time is lengthened when the FB pin voltage falls below 0.8V; this foldback behavior helps control the output current during start-up and overload. Figure 1 shows several waveforms of an LT1934 producing 3.3V from a 10V input. When the switch is on, the SW pin voltage is at 10V. When the switch is off, the inductor current pulls the SW pin down until it is clamped near ground by the external catch diode. The switch driver operates from either the input or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to fully saturate the bipolar switch for efficient operation. If the SHDN pin is grounded, all internal circuits are turned off and VIN current reduces to the device leakage current, typically a few nA. VOUT 50mV/DIV VSW 10V/DIV ISW 0.5A/DIV ILI 0.5A/DIV 5µs/DIV 1934 F01a Figure 1. Operating Waveforms of the LT1934 Converting 10V to 3.3V at 180mA (Front Page Schematic) 1934f LT1934/LT1934-1 APPLICATIO S I FOR ATIO Which One to Use: LT1934 or LT1934-1? The only difference between the LT1934 and LT1934-1 is the peak current through the internal switch and the inductor. If your maximum load current is less than 60mA, use the LT1934-1. If your maximum load is higher, use the LT1934; it can supply up to ~300mA. While the LT1934-1 can’t deliver as much output current, it has other advantages. The lower peak switch current allows the use of smaller components (input capacitor, inductor and output capacitor). The ripple current at the input of the LT1934-1 circuit will be smaller and may be an important consideration if the input supply is current limited or has high impedance. The LT1934-1’s current draw during faults (output overload or short) and start-up is lower. The maximum load current that the LT1934 or LT1934-1 can deliver depends on the value of the inductor used. Table 1 lists inductor value, minimum output capacitor and maximum load for 3.3V and 5V circuits. Increasing the value of the capacitor will lower the output voltage ripple. Component selection is covered in more detail in the following sections. Minimum Input Voltage The minimum input voltage required to generate a particular output voltage is determined by either the LT1934’s undervoltage lockout of ~3V or by its maximum duty Table 1 PART LT1934 VOUT 3.3V L 100µH 47µH 33µH 150µH 68µH 47µH 150µH 100µH 68µH 220µH 150µH 100µH MINIMUM COUT 100µF 47µF 33µF 47µF 33µF 22µF 15µF 10µF 10µF 10µF 4.7µF 4.7µF MAXIMUM LOAD 300mA 250mA 200mA 300mA 250mA 200mA 60mA 45mA 20mA 60mA 45mA 20mA 5V LT1934-1 3.3V 5V U cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: DC = (VOUT + VD)/(VIN – VSW + VD) where VD is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.3V at maximum load for the LT1934, ~0.1V for the LT1934-1). This leads to a minimum input voltage of: VIN(MIN) = (VOUT + VD)/DCMAX – VD + VSW with DCMAX = 0.85. Inductor Selection A good first choice for the inductor value is: L = 2.5 • (VOUT + VD) • 1.8µs/ILIM where ILIM is the switch current limit (400mA for the LT1934 and 120mA for the LT1934-1). This choice provides a worst-case maximum load current of 250mA (60mA for the LT1934-1). The inductor’s RMS current rating must be greater than the load current and its saturation current should be greater than ILIM. To keep efficiency high, the series resistance (DCR) should be less than 0.3Ω (1Ω for the LT1934-1). Table 2 lists several vendors and types that are suitable. This simple rule may not provide the optimum value for your application. If the load current is less, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. The following provides more details to guide inductor selection. First, the value must be chosen so that the LT1934 can supply the maximum load current drawn from the output. Second, the inductor must be rated appropriately so that the LT1934 will function reliably and the inductor itself will not be overly stressed. Detailed Inductor Selection and Maximum Load Current The square wave that the LT1934 produces at its switch pin results in a triangle wave of current in the inductor. The LT1934 limits the peak inductor current to ILIM. Because 1934f W UU 7 LT1934/LT1934-1 APPLICATIO S I FOR ATIO Table 2. Inductor Vendors Vendor Murata Sumida Phone (404) 426-1300 (847) 956-0666 URL www.murata.com www.sumida.com Part Series LQH3C CR43 CDRH4D28 CDRH5D28 DO1607C DO1608C DT1608C WE-PD1, 2, 3, 4 Comments Small, Low Cost, 2mm Height Coilcraft (847) 639-6400 www.coilcraft.com Wurth Electronics (866) 362-6673 www.we-online.com the average inductor current equals the load current, the maximum load current is: IOUT(MAX) = IPK – ∆IL /2 where IPK is the peak inductor current and ∆IL is the peakto-peak ripple current in the inductor. The ripple current is determined by the off time, tOFF = 1.8µs, and the inductor value: ∆IL = (VOUT + VD) • tOFF /L IPK is nominally equal to ILIM. However, there is a slight delay in the control circuitry that results in a higher peak current and a more accurate value is: IPK = ILIM + 150ns • (VIN – VOUT)/L These expressions are combined to give the maximum load current that the LT1934 will deliver: IOUT(MAX) = 350mA + 150ns • (VIN – VOUT)/L – 1.8µs • (VOUT + VD)/2L (LT1934) IOUT(MAX) = 90mA + 150ns • (VIN – VOUT)/L – 1.8µs • (VOUT + VD)/2L (LT1934-1) The minimum current limit is used here to be conservative. The third term is generally larger than the second term, so that increasing the inductor value results in a higher output current. This equation can be used to evaluate a chosen inductor or it can be used to choose L for a given maximum load current. The simple, single equation rule given above for choosing L was found by setting ∆IL = ILIM /2.5. This results in IOUT(MAX) ~ 0.8ILIM (ignoring the delay term). Note that this analysis assumes that the inductor current is continuous, which is true if the ripple current is less than the peak current or ∆IL < IPK. 8 U The inductor must carry the peak current without saturating excessively. When an inductor carries too much current, its core material can no longer generate additional magnetic flux (it saturates) and the inductance drops, sometimes very rapidly with increasing current. This condition allows the inductor current to increase at a very high rate, leading to high ripple current and decreased overload protection. Inductor vendors provide current ratings for power inductors. These are based on either the saturation current or on the RMS current that the inductor can carry without dissipating too much power. In some cases it is not clear which of these two determine the current rating. Some data sheets are more thorough and show two current ratings, one for saturation and one for dissipation. For LT1934 applications, the RMS current rating should be higher than the load current, while the saturation current should be higher than the peak inductor current calculated above. Input Capacitor Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT1934 and to force this switching current into a tight local loop, minimizing EMI. The input capacitor must have low impedance at the switching frequency to do this effectively. A 2.2µF ceramic capacitor (1µF for the LT1934-1) satisfies these requirements. If the input source impedance is high, a larger value capacitor may be required to keep input ripple low. In this case, an electrolytic of 10µF or more in parallel with a 1µF ceramic is a good combination. Be aware that the input 1934f W UU LT1934/LT1934-1 APPLICATIO S I FOR ATIO capacitor is subject to large surge currents if the LT1934 circuit is connected to a low impedance supply, and that some electrolytic capacitors (in particular tantalum) must be specified for such use. Output Capacitor and Output Ripple The output capacitor filters the inductor’s ripple current and stores energy to satisfy the load current when the LT1934 is quiescent. In order to keep output voltage ripple low, the impedance of the capacitor must be low at the LT1934’s switching frequency. The capacitor’s equivalent series resistance (ESR) determines this impedance. Choose one with low ESR intended for use in switching regulators. The contribution to ripple voltage due to the ESR is approximately ILIM • ESR. ESR should be less than ~150mΩ for the LT1934 and less than ~500mΩ for the LT1934-1. The value of the output capacitor must be large enough to accept the energy stored in the inductor without a large change in output voltage. Setting this voltage step equal to 1% of the output voltage, the output capacitor must be: COUT > 50 • L • (ILIM /VOUT)2 For example, an LT1934 producing 3.3V with L = 47µH requires 33µF. This value can be relaxed if small circuit size is more important than low output ripple. Sanyo’s POSCAP series in B-case and C-case sizes provides very good performance in a small package for the LT1934. Similar performance in traditional tantalum capacitors requires a larger package (C- or D-case). The Table 3. Capacitor Vendors Vendor Panasonic Phone (714) 373-7366 URL Kemet Sanyo (864) 963-6300 (408) 749-9714 Murata AVX Taiyo Yuden (404) 436-1300 (864) 963-6300 U LT1934-1, with its lower switch current, can use a B-case tantalum capacitor. With a high quality capacitor filtering the ripple current from the inductor, the output voltage ripple is determined by the hysteresis and delay in the LT1934’s feedback comparator. This ripple can be reduced further by adding a small (typically 10pF) phase lead capacitor between the output and the feedback pin. Ceramic Capacitors Ceramic capacitors are small, robust and have very low ESR. However, ceramic capacitors can cause problems when used with the LT1934. Not all ceramic capacitors are suitable. X5R and X7R types are stable over temperature and applied voltage and give dependable service. Other types (Y5V and Z5U) have very large temperature and voltage coefficients of capacitance. In the application circuit they may have only a small fraction of their nominal capacitance and voltage ripple may be much larger than expected. Ceramic capacitors are piezoelectric. The LT1934’s switching frequency depends on the load current, and at light loads the LT1934 can excite the ceramic capacitor at audio frequencies, generating audible noise. If this is unacceptable, use a high performance electrolytic capacitor at the output. The input capacitor can be a parallel combination of a 2.2µF ceramic capacitor and a low cost electrolytic capacitor. The level of noise produced by the LT1934-1 Part Series Ceramic, Polymer, Tantalum Ceramic, Tantalum Ceramic, Polymer, Tantalum Ceramic Ceramic, Tantalum Ceramic 1934f W UU Comments EEF Series www.panasonic.com www.kemet.com www.sanyovideo.com T494, T495 POSCAP www.murata.com www.avxcorp.com www.taiyo-yuden.com TPS Series 9 LT1934/LT1934-1 APPLICATIO S I FOR ATIO when used with ceramic capacitors will be lower and may be acceptable. BOOST C3 SW GND VBOOST – VSW ≅ VOUT MAX VBOOST ≅ VIN + VOUT A final precaution regarding ceramic capacitors concerns the maximum input voltage rating of the LT1934. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT1934 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT1934’s rating. This situation is easily avoided; see the Hot Plugging Safely section. Catch Diode A 0.5A Schottky diode is recommended for the catch diode, D1. The diode must have a reverse voltage rating equal to or greater than the maximum input voltage. The ON Semiconductor MBR0540 is a good choice; it is rated for 0.5A forward current and a maximum reverse voltage of 40V. Schottky diodes with lower reverse voltage ratings usually have a lower forward drop and may result in higher efficiency with moderate to high load currents. However, these diodes also have higher leakage currents. This leakage current mimics a load current at the output and can raise the quiescent current of the LT1934 circuit, especially at elevated temperatures. BOOST Pin Considerations Capacitor C3 and diode D2 are used to generate a boost voltage that is higher than the input voltage. In most cases a 0.1µF capacitor and fast switching diode (such as the 1N4148 or 1N914) will work well. Figure 2 shows two ways to arrange the boost circuit. The BOOST pin must be more than 2.5V above the SW pin for best efficiency. For outputs of 3.3V and above, the standard circuit (Figure 2a) is best. For outputs between 2.8V and 3V, use a 0.22µF capacitor and a small Schottky diode (such as the BAT-54). For lower output voltages the boost diode can be tied to the input (Figure 2b). The circuit in Figure 2a is more efficient because the BOOST pin current comes from a lower voltage source. You must also be sure that the maximum voltage rating of the BOOST pin is not exceeded. The minimum operating voltage of an LT1934 application is limited by the undervoltage lockout (~3V) and by the 10 U D2 LT1934 VIN VIN VOUT W UU (2a) D2 BOOST LT1934 VIN VIN GND SW C3 VOUT 1934 F02 VBOOST – VSW ≅ VIN MAX VBOOST ≅ 2VIN (2b) Figure 2. Two Circuits for Generating the Boost Voltage maximum duty cycle as outlined above. For proper startup, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, or the LT1934 is turned on with its SHDN pin when the output is already in regulation, then the boost capacitor may not be fully charged. Because the boost capacitor is charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. Figure 3 shows a plot of minimum load to start and to run as a function of input voltage. In many cases the discharged output capacitor will present a load to the switcher which will allow it to start. The plots show the worst-case situation where VIN is ramping very slowly. Use a Schottky diode (such as the BAT-54) for the lowest start-up voltage. At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This reduces the minimum input voltage to approximately 300mV above VOUT. At higher load currents, the inductor current is continuous and the duty cycle is limited by the 1934f LT1934/LT1934-1 APPLICATIO S I FOR ATIO Minimum Input Voltage VOUT = 3.3V 6.0 LT1934 VOUT = 3.3V 5.5 TA = 25°C BOOST DIODE TIED TO OUTPUT INPUT VOLTAGE (V) 5.0 4.5 4.0 VIN TO START VIN TO RUN 3.5 3.0 0.1 1 10 100 LOAD CURRENT (mA) 1934 G12 Minimum Input Voltage VOUT = 5V 8 LT1934 VOUT = 5V TA = 25°C 7 BOOST DIODE TIED TO OUTPUT INPUT VOLTAGE (V) VIN TO START 6 VIN TO RUN 5 4 0.1 1 10 100 LOAD CURRENT (mA) 1934 G13 Figure 3. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit maximum duty cycle of the LT1934, requiring a higher input voltage to maintain regulation. Shorted Input Protection If the inductor is chosen so that it won’t saturate excessively, an LT1934 buck regulator will tolerate a shorted output. There is another situation to consider in systems where the output will be held high when the input to the LT1934 is absent. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode OR-ed with the LT1934’s output. If the VIN pin is allowed to float and the SHDN pin is held high (either by a logic signal or because it is tied to U VIN), then the LT1934’s internal circuitry will pull its quiescent current through its SW pin. This is fine if your system can tolerate a few mA in this state. If you ground the SHDN pin, the SW pin current will drop to essentially zero. However, if the VIN pin is grounded while the output is held high, then parasitic diodes inside the LT1934 can pull large currents from the output through the SW pin and the VIN pin. Figure 4 shows a circuit that will run only when the input voltage is present and that protects against a shorted or reversed input. D4 VIN 5 VIN BOOST LT1934 100k 4 SHDN GND 1M 2 SW FB 3 BACKUP 6 VOUT 1 D4: MBR0530 1934 F07 W UU Figure 4. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output; It Also Protects the Circuit from a Reversed Input. The LT1934 Runs Only When the Input is Present PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 5 shows the high current paths in the buck regulator circuit. Note that large, switched currents flow in the power switch, the catch diode (D1) and the input capacitor (C2). The loop formed by these components should be as small as possible. Furthermore, the system ground should be tied to the regulator ground in only one place; this prevents the switched current from injecting noise into the system ground. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components, and tie this ground plane to system ground at one location, ideally at the ground terminal of the output capacitor C1. Additionally, the SW and BOOST nodes should be kept as small as possible. Finally, keep the FB node as small as possible so that the ground pin and 1934f 11 LT1934/LT1934-1 APPLICATIO S I FOR ATIO VIN SW GND (5a) IC1 VIN SW L1 C2 Figure 5. Subtracting the Current When the Switch is On (a) from the Current When the Switch is Off (b) Reveals the Path of the High Frequency Switching Current (c). Keep This Loop Small. The Voltage on the SW and BOOST Nodes Will Also be Switched; Keep These Nodes as Small as Possible. Finally, Make Sure the Circuit is Shielded with a Local Ground Plane SHUTDOWN VIN VOUT SYSTEM GROUND VIAS TO LOCAL GROUND PLANE OUTLINE OF LOCAL GROUND PLANE Figure 6. A Good PCB Layout Ensures Proper, Low EMI Operation ground traces will shield it from the SW and BOOST nodes. Figure 6 shows component placement with trace, ground plane and via locations. Include two vias near the GND pin of the LT1934 to help remove heat from the LT1934 to the ground plane. Hot Plugging Safely The small size, robustness and low impedance of ceramic capacitors make them an attractive option for the input bypass capacitor of LT1934 and LT1934-1 circuits. However, these capacitors can cause problems if the LT1934 12 U VIN SW GND W UU (5b) VSW GND D1 C1 1934 F05 (5c) 1934 F06 is plugged into a live supply (see Linear Technology Application Note 88 for a complete discussion). The low loss ceramic capacitor combined with stray inductance in series with the power source forms an under damped tank circuit, and the voltage at the VIN pin of the LT1934 can ring to twice the nominal input voltage, possibly exceeding the LT1934’s rating and damaging the part. If the input supply is poorly controlled or the user will be plugging the LT1934 into an energized supply, the input network should be designed to prevent this overshoot. 1934f LT1934/LT1934-1 APPLICATIO S I FOR ATIO Figure 7 shows the waveforms that result when an LT1934 circuit is connected to a 24V supply through six feet of 24-gauge twisted pair. The first plot is the response with a 2.2µF ceramic capacitor at the input. The input voltage rings as high as 35V and the input current peaks at 20A. One method of damping the tank circuit is to add another capacitor with a series resistor to the circuit. In Figure 7b CLOSING SWITCH SIMULATES HOT PLUG IIN VIN LT1934 + LOW IMPEDANCE ENERGIZED 24V SUPPLY STRAY INDUCTANCE DUE TO 6 FEET (2 METERS) OF TWISTED PAIR 10µF 35V AI.EI. + 1Ω LT1934 0.1µF 2.2µF 4.7Ω LT1934-1 0.1µF 1µF Figure 7. A Well Chosen Input Network Prevents Input Voltage Overshoot and Ensures Reliable Operation When the LT1934 is Connected to a Live Supply 1934f U an aluminum electrolytic capacitor has been added. This capacitor’s high equivalent series resistance damps the circuit and eliminates the voltage overshoot. The extra capacitor improves low frequency ripple filtering and can slightly improve the efficiency of the circuit, though it is likely to be the largest component in the circuit. An alternative solution is shown in Figure 7c. A 1Ω resistor is VIN 10V/DIV 2.2µF IIN 10A/DIV 10µs/DIV W UU (7a) LT1934 2.2µF (7b) (7c) LT1934-1 1µF (7d) (7e) 1934 F07 13 LT1934/LT1934-1 APPLICATIO S I FOR ATIO added in series with the input to eliminate the voltage overshoot (it also reduces the peak input current). A 0.1µF capacitor improves high frequency filtering. This solution is smaller and less expensive than the electrolytic capacitor. For high input voltages its impact on efficiency is minor, reducing efficiency less than one half percent for a 5V output at full load operating from 24V. Voltage overshoot gets worse with reduced input capacitance. Figure 7d shows the hot plug response with a 1µF ceramic input capacitor, with the input ringing above 40V. The LT1934-1 can tolerate a larger input resistance, such as shown in Figure 7e where a 4.7Ω resistor damps the voltage transient and greatly reduces the input current glitch on the 24V supply. High Temperature Considerations The die temperature of the LT1934 must be lower than the maximum rating of 125°C. This is generally not a concern unless the ambient temperature is above 85°C. For higher temperatures, care should be taken in the layout of the circuit to ensure good heat sinking of the LT1934. The maximum load current should be derated as the ambient temperature approaches 125°C. The die temperature is calculated by multiplying the LT1934 power dissipation by the thermal resistance from junction to ambient. Power dissipation within the LT1934 can be 14 U estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss. The resulting temperature rise at full load is nearly independent of input voltage. Thermal resistance depends on the layout of the circuit board, but a value of 150°C/W is typical. The temperature rise for an LT1934 producing 5V at 250mA is approximately 25°C, allowing it to deliver full load to 100°C ambient. Above this temperature the load current should be reduced. For 3.3V at 250mA the temperature rise is 15°C. Finally, be aware that at high ambient temperatures the external Schottky diode, D1, is likely to have significant leakage current, increasing the quiescent current of the LT1934 converter. Outputs Greater Than 6V For outputs greater than 6V, tie a diode (such as a 1N4148) from the SW pin to VIN to prevent the SW pin from ringing above VIN during discontinuous mode operation. The 12V output circuit in Typical Applications shows the location of this diode. Also note that for outputs above 6V, the input voltage range will be limited by the maximum rating of the BOOST pin. The 12V circuit shows how to overcome this limitation using an additional Zener diode. 1934f W UU LT1934/LT1934-1 TYPICAL APPLICATIO S 3.3V Step-Down Converter D2 VIN 4.5V TO 34V VIN 6.5V TO 34V U BOOST VIN C2 1µF SW 0.1µF L1 100µH LT1934-1 SHDN GND FB D1 10pF 1M VOUT 3.3V 45mA + ON OFF C1 22µF 604k C1: TAIYO YUDEN JMK316BJ226ML C2: TAIYO YUDEN GMK316BJ105ML D1: ZETEX ZHCS400 OR ON SEMI MBR0540 D2: CENTRAL CMDSH-3 L1: COILCRAFT DO1608C-104 OR WURTH ELECTRONICS WE-PD4 TYPE S 1934 TA04 5V Step-Down Converter D2 BOOST VIN C2 1µF SW 0.1µF L1 150µH LT1934-1 SHDN GND FB D1 10pF 1M VOUT 5V 45mA + ON OFF C1 22µF 332k C1: TAIYO YUDEN JMK316BJ226ML C2: TAIYO YUDEN GMK316BJ105ML D1: ZETEX ZHCS400 OR ON SEMI MBR0540 D2: CENTRAL CMPD914 L1: COILCRAFT DO1608C-154 OR WURTH ELECTRONICS WE-PD4 TYPE S 1934 TA05 1934f 15 LT1934/LT1934-1 TYPICAL APPLICATIO S 1.8V Step-Down Converter D2 VIN 3.6V TO 16V Loop Powered 3.3V Supply with Additional Isolated Output D3 ISOLATED OUT 3V 10µF 3mA BOOST VIN 14V TO 32V