LT1936EMS8E-PBF

LT1936 1.4A, 500kHz Step-Down Switching Regulator FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTION The LT®1936 is a current mode PWM step-down DC/DC converter with an internal 1.9A power switch, packaged in a tiny, thermally enhanced 8-lead MSOP The wide in. put range of 3.6V to 36V makes the LT1936 suitable for regulating power from a wide variety of sources, including automotive batteries, 24V industrial supplies and unregulated wall adapters. Its high operating frequency allows the use of small, low cost inductors and ceramic capacitors, resulting in low, predictable output ripple. Cycle-by-cycle current limit, frequency foldback and thermal shutdown provide protection against shorted outputs, and soft-start eliminates input current surge during start-up. Transient response can be optimized by using external compensation components, or board space can be minimized by using internal compensation. The low current (30V), the saturation current should be above 2.6A. To keep the efficiency high, the series resistance (DCR) should be less than 0.1Ω, and the core material should be intended for high frequency applications. Table 1 lists several vendors and suitable types. Table 1. Inductor Vendors VENDOR Murata TDK Toko URL www.murata.com www.component.tdk.com www.toko.com PART SERIES LQH55D SLF7045 SLF10145 D62CB D63CB D75C D75F CR54 CDRH74 CDRH6D38 CR75 TYPE Open Shielded Shielded Shielded Shielded Shielded Open Open Shielded Shielded Open where VD is the forward voltage drop of the catch diode (~0.5V) and VSW is the voltage drop of the internal switch (~0.5V at maximum load). This leads to a minimum input voltage of: VIN(MIN) = VOUT + VD – VD + VSW DCMAX Sumida www.sumida.com with DCMAX = 0.87. The maximum input voltage is determined by the absolute maximum ratings of the VIN and BOOST pins and by the minimum duty cycle DCMIN = 0.08: VIN(MAX ) = VOUT + VD – VD + VSW DCMIN Note that this is a restriction on the operating input voltage; the circuit will tolerate transient inputs up to the absolute maximum ratings of the VIN and BOOST pins. Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger value provides a slightly higher maximum load current and will reduce the output voltage ripple. If your load is lower than 1.2A, then you can decrease the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. There are several graphs in the Typical Performance Characteristics section of this data sheet that show the maximum load current as a function of input voltage and inductor value for several popular output voltages. Low 1936fd 8 LT1936 APPLICATIONS INFORMATION inductance may result in discontinuous mode operation, which is okay but further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), there is a minimum inductance required to avoid subharmonic oscillations. Choosing L greater than 1.6 (VOUT + VD) μH prevents subharmonic oscillations at all duty cycles. Catch Diode A 1A Schottky diode is recommended for the catch diode, D1. The diode must have a reverse voltage rating equal to or greater than the maximum input voltage. The ON Semiconductor MBRM140 is a good choice. It is rated for 1A DC at a case temperature of 110°C and 1.5A at a case temperature of 95°C. Diode Incorporated’s DFLS140L is rated for 1.1A average current; the DFLS240L is rated for 2A average current. The average diode current in an LT1936 application is approximately IOUT (1 – DC). Input Capacitor Bypass the input of the LT1936 circuit with a 4.7μF or higher value ceramic capacitor of X7R or X5R type. Y5V types have poor performance over temperature and applied voltage, and should not be used. A 4.7μF ceramic is adequate to bypass the LT1936 and will easily handle the ripple current. However, if the input power source has high impedance, or there is significant inductance due to long wires or cables, additional bulk capacitance may be necessary. This can be provided with a low performance electrolytic capacitor. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT1936 and to force this very high frequency switching current into a tight local loop, minimizing EMI. A 4.7μF capacitor is capable of this task, but only if it is placed close to the LT1936 and the catch diode; see the PCB Layout section. A second precaution regarding the ceramic input capacitor concerns the maximum input voltage rating of the LT1936. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT1936 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT1936’s voltage rating. This situation is easily avoided; see the Hot Plugging Safety section. For space sensitive applications, a 2.2μF ceramic capacitor can be used for local bypassing of the LT1936 input. However, the lower input capacitance will result in increased input current ripple and input voltage ripple, and may couple noise into other circuitry. Also, the increased voltage ripple will raise the minimum operating voltage of the LT1936 to ~3.7V. Output Capacitor The output capacitor has two essential functions. Along with the inductor, it filters the square wave generated by the LT1936 to produce the DC output. In this role it determines the output ripple, and low impedance at the switching frequency is important. The second function is to store energy in order to satisfy transient loads and stabilize the LT1936’s control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A good value is: COUT = 150 VOUT where COUT is in μF Use X5R or X7R types. This choice . will provide low output ripple and good transient response. Transient performance can be improved with a high value capacitor if the compensation network is also adjusted to maintain the loop bandwidth. A lower value of output capacitor can be used, but transient performance will suffer. With an external compensation network, the loop gain can be lowered to compensate for the lower capacitor value. When using the internal compensation network, the lowest value for stable operation is: COUT > 66 VOUT 1936fd 9 LT1936 APPLICATIONS INFORMATION Table 2. Capacitor Vendors VENDOR Panasonic PHONE (714) 373-7366 URL www.panasonic.com PART SERIES Ceramic, Polymer, Tantalum Ceramic, Tantalum Ceramic, Polymer, Tantalum Ceramic Ceramic, Tantalum Ceramic TPS Series COMMENTS EEF Series Kemet Sanyo (864) 963-6300 (408) 749-9714 www.kemet.com www.sanyovideo.com T494, T495 POSCAP Murata AVX Taiyo Yuden (404) 436-1300 www.murata.com www.avxcorp.com (864) 963-6300 www.taiyo-yuden.com This is the minimum output capacitance required, not the nominal capacitor value. For example, a 3.3V output requires 20μF of output capacitance. If a small 22μF 6.3V , ceramic capacitor is used, the circuit may be unstable because the effective capacitance is lower than the nominal capacitance when biased at 3.3V. Look carefully at the capacitor’s data sheet to find out what the actual capacitance is under operating conditions (applied voltage and temperature). A physically larger capacitor, or one with a higher voltage rating, may be required. High performance electrolytic capacitors can be used for the output capacitor. Low ESR is important, so choose one that is intended for use in switching regulators. The ESR should be specified by the supplier, and should be 0.05Ω or less. Such a capacitor will be larger than a ceramic capacitor and will have a larger capacitance, because the capacitor must be large to achieve low ESR. Table 2 lists several capacitor vendors. Frequency Compensation The LT1936 uses current mode control to regulate the output. This simplifies loop compensation. In particular, the LT1936 does not require the ESR of the output capacitor for stability, so you are free to use ceramic capacitors to achieve low output ripple and small circuit size. Frequency compensation is provided by the components tied to the VC pin, as shown in Figure 1. Generally a capacitor (CC) and a resistor (RC) in series to ground are used. In addition, there may be lower value capacitor in parallel. This capacitor (CF) is not part of the loop compensation but is used to filter noise at the switching frequency, and is required only if a phase-lead capacitor is used or if the output capacitor has high ESR. An alternative to using external compensation components is to use the internal RC network by tying the COMP pin to the VC pin. This reduces component count but does not provide the optimum transient response when the output capacitor value is high, and the circuit may not be stable when the output capacitor value is low. If the internal compensation network is not used, tie COMP to ground or leave it floating. Loop compensation determines the stability and transient performance. Designing the compensation network is a bit LT1936 CURRENT MODE POWER STAGE gm = 2mho SW ERROR AMPLIFIER FB ESR gm = 250μmho 600k C1 50k VC CF RC CC 1936 F01 OUTPUT R1 CPL 150pF GND POLYMER OR TANTALUM R2 CERAMIC COMP Figure 1. Model for Loop Response 1936fd 10 – + 1.2V + C1 LT1936 APPLICATIONS INFORMATION complicated and the best values depend on the application and in particular the type of output capacitor. A practical approach is to start with one of the circuits in this data sheet that is similar to your application and tune the compensation network to optimize the performance. Stability should then be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the stability using a transient load. Figure 1 shows an equivalent circuit for the LT1936 control loop. The error amplifier is a transconductance amplifier with finite output impedance. The power section, consisting of the modulator, power switch and inductor, is modeled as a transconductance amplifier generating an output COUT = 22μF (AVX 1210ZD226MAT) (2a) COMP VC VOUT 100mV/DIV current proportional to the voltage at the VC pin. Note that the output capacitor integrates this current, and that the capacitor on the VC pin (CC) integrates the error amplifier output current, resulting in two poles in the loop. In most cases a zero is required and comes from either the output capacitor ESR or from a resistor RC in series with CC. This simple model works well as long as the value of the inductor is not too high and the loop crossover frequency is much lower than the switching frequency. A phase lead capacitor (CPL) across the feedback divider may improve the transient response. Figure 2 compares the transient response across several output capacitor choices and compensation schemes. In each case the load current is stepped from 200mA to 800mA and back to 200mA. COUT = 22μF ×2 (2b) COMP VC VOUT 100mV/DIV COUT = 150μF (4TPC150M) (2c) COMP VC VOUT 100mV/DIV COUT = 150μF (4TPC150M) (2d) COMP VC 220k 100pF VOUT 100mV/DIV IOUT 500mA/DIV 800mA 200mA 1936 F02 50μs/DIV Figure 2. Transient Load Response of the LT1936 with Different Output Capacitors as the Load Current is Stepped from 200mA to 800mA. VOUT = 3.3V 1936fd 11 LT1936 APPLICATIONS INFORMATION BOOST Pin Considerations Capacitor C3 and diode D2 are used to generate a boost voltage that is higher than the input voltage. In most cases a 0.22μF capacitor and fast switching diode (such as the 1N4148 or 1N914) will work well. Figure 3 shows two ways to arrange the boost circuit. The BOOST pin must be at least 2.3V above the SW pin for best efficiency. For outputs of 3V and above, the standard circuit (Figure 3a) is best. For outputs between 2.8V and 3V, use a 0.47μF capacitor and a Schottky diode. For lower output voltages the boost diode can be tied to the input (Figure 3b), or to another supply greater than 2.8V. The circuit in Figure 3a is more efficient because the BOOST pin current comes from a lower voltage. You must also be sure that the maximum voltage rating of the BOOST pin is not exceeded. A 2.5V output presents a special case. This is a popular output voltage, and the advantage of connecting the boost circuit to the output is that the circuit will accept a 36V maximum input voltage rather than 20V (due to the BOOST pin rating). However, 2.5V is marginally adequate to support the boosted drive stage at low ambient temperatures. Therefore, special care and some restrictions on operation are necessary when powering the BOOST pin from a 2.5V output. Minimize the voltage loss in the boost D2 circuit by using a 1μF boost capacitor and a good, low drop Schottky diode (such as the ON Semi MBR0540). Because the required boost voltage increases at low temperatures, the circuit will supply only 1A of output current when the ambient temperature is –45°C, increasing to 1.2A at 0°C. Also, the minimum input voltage to start the boost circuit is higher at low temperature. See the Typical Applications section for a 2.5V schematic and performance curves. The minimum operating voltage of an LT1936 application is limited by the undervoltage lockout (~3.45V) and by the maximum duty cycle as outlined above. For proper start-up, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, or the LT1936 is turned on with its SHDN pin when the output is already in regulation, then the boost capacitor may not be fully charged. Because the boost capacitor is charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. Figure 4 shows a plot of minimum load to start and to run as a function of input voltage. In many cases the discharged output capacitor will present a load to the switcher, which will allow it to start. The plots show the worst-case situation where VIN is ramping very slowly. For lower start-up voltage, the boost diode can be tied to VIN; however, this restricts the input range to one-half of the absolute maximum rating of the BOOST pin. At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This reduces the minimum input voltage to approximately 300mV above VOUT. At higher load currents, the inductor current is continuous and the duty cycle is limited by the maximum duty cycle of the LT1936, requiring a higher input voltage to maintain regulation. Soft-Start The SHDN pin can be used to soft-start the LT1936, reducing the maximum input current during start-up. The SHDN pin is driven through an external RC filter to create a voltage ramp at this pin. Figure 5 shows the start-up waveforms with and without the soft-start circuit. By choosing a large 1936fd BOOST LT1936 VIN VIN GND VBOOST – VSW ≅ VOUT MAX VBOOST ≅ VIN + VOUT D2 (3a) SW C3 VOUT BOOST LT1936 VIN VIN GND SW C3 VOUT 1933 F03 VBOOST – VSW ≅ VIN MAX VBOOST ≅ 2VIN (3b) Figure 3. Two Circuits for Generating the Boost Voltage 12 LT1936 APPLICATIONS INFORMATION Minimum Input Voltage VOUT = 3.3V 6.0 5.5 INPUT VOLTAGE (V) 5.0 4.5 4.0 3.5 3.0 0 10 100 LOAD CURRENT (mA) 1000 1936 F04a Minimum Input Voltage VOUT = 5V 8 VOUT = 5V TA = 25°C L = 15μH TO START VOUT = 3.3V TA = 25°C L = 10μH INPUT VOLTAGE (V) TO START 7 6 TO RUN 5 TO RUN 4 1 10 100 LOAD CURRENT (mA) 1000 1936 F04b Figure 4. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit RUN 5V/DIV RUN SHDN GND VOUT 5V/DIV 50μs/DIV 1936 F05a IIN 500mA/DIV RUN 15k SHDN 0.22μF GND RUN 5V/DIV IIN 500mA/DIV VOUT 5V/DIV 0.5ms/DIV 1936 F05b Figure 5. To Soft-Start the LT1936, Add a Resistor and Capacitor to the SHDN Pin. VIN = 12V, VOUT = 3.3V, COUT = 2 × 22μF RLOAD = 3.3Ω , RC time constant, the peak start-up current can be reduced to the current that is required to regulate the output, with no overshoot. Choose the value of the resistor so that it can supply 60μA when the SHDN pin reaches 2.3V. Shorted and Reversed Input Protection If the inductor is chosen so that it won’t saturate excessively, an LT1936 buck regulator will tolerate a shorted output. There is another situation to consider in systems where the output will be held high when the input to the LT1936 is absent. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode OR-ed with the LT1936’s output. If the VIN pin is allowed to float and the SHDN pin is held high (either by a logic signal or because it is tied to VIN), then the LT1936’s internal circuitry will pull its quiescent current through its SW pin. This is fine if your system can tolerate a few mA in this state. If you ground 1936fd 13 LT1936 APPLICATIONS INFORMATION the SHDN pin, the SW pin current will drop to essentially zero. However, if the VIN pin is grounded while the output is held high, then parasitic diodes inside the LT1936 can pull large currents from the output through the SW pin and the VIN pin. Figure 6 shows a circuit that will run only when the input voltage is present and that protects against a shorted or reversed input. D4 MBRS140 VIN L1 VIN SHDN VC COMP GND FB BACKUP BOOST LT1936 SW VOUT OUT VIAS 1936 F07 IN MINIMIZE LT1936 C2, D1 LOOP D2 C2 GND R4 C3 R2 R1 C1 D1 GND Figure 7. A Good PCB Layout Ensures Low EMI Operation High Temperature Considerations 1936 F06 Figure 6. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output; It Also Protects the Circuit from a Reversed Input. The LT1936 Runs Only When the Input is Present PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 7 shows the recommended component placement with trace, ground plane and via locations. Note that large, switched currents flow in the LT1936’s VIN and SW pins, the catch diode (D1) and the input capacitor (C2). The loop formed by these components should be as small as possible. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components. The SW and BOOST nodes should be as small as possible. Finally, keep the FB and VC nodes small so that the ground traces will shield them from the SW and BOOST nodes. The Exposed Pad on the bottom of the package must be soldered to ground so that the pad acts as a heat sink. To keep thermal resistance low, extend the ground plane as much as possible, and add thermal vias under and near the LT1936 to additional ground planes within the circuit board and on the bottom side. The die temperature of the LT1936 must be lower than the maximum rating of 125°C (150°C for the H grade). This is generally not a concern unless the ambient temperature is above 85°C. For higher temperatures, care should be taken in the layout of the circuit to ensure good heat sinking of the LT1936. The maximum load current should be derated as the ambient temperature approaches 125°C (150°C for the H grade). The die temperature is calculated by multiplying the LT1936 power dissipation by the thermal resistance from junction to ambient. Power dissipation within the LT1936 can be estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss. The resulting temperature rise at full load is nearly independent of input voltage. Thermal resistance depends on the layout of the circuit board, but values from 40°C/W to 60°C/W are typical. Die temperature rise was measured on a 4-layer, 5cm × 6.5cm circuit board in still air at a load current of 1.4A. For 12V input to 3.3V output the die temperature elevation above ambient was 26°C; for 24V in to 3.3V out the rise was 31°C; for 12V in to 5V the rise was 31°C and for 24V in to 5V the rise was 34°C. 1936fd 14 LT1936 APPLICATIONS INFORMATION Hot Plugging Safely The small size, robustness and low impedance of ceramic capacitors make them an attractive option for the input bypass capacitor of LT1936 circuits. However, these capacitors can cause problems if the LT1936 is plugged into a live supply (see Linear Technology Application Note 88 for a complete discussion). The low loss ceramic capacitor combined with stray inductance in series with the power source forms an under damped tank circuit, and the voltage at the VIN pin of the LT1936 can ring to twice the nominal CLOSING SWITCH SIMULATES HOT PLUG IIN VIN LT1936 input voltage, possibly exceeding the LT1936’s rating and damaging the part. If the input supply is poorly controlled or the user will be plugging the LT1936 into an energized supply, the input network should be designed to prevent this overshoot. Figure 8 shows the waveforms that result when an LT1936 circuit is connected to a 24V supply through six feet of 24-gauge twisted pair. The first plot is the response with a 4.7μF ceramic capacitor at the input. The input voltage rings as high as 50V and the input current peaks at 26A. DANGER VIN 20V/DIV RINGING VIN MAY EXCEED ABSOLUTE MAXIMUM RATING OF THE LT1936 + 4.7μF LOW IMPEDANCE ENERGIZED 24V SUPPLY STRAY INDUCTANCE DUE TO 6 FEET (2 METERS) OF TWISTED PAIR IIN 10A/DIV 20μs/DIV (8a) LT1936 VIN 20V/DIV + 22μF 35V AI.EI. + 4.7μF IIN 10A/DIV (8b) 20μs/DIV 0.7Ω LT1936 VIN 20V/DIV + 0.1μF 4.7μF IIN 10A/DIV (8c) 20μs/DIV 1936 F08 Figure 8. A Well Chosen Input Network Prevents Input Voltage Overshoot and Ensures Reliable Operation When the LT1936 is Connected to a Live Supply 1936fd 15 LT1936 APPLICATIONS INFORMATION One method of damping the tank circuit is to add another capacitor with a series resistor to the circuit. In Figure 8b an aluminum electrolytic capacitor has been added. This capacitor’s high equivalent series resistance damps the circuit and eliminates the voltage overshoot. The extra capacitor improves low frequency ripple filtering and can slightly improve the efficiency of the circuit, though it is likely to be the largest component in the circuit. An alternative solution is shown in Figure 8c. A 0.7Ω resistor is added in series with the input to eliminate the voltage overshoot (it also reduces the peak input current). A 0.1μF capacitor improves high frequency filtering. This solution is smaller and less expensive than the electrolytic capacitor. For high input voltages its impact on efficiency is minor, reducing efficiency by one percent for a 5V output at full load operating from 24V. Other Linear Technology Publications Application Notes 19, 35 and 44 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design Note 100 shows how to generate a bipolar output supply using a buck regulator. Outputs Greater Than 6V For outputs greater than 6V, add a resistor of 1k to 2.5k across the inductor to damp the discontinuous ringing of the SW node, preventing unintended SW current. The 12V Step-Down Converter circuit in the Typical Applications section shows the location of this resistor. Also note that for outputs above 6V, the input voltage range will be limited by the maximum rating of the BOOST pin. The 12V circuit shows how to overcome this limitation using an additional Zener diode. TYPICAL APPLICATIONS 3.3V Step-Down Converter D2 VIN 4.5V TO 36V VIN ON OFF C1 4.7μF SHDN LT1936 COMP VC FB GND R2 10k C2 47μF BOOST SW D1 R1 17.4k C3 0.22μF L1 10μH VOUT 3.3V 1.2A 1936 TA03 1936fd 16 LT1936 TYPICAL APPLICATIONS 5V Step-Down Converter D2 VIN 6.3V TO 36V VIN ON OFF C1 4.7μF SHDN LT1936 COMP VC FB GND R2 10k C2 22μF BOOST SW D1 R1 31.6k C3 0.22μF L1 15μH VOUT 5V 1.2A 1936 TA04 1.8V Step-Down Converter VIN 3.6V TO 20V VIN ON OFF C1 4.7μF SHDN LT1936 COMP VC FB GND R2 20k C2 47μF ×2 1936 TA05a Efficiency, 1.8V Output 90 VOUT = 1.8V TA = 25°C VIN = 5V VIN = 12V 70 1.0 2.0 D2 C3 0.22μF BOOST SW L1 4.7μH R1 10k D1 EFFICIENCY (%) VOUT 1.8V 1.3A 80 1.5 POWER LOSS (W) 60 POWER LOSS 50 0 0.5 1 LOAD CURRENT (A) 0.5 D1: DFLS140L D2: 1N4148 L1: TOKO D63CB 0 1.5 1936 TA05b 1.2V Step-Down Converter VIN 3.6V TO 20V VIN ON OFF C1 4.7μF SHDN LT1936 COMP VC FB GND 100k C2 47μF ×2 1936 TA06a Efficiency, 1.2V Output 80 VOUT = 1.2V TA = 25°C VIN = 5V EFFICIENCY (%) 70 65 60 0.5 55 POWER LOSS 0 1.5 1936 TA06b D2 C3 0.22μF 2.0 BOOST SW L1 3.3μH D1 75 VOUT 1.2V 1.3A 1.5 POWER LOSS (W) VIN = 12V 1.0 D1: DFLS140L D2: 1N4148 L1: TOKO D63CB 50 0 0.5 1 LOAD CURRENT (A) 1936fd 17 LT1936 TYPICAL APPLICATIONS 2.5V Step-Down Converter D2 VIN 3.6V TO 36V VIN ON OFF C1 4.7μF SHDN LT1936 COMP VC FB GND R2 10k C2 47μF BOOST SW D1 R1 11k C3 1μF L1 6.2μH VOUT 2.5V 1.2A TA > 0°C D1: DFLS140L D2: MBRO540 L1: TOKO D63CB 1936 TA07a Efficiency, 2.5V Output 100 VOUT = 2.5V TA = 25°C 5.5 Minimum Input Voltage VOUT = 2.5V 90 VIN = 5V 80 VIN = 12V INPUT VOLTAGE (V) EFFICIENCY (%) 5.0 TO START TA = –45°C 4.5 TO START TA = 25°C TO RUN TA = –45°C 3.5 TO RUN TA = 25°C 4.0 70 60 0 0.5 1.0 LOAD CURRENT (A) 1.5 1936 TA07b 3.0 1 100 10 LOAD CURRENT (mA) 1000 1936 TA07c 12V Step-Down Converter D2 VIN 14.5V TO 36V VIN ON OFF C1 2.2μF SHDN LT1936 COMP VC FB GND R2 20k R1 182k C2 22μF BOOST SW D1 C3 0.22μF L1 22μH 1.8k VOUT 12V 1.2A D3 6.8V D1: MBRM140 D2: 1N4148 D3: CMDZ5235B 1936 TA08 1936fd 18 LT1936 PACKAGE DESCRIPTION MS8E Package 8-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1662 Rev E) BOTTOM VIEW OF EXPOSED PAD OPTION 1 2.06 ± 0.102 (.081 ± .004) 1.83 ± 0.102 (.072 ± .004) 0.29 REF 2.794 ± 0.102 (.110 ± .004) 0.889 ± 0.127 (.035 ± .005) 0.05 REF 5.23 (.206) MIN 2.083 ± 0.102 3.20 – 3.45 (.082 ± .004) (.126 – .136) 8 DETAIL “B” CORNER TAIL IS PART OF THE LEADFRAME FEATURE. FOR REFERENCE ONLY NO MEASUREMENT PURPOSE DETAIL “B” 0.42 ± 0.038 (.0165 ± .0015) TYP 0.65 (.0256) BSC 3.00 ± 0.102 (.118 ± .004) (NOTE 3) 8 7 65 0.52 (.0205) REF RECOMMENDED SOLDER PAD LAYOUT DETAIL “A” 0° – 6° TYP 4.90 ± 0.152 (.193 ± .006) 3.00 ± 0.102 (.118 ± .004) (NOTE 4) 0.254 (.010) GAUGE PLANE 1 0.53 ± 0.152 (.021 ± .006) DETAIL “A” 0.18 (.007) SEATING PLANE 0.22 – 0.38 (.009 – .015) TYP 1.10 (.043) MAX 23 4 0.86 (.034) REF NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 0.65 (.0256) BSC 0.1016 ± 0.0508 (.004 ± .002) MSOP (MS8E) 0908 REV E 1936fd Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LT1936 TYPICAL APPLICATION 2.5V Step-Down Converter VIN 3.6V TO 20V VIN ON OFF C1 4.7μF SHDN LT1936 COMP VC FB GND R2 10k C2 47μF D2 C3 0.22μF 5.5 Minimum Input Voltage VOUT = 2.5V CONNECTING THE BOOST CIRCUIT TO THE INPUT LOWERS THE MINIMUM INPUT VOLTAGE TO RUN AND TO START TO LESS THAN 3.7V AT ALL LOADS BOOST SW L1 8.2μH R1 11k 5.0 INPUT VOLTAGE (V) VOUT 2.5V 1.3A D1 4.5 4.0 3.5 D1: DFLS140L D2: 1N4148 L1: TOKO D63CB 1936 TA09a 3.0 1 100 10 LOAD CURRENT (mA) 1000 1936 TA09b RELATED PARTS PART NUMBER LT1676 LT1765 LT1766 LT1767 LT1776 LT1933 LT1940 LT1956 LT1976 LT3010 LTC®3407 LTC3412 LTC3414 LT3430/LT3431 DESCRIPTION 60V, 440mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter 25V, 2.75A (IOUT), 1.25MHz, High Efficiency Step-Down DC/DC Converter 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 25V, 1.2A (IOUT), 1.25MHz, High Efficiency Step-Down DC/DC Converter 40V, 550mA (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 600mA, 500kHz, Step-Down Switching Regulator in SOT-23 25V, Dual 1.4A (IOUT), 1.1MHz, High Efficiency Step-Down DC/DC Converter 60V, 1.2A (IOUT), 500kHz, High Efficiency Step-Down DC/DC Converter 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter with Burst Mode® Operation 80V, 50mA, Low Noise Linear Regulator Dual 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 4A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 60V, 2.75A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC Converters COMMENTS VIN: 7.4V to 60V, VOUT(MIN) = 1.24V, IQ = 3.2mA, ISD = 2.5μA, SO-8 Package VIN: 3V to 25V, VOUT(MIN) = 1.20V, IQ = 1mA, ISD = 15μA, SO-8 and 16-Lead TSSOPE Packages VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD = 25μA, 16-Lead TSSOP/TSSOPE Packages VIN: 3V to 25V, VOUT(MIN) = 1.20V, IQ = 1mA, ISD = 6μA, MS8/MS8E Packages VIN: 7.4V to 40V, VOUT(MIN) = 1.24V, IQ = 3.2mA, ISD = 30μA, N8/SO-8 Packages VIN: 3.6V to 36V, VOUT(MIN) = 1.25V, IQ = 1.6mA, ISD < 1μA, ThinSOT™ Package VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 3.8mA, ISD < 1μA, 16-Lead TSSOPE Package VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD = 25μA, 16-Lead TSSOP/TSSOPE Packages VIN: 3.3V to 60V, VOUT(MIN) = 1.20V, IQ = 100μA, ISD < 1μA, 16-Lead TSSOPE Package VIN: 1.5V to 80V, VOUT(MIN) = 1.28V, IQ = 30μA, ISD < 1μA, MS8E Package VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40μA, ISD < 1μA, 10-Lead MSE Package VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60μA, ISD < 1μA, 16-Lead TSSOPE Package VIN: 2.3V to 5.5V, VOUT(MIN) = 0.8V, IQ = 64μA, ISD < 1μA, 20-Lead TSSOPE Package VIN: 5.5V to 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA, ISD = 30μA, 16-Lead TSSOPE Package Burst Mode is a registered trademark of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation. 1936fd 20 Linear Technology Corporation (408) 432-1900 ● FAX: (408) 434-0507 ● LT 1108 REV D • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 www.linear.com © LINEAR TECHNOLOGY CORPORATION 2006