LT3475EFE-TRPBF

LT3475/LT3475-1 Dual Step-Down 1.5A LED Driver FEATURES ■ ■ DESCRIPTIO ■ ■ ■ ■ ■ ■ ■ ■ True Color PWMTM Delivers Constant Color with 3000:1 Dimming Range Wide Input Range: 4V to 36V Operating, 40V Maximum Accurate and Adjustable Control of LED Current from 50mA to 1.5A High Side Current Sense Allows Grounded Cathode LED Operation Open LED (LT3475) and Short Circuit Protection LT3475-1 Drives LED Strings Up to 25V Accurate and Adjustable 200kHz to 2MHz Switching Frequency Anti-Phase Switching Reduces Ripple Uses Small Inductors and Ceramic Capacitors Available in the Compact 20-Lead TSSOP Thermally Enhanced Surface Mount Package The LT®3475/LT3475-1 are dual step-down DC/DC converters designed to operate as a constant-current source. An internal sense resistor monitors the output current allowing accurate current regulation ideal for driving high current LEDs. The high side current sense allows grounded cathode LED operation. High output current accuracy is maintained over a wide current range, from 50mA to 1.5A, allowing a wide dimming range. Unique PWM circuitry allows a dimming range of 3000:1, avoiding the color shift normally associated with LED current dimming. The high switching frequency offers several advantages, permitting the use of small inductors and ceramic capacitors. Small inductors combined with the 20 lead TSSOP surface mount package save space and cost versus alternative solutions. The constant switching frequency combined with low-impedance ceramic capacitors result in low, predictable output ripple. With its wide input range of 4V to 36V, the LT3475/LT3475-1 regulate a broad array of power sources. A current mode PWM architecture provides fast transient response and cycle-by-cycle current limiting. Frequency foldback and thermal shutdown provide additional protection. APPLICATIO S ■ ■ ■ ■ Automotive and Avionic Lighting Architectural Detail Lighting Display Backlighting Constant-Current Sources , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Patents Pending. TYPICAL APPLICATIO VIN 5V TO 36V 4.7μF VIN BOOST1 0.22μF 10μH SW1 LT3475 Dual Step-Down 1.5A LED Driver Efficiency SHDN BOOST2 0.22μF 10μH SW2 95 90 85 EFFICIENCY (%) 80 75 70 65 0.1μF 24.3k 1.5A LED CURRENT 2.2μF VIN = 12V OUT1 LED1 *DIMMING CONTROL PWM1 VC1 REF VADJ1 GND OUT2 LED2 PWM2 VC2 RT VADJ2 DIMMING* CONTROL 2.2μF 0.1μF 1.5A LED CURRENT 60 55 0 0.5 1 1.5 3475 TA01b 3475 TA01 *SEE APPLICATIONS SECTION FOR DETAILS fSW = 600kHz U TWO SERIES CONNECTED WHITE 1.5A LEDS SINGLE WHITE 1.5A LED LED CURRENT (A) 3475fb U U 1 LT3475/LT3475-1 ABSOLUTE (Note 1) TOP VIEW AXI U RATI GS VIN Pin .........................................................(-0.3V), 40V BOOST Pin Voltage ...................................................60V BOOST Above SW Pin ...............................................30V OUT, LED, Pins (LT3475) ...........................................15V OUT, LED Pins (LT3475-1).........................................25V PWM Pin ...................................................................15V VADJ Pin ......................................................................6V VC, RT, REF Pins ..........................................................3V SHDN Pin ...................................................................VIN Maximum Junction Temperature (Note 2)............. 125°C Operating Temperature Range (Note 3) LT3475E/LT3475E-1 ............................. –40°C to 85°C LT3475I/LT3475I-1 ............................. –40°C to 125°C Storage Temperature Range................... –65°C to 150°C Lead Temperature Range (Soldering, 10 sec) ....... 300°C ORDER INFORMATION LEAD FREE FINISH LT3475EFE#PBF LT3475IFE#PBF LT3475EFE-1#PBF LT3475IFE-1#PBF TAPE AND REEL LT3475EFE#TRPBF LT3475IFE#TRPBF LT3475EFE-1#TRPBF LT3475IFE-1#TRPBF PART MARKING* LT3475EFE LT3475IFE LT3475FE-1 LT3475FE-1 PACKAGE DESCRIPTION 20-Lead Plastic TSSOP 20-Lead Plastic TSSOP 20-Lead Plastic TSSOP 20-Lead Plastic TSSOP TEMPERATURE RANGE –40°C to 85°C –40°C to 125°C –40°C to 85°C –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3) PARAMETER Minimum Input Voltage Input Quiescent Current Shutdown Current Not Switching SHDN = 0.3V, VBOOST = VOUT = 0V CONDITIONS ● ELECTRICAL CHARACTERISTICS 2 U WW W PIN CONFIGURATION OUT1 LED1 BOOST1 SW1 VIN VIN SW2 BOOST2 LED2 1 2 3 4 5 6 7 8 9 21 20 PWM1 19 VADJ1 18 VC1 17 REF 16 SHDN 15 GND 14 RT 13 VC2 12 VADJ2 11 PWM2 OUT2 10 FE PACKAGE 20-LEAD PLASTIC TSSOP TJMAX = 125°C, θJA = 30°C/W, θJC = 8°C/W EXPOSED PAD (PIN 21) IS GROUND AND MUST BE ELECTRICALLY CONNECTED TO THE PCB. MIN TYP 3.7 6 0.01 MAX 4 8 2 UNITS V mA μA 3475fb LT3475/LT3475-1 ELECTRICAL CHARACTERISTICS PARAMETER LED Pin Current The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3) CONDITIONS VADJ Tied to VREF • 2/3 VADJ Tied to VREF • 7/30 LT3475E/LT3475E-1 0°C to 85°C ● ● MIN 0.97 0.94 0.336 0.325 0.31 1.22 TYP 1.00 0.350 MAX 1.03 1.04 0.364 0.375 0.385 1.27 UNITS A A A A A V %/V %/μA REF Voltage Reference Voltage Line Regulation Reference Voltage Load Regulation VADJ Pin Bias Current (Note 4) Switching Frequency Maximum Duty Cycle RT = 24.3k RT = 24.3k RT = 4.32k RT = 100k RT = 24.3k RT = 24.3k, VOUT = 0V 4V < VIN < 40V 0 < IREF < 500μA ● 1.25 0.05 0.0002 ● ● ● 40 530 90 600 95 80 98 180 80 2.5 2.6 9 0.8 0.8 50 50 500 1 2.6 1.8 400 640 nA kHz % % % Switching Phase Foldback Frequency SHDN Threshold (to Switch) SHDN Pin Current (Note 5) PWM Threshold VC Switching Threshold VC Source Current VC Sink Current LED to VC Transresistance LED to VC Current Gain VC to Switch Current Gain VC Clamp Voltage VC Pin Current in PWM Mode OUT Pin Clamp Voltage (LT3475) OUT Pin Current in PWM Mode Switch Current Limit (Note 6) Switch VCESAT BOOST Pin Current Switch Leakage Current Minimum Boost Voltage Above SW 150 210 2.74 11 1.2 Deg kHz V μA V V μA μA V/A mA/μA A/V V VSHDN = 2.6V 7 0.3 VC = 1V VC = 1V VC = 1V, VPWM = 0.3V VOUT = 4V, VPWM = 0.3V ISW =1.5A ISW =1.5A ● 10 13.5 14 25 2.3 2.7 350 25 0.1 1.8 400 14.5 50 3.2 500 40 10 2.5 nA V μA A mV mA μA V ● Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 3: The LT3475E and LT3475E-1 are guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LT3475I and LT3475I-1 are guaranteed to meet performance specifications over the –40°C to 125°C operating temperature range. Note 4: Current flows out of pin. Note 5: Current flows into pin. Note 6: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycles. 3475fb 3 LT3475/LT3475-1 TYPICAL PERFOR A CE CHARACTERISTICS LED Current vs VADJ 1.50 1.25 LED CURRENT (A) LED CURRENT (A) 1.00 0.75 0.50 0.25 0 0 0.25 0.5 0.75 VADJ (V) 1 1.25 3475 G01 TA = 25°C SWITCH ON VOLTAGE (mV) Switch Current Limit vs Duty Cycle 3.0 2.5 CURRENT LIMIT (A) 2.0 1.5 1.0 0.5 0 0 20 40 60 DUTY CYCLE (%) TA = 25°C 80 100 3475 G04 TYPICAL CURRENT LIMIT (A) CURRENT LIMIT (A) MINIMUM Oscillator Frequency vs Temperature 700 OSCILLATOR FREQUENCY (kHz) 650 600 550 500 450 400 –50 –25 RT = 24.3kΩ OSCILLATOR FREQUENCY (kHz) 700 600 500 400 300 200 100 0 50 25 75 0 TEMPERATURE (˚C) 100 125 OSCILLATOR FREQUENCY (kHz) 4 UW LED Current vs Temperature 1.2 VADJ = VREF • 2/3 1.0 0.8 0.6 0.4 0.2 0 –50 –25 VADJ = VREF • 7/30 600 500 400 300 200 100 0 50 25 75 0 TEMPERATURE (˚C) 100 125 Switch On Voltage TA = 25°C 0 1.5 0.5 1.0 SWITCH CURRENT (A) 2.0 3475 G03 3475 G02 Switch Current Limit vs Temperature 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 –50 –25 3.0 2.5 2.0 1.5 1.0 0.5 0 Current Limit vs Output Voltage TA = 25°C 50 25 75 0 TEMPERATURE (°C) 100 125 0 0.5 1.0 1.5 2.0 2.5 VOUT (V) 3.0 3.5 4.0 3475 G05 3475 G06 Oscillator Frequency Foldback TA = 25°C RT = 24.3kΩ 1000 Oscillator Frequency vs RT TA = 25°C 10 0 0.5 1.0 1.5 2.0 2.5 3475 G08 1 VOUT (V) 3475 G07 10 RT (kΩ) 100 3475 G09 3475fb LT3475/LT3475-1 TYPICAL PERFOR A CE CHARACTERISTICS Boost Pin Current 35 30 BOOST PIN CURRENT (mA) INPUT CURRENT (mA) 25 20 15 10 5 0 0 0.5 1.5 SWITCH CURRENT (A) 1.0 2.0 3475 G10 TA = 25°C 5 4 3 2 1 0 0 10 20 VIN (V) 3475 G11 OUTPUT VOLTAGE (V) Reference Voltage 1.28 1.27 1.26 VREF (V) 1.25 1.24 1.23 1.22 –50 –25 VIN (V) 6 5 4 3 2 1 0 VIN (V) 50 25 75 0 TEMPERATURE (˚C) PI FU CTIO S OUT1, OUT2 (Pins 1, 10): The OUT pin is the input to the current sense resistor. Connect this pin to the inductor and the output capacitor. LED1, LED2 (Pins 2, 9): The LED pin is the output of the current sense resistor. Connect the anode of the LED here. VIN (Pins 5, 6): The VIN pins supply current to the internal circuitry and to the internal power switches and must be locally bypassed. SW1, SW2 (Pins 4, 7): The SW pin is the output of the internal power switch. Connect this pin to the inductor, switching diode and boost capacitor. 3475fb UW 100 3475 G13 Quiescent Current 7 6 TA = 25°C 50 45 40 35 30 25 20 15 10 5 30 40 0 Open-Circuit Output Voltage and Input Current TA = 25°C 14 INPUT CURRENT LT3475-1 12 INPUT CURRENT (mA) 10 LT3475 8 LT3475-1 OUTPUT VOLTAGE LT3475 6 4 2 0 0 10 20 VIN (V) 30 40 3475 G12 Minimum Input Voltage, Single 1.5A White LED TA = 25°C TO START 9 TO RUN 10 Minimum Input Voltage, Two Series Connected 1.5A White LEDs TA = 25°C LED VOLTAGE 8 TO START LED VOLTAGE TO RUN 7 6 125 5 0 0.5 1 LED CURRENT (A) 1.5 3475 G14 0 0.5 1 LED CURRENT (A) 1.5 3475 G15 U U U BOOST1, BOOST2 (Pins 3, 8): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. GND (Pins 15, Exposed Pad Pin 21): Ground. Tie the GND pin and the exposed pad directly to the ground plane. The exposed pad metal of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The exposed pad must be soldered to the circuit board for proper operation. Use a large ground plane and thermal vias to optimize thermal performance. 5 LT3475/LT3475-1 PI FU CTIO S RT (Pin 14): The RT pin is used to set the internal oscillator frequency. Tie a 24.3k resistor from RT to GND for a 600kHz switching frequency. SHDN (Pin 16): The SHDN pin is used to shut down the switching regulator and the internal bias circuits. The 2.6V switching threshold can function as an accurate undervoltage lockout. Pull below 0.3V to shut down the LT3475/LT3475-1. Pull above 2.6V to enable the LT3475/ LT3475-1. Tie to VIN if the SHDN function is unused. REF (Pin 17): The REF pin is the buffered output of the internal reference. Either tie the REF pin to the VADJ pin for a 1.5A output current, or use a resistor divider to generate a lower voltage at the VADJ pin. Leave this pin unconnected if unused. VC1, VC2 (Pins 18, 13): The VC pin is the output of the internal error amp. The voltage on this pin controls the peak switch current. Use this pin to compensate the control loop. VADJ1, VADJ2 (Pins 19, 12): The VADJ pin is the input to the internal voltage-to-current amplifier. Connect the VADJ pin to the REF pin for a 1.5A output current. For lower output currents, program the VADJ pin using the following formula: ILED = 1.5A • VADJ/1.25V. PWM1, PWM2 (Pins 20, 11): The PWM pin controls the connection of the VC pin to the internal circuitry. When the PWM pin is low, the VC pin is disconnected from the internal circuitry and draws minimal current. If the PWM feature is unused, leave this pin unconnected. BLOCK DIAGRAM VIN CIN RT D1 BOOST1 C1 Q Q DRIVER D3 OUT1 COUT1 LED1 R S L1 SW1 Q1 0.067Ω 100Ω 2V 2V DLED1 gm1 1.25V PWM 1 Q3 VC1 1.25k 1.25k Q4 CC1 VADJ1 REF 6 + – + – U U U VIN SHDN RT VIN INT REG AND UVLO MASTER OSC BOOST2 D2 C1 ∑ SLOPE COMP MOSC 1 SLAVE OSC MOSC 2 SLOPE COMP ∑ C2 R S Q Q DRIVER D4 OUT2 100Ω 0.067Ω LED2 C2 SLAVE OSC Q2 SW2 L2 FREQUENCY FOLDBACK FREQUENCY FOLDBACK COUT2 gm2 DLED 2 PWM2 VC2 CC2 VADJ2 EXPOSED PAD GND 3475 BD 3475fb LT3475/LT3475-1 OPERATION The LT3475 is a dual constant frequency, current mode regulator with internal power switches capable of generating constant 1.5A outputs. Operation can be best understood by referring to the Block Diagram. If the SHDN pin is tied to ground, the LT3475 is shut down and draws minimal current from the input source tied to VIN. If the SHDN pin exceeds 1V, the internal bias circuits turn on, including the internal regulator, reference and oscillator. The switching regulators will only begin to operate when the SHDN pin exceeds 2.6V. The switcher is a current mode regulator. Instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. Compared to voltage mode control, current mode control improves loop dynamics and provides cycle-bycycle current limit. A pulse from the oscillator sets the RS flip-flop and turns on the internal NPN bipolar power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level determined by the voltage at VC, current comparator C1 resets the flip-flop, turning off the switch. The current in the inductor flows through the external Schottky diode and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way, the voltage on the VC pin controls the current through the inductor to the output. The internal error amplifier regulates the output current by continually adjusting the VC pin voltage. The threshold for switching on the VC pin is 0.8V, and an active clamp of 1.8V limits the output current. The voltage on the VADJ pin sets the current through the LED pin. The NPN, Q3, pulls a current proportional to the voltage on the VADJ pin through the 100Ω resistor. The gm amplifier servos the VC pin to set the current through the 0.067Ω resistor and the LED pin. When the voltage drop across the 0.067Ω resistor is equal to the voltage drop across the 100Ω resistor, the servo loop is balanced. Tying the REF pin to the VADJ pin sets the LED pin current to 1.5A. Tying a resistor divider to the REF pin allows the programming of LED pin currents of less than 1.5A. LED pin current can also be programmed by tying the VADJ pin directly to a voltage source. An LED can be dimmed with pulse width modulation using the PWM pin and an external NFET. If the PWM pin is unconnected or is pulled high, the part operates nominally. If the PWM pin is pulled low, the VC pin is disconnected from the internal circuitry and draws minimal current from the compensation capacitor. Circuitry drawing current from the OUT pin is also disabled. This way, the VC pin and the output capacitor store the state of the LED pin current until the PWM is pulled high again. This leads to a highly linear relationship between pulse width and output light, allowing for a large and accurate dimming range. The RT pin allows programming of the switching frequency. For applications requiring the smallest external components possible, a fast switching frequency can be used. If low dropout or very high input voltages are required, a slower switching frequency can be programmed. During startup VOUT will be at a low voltage. The NPN, Q3, can only operate correctly with sufficient voltage of ≈1.7V at VOUT, A comparator senses VOUT and forces the VC pin high until VOUT rises above 2V, and Q3 is operating correctly. The switching regulator performs frequency foldback during overload conditions. An amplifier senses when VOUT is less than 2V and begins decreasing the oscillator frequency down from full frequency to 15% of the nominal frequency when VOUT = 0V. The OUT pin is less than 2V during startup, short circuit, and overload conditions. Frequency foldback helps limit switch current under these conditions. The switch driver operates either from VIN or from the BOOST pin. An external capacitor and Schottky diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to saturate the internal bipolar NPN power switch for efficient operation. 3475fb 7 LT3475/LT3475-1 APPLICATIONS INFORMATION Open Circuit Protection The LT3475 has internal open-circuit protection. If the LED is absent or is open circuit, the LT3475 clamps the voltage on the LED pin at 14V. The switching regulator then operates at a very low frequency to limit the input current. The LT3475-1 has no internal open circuit protection. With the LT3475-1, be careful not to violate the ABSMAX voltage of th BOOST pin; if VIN > 25V, external open circuit protection circuitry (as shown in Figure 1) may be necessary.The output voltage during an open LED condition is shown in the Typical Performance Characteristics section. Undervoltage Lockout Undervoltage lockout (UVLO) is typically used in situations where the input supply is current limited, or has high source resistance. A switching regulator draws constant power from the source, so the source current increases as the source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where these problems might occur. An internal comparator will force the part into shutdown when VIN falls below 3.7V. If an adjustable UVLO threshold is required, the SHDN pin can be used. The threshold voltage of the SHDN pin comparator is 2.6V. An internal resistor pulls 9μA to ground from the SHDN pin at the UVLO threshold. Choose resistors according to the following formula: R2 = 2.6V VTH – 2.6V – 9μ A R1 OUT 10k 22V VC 100k 3475 F01 Figure 1. External Overvoltage Protection Circuitry for the LT3475-1 LT3475 VIN R1 SHDN 9μA C1 R2 GND 3475 F02 VIN 2.6V VC Figure 2. Undervoltage Lockout Keep the connections from the resistors to the SHDN pin short and make sure the coupling to the SW and BOOST pins is minimized. If high resistance values are used, the SHDN pin should be bypassed with a 1nF capacitor to prevent coupling problems from switching nodes. Setting the Switching Frequency The LT3475 uses a constant frequency architecture that can be programmed over a 200kHz to 2MHz range with a single external timing resistor from the RT pin to ground. A graph for selecting the value of RT for a given operating frequency is shown in the Typical Applications section. Table 1. Switching Frequencies SWITCHING FREQUENCY (MHz) RT (kΩ) 4.32 6.81 9.09 11.8 16.9 24.3 40.2 57.6 100 3475fb VTH = UVLO Threshold Example: Switching should not start until the input is above 8V. VTH = 8V R1=100k 2.6V R2 = = 57.6k 8V – 2.6V – 9μ A 100k 2 1.5 1.2 1 0.8 0.6 0.4 0.3 0.2 8 LT3475/LT3475-1 APPLICATIONS INFORMATION Table 1 shows suggested RT selections for a variety of switching frequencies. Operating Frequency Selection The choice of operating frequency is determined by several factors. There is a tradeoff between efficiency and component size. A higher switching frequency allows the use of smaller inductors at the cost of increased switching losses and decreased efficiency. Another consideration is the maximum duty cycle. In certain applications, the converter needs to operate at a high duty cycle in order to work at the lowest input voltage possible. The LT3475 has a fixed oscillator off time and a variable on time. As a result, the maximum duty cycle increases as the switching frequency is decreased. Input Voltage Range The minimum operating voltage is determined either by the LT3475’s undervoltage lockout of 4V, or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: ( VOUT + VF ) DC = ( VIN – VSW + VF ) where VF is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.4V at maximum load). This leads to a minimum input voltage of: V +V VIN(MIN ) = OUT F – VF + VSW DCMAX with DCMAX = 1–tOFF(MIN) • f where t0FF(MIN) is equal to 167ns and f is the switching frequency. Example: f = 600kHz, VOUT = 4V DCMAX = 1− 167ns • 600kHz = 0.90 4V + 0.4V VIN(MIN ) = – 0.4V + 0.4V = 4.9V 0.9 VSW 20V/DIV The maximum operating voltage is determined by the absolute maximum ratings of the VIN and BOOST pins, and by the minimum duty cycle. V +V VIN(MAX ) = OUT F – VF + VSW DCMIN with DCMIN = tON(MIN) • f where tON(MIN) is equal to 140ns and f is the switching frequency. Example: f = 750kHz, VOUT = 3.4V DCMIN = 140ns • 750kHz = 0.105 3.4V + 0.4V VIN(MAX ) = – 0.4V + 0.4V = 36V 0.105 The minimum duty cycle depends on the switching frequency. Running at a lower switching frequency might allow a higher maximum operating voltage. Note that this is a restriction on the operating input voltage; the circuit will tolerate transient inputs up to the Absolute Maximum Ratings of the VIN and BOOST pins. The input voltage should be limited to the VIN operating range (36V) during overload conditions (short circuit or start up). Minimum On Time The LT3475 will regulate the output current at input voltages greater than VIN(MAX). For example, an application with an output voltage of 3V and switching frequency of 1.2MHz has a VIN(MAX) of 20V, as shown in Figure 3. Figure 4 shows operation at 35V. Output ripple and peak inductor VOUT 500mV/DIV (AC COUPLED) IL 1A/DIV 3475 F03 Figure 3. Operation at VIN(MAX) = 20V. VOUT = 3V and fSW = 1.2MHHz 3475fb 9 LT3475/LT3475-1 APPLICATIONS INFORMATION current have significantly increased. Exceeding VIN(MAX) is safe if the external components have adequate ratings to handle the peak conditions and if the peak inductor current does not exceed 3.2A. A saturating inductor may further reduce performance. Table 2. Inductors PART NUMBER Sumida CR43-3R3 CR43-4R7 CDRH4D16-3R3 VOUT 500mV/DIV (AC COUPLED) IL 1A/DIV VALUE (μH) 3.3 4.7 3.3 3.3 4.7 5.0 5.6 10 15 10 15 15 3.3 4.7 3.3 4.7 3.3 10 10 15 IRMS (A) 1.44 1.15 1.10 1.57 1.32 2.20 2.0 1.30 1.10 1.68 1.33 3.1 1.30 1.10 2.00 1.50 1.80 1.50 3.9 3.1 DCR () 0.086 0.109 0.063 0.049 0.072 0.032 0.036 0.048 0.076 0.072 0.130 0.050 0.100 0.120 0.080 0.090 0.046 0.050 0.038 0.046 HEIGHT (mm) 3.5 3.5 1.8 3.0 3.0 2.8 2.8 3.0 3.0 3.4 3.4 4.0 2.0 2.0 2.9 2.9 2.0 2.0 5.2 5.2 CDRH4D28-3R3 CDRH4D28-4R7 CDRH6D26-5R0 CDRH6D26-5R6 CDRH5D28-100 CDRH5D28-150 3475 F04 VSW 20V/DIV CDRH73-100 CDRH73-150 CDRH104R-150 Coilcraft DO1606T-332 DO1606T-472 DO1608C-332 DO1608C-472 MOS6020-332 MOS6020-472 DO3316P-103 DO3316P-153 Figure 4. Operation above VIN(MAX). Output Ripple and Peak Inductor Current Increases Inductor Selection and Maximum Output Current A good first choice for the inductor value is: L = (VOUT + VF ) • 1.2MHz f where VF is the voltage drop of the catch diode (~0.4V), f is the switching frequency and L is in μH. With this value the maximum load current will be above 1.6A at all duty cycles. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For highest efficiency, the series resistance (DCR) should be less than 0.15Ω. Table 2 lists several vendors and types that are suitable. For robust operation at full load and high input voltages (VIN > 30V), use an inductor with a saturation current higher than 3.2A. The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a higher maximum load current, and reduces the output voltage ripple. If your load is lower than the maximum load current, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology’s Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), a minimum inductance is required to avoid sub-harmonic oscillations: L MIN = (VOUT + VF ) • 800kHz f 3475fb 10 LT3475/LT3475-1 APPLICATIONS INFORMATION The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to-peak inductor ripple current. The LT3475 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3475 will deliver depends on the switch current limit, the inductor value, and the input and output voltages. When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor Δ IL = Input Capacitor Selection Bypass the input of the LT3475 circuit with a 4.7μF or higher ceramic capacitor of X7R or X5R type. A lower value or a less expensive Y5V type will work if there is additional bypassing provided by bulk electrolytic capacitors or if the input source impedance is low. The following paragraphs describe the input capacitor considerations in more detail. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3475 input and to force this switching current into a tight local loop, minimizing EMI. The input capacitor must have low impedance at the switching frequency to do this effectively, and it must have an adequate ripple current rating. With two switchers operating at the same frequency but with different phases and duty cycles, calculating the input capacitor RMS current is not simple. However, a conservative value is the RMS input current for the channel that is delivering most power (VOUT • IOUT): CINRMS = IOUT • VOUT (VIN – VOUT ) IOUT < VIN 2 (1– DC)( VOUT + VF ) (L • f ) where f is the switching frequency of the LT3475 and L is the value of the inductor. The peak inductor and switch current is ΔI ISW (PK ) = IL (PK ) = IOUT + L 2 To maintain output regulation, this peak current must be less than the LT3475’s switch current limit ILIM. ILIM is at least 2.3A at low duty cycles and decreases linearly to 1.8A at DC = 0.9. The maximum output current is a function of the chosen inductor value: IOUT (MAX ) = ILIM – Δ IL 2 Δ IL 2 = 2.3A• (1–0.25•DC) – Choosing an inductor value so that the ripple current is small will allow a maximum output current near the switch current limit. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then use these equations to check that the LT3475 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT is less than ΔIL/2. and is largest when VIN = 2VOUT (50% duty cycle). As the second, lower power channel draws input current, the input capacitor’s RMS current actually decreases as the out-of-phase current cancels the current drawn by the higher power channel. Considering that the maximum load current from a single channel is ~1.5A, RMS ripple current will always be less than 0.75A. The high frequency of the LT3475 reduces the energy storage requirements of the input capacitor, so that the capacitance required is less than 10μF The combination . of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes them the preferred choice. The low ESR results in very low voltage ripple. Ceramic capacitors can handle larger magnitudes of ripple current than other capacitor types of the same value. Use X5R and X7R types. 3475fb 11 LT3475/LT3475-1 APPLICATIONS INFORMATION An alternative to a high value ceramic capacitor is a lower value ceramic along with a larger electrolytic capacitor. The electrolytic capacitor likely needs to be greater than 10μF in order to meet the ESR and ripple current requirements. The input capacitor is likely to see high surge currents when the input source is applied. Tantalum capacitors can fail due to an over-surge of current. Only use tantalum capacitors with the appropriate surge current rating. The manufacturer may also recommend operation below the rated voltage of the capacitor. A final caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plugging the circuit into a live power source) this tank can ring, doubling the input voltage and damaging the LT3475. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor in parallel with the ceramic capacitor. For details, see Application Note 88. Output Capacitor Selection For most LEDs, a 2.2μF 6.3V ceramic capacitor (X5R or , X7R) at the output results in very low output voltage ripple and good transient response. Other types and values will also work. The following discusses tradeoffs in output ripple and transient performance. The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilizes the LT3475’s control loop. Because the LT3475 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option. You can estimate output ripple with the following equation: VRIPPLE = ΔIL / (8 • f • COUT) for ceramic capacitors where ΔIL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula: IC(RMS) = Δ IL / 12 The low ESR and small size of ceramic capacitors make them the preferred type for LT3475 applications. Not all ceramic capacitors are the same, however. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT, this loss may be unacceptable. Use X7R and X5R types. Table 3 lists several capacitor vendors. Table 3. Low ESR Surface Mount Capacitors. VENDOR Taiyo-Yuden AVX TDK TYPE Ceramic Ceramic Ceramic SERIES X5R, X7R X5R, X7R X5R, X7R Diode Selection The catch diode (D3 from the Block Diagram) conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID(AVG) = IOUT (VIN – VOUT)/VIN The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to one half the typical peak switch current limit. Peak reverse voltage is equal to the regulator input voltage. Use a diode with a reverse voltage rating greater than the input voltage. Table 4 lists several Schottky diodes and their manufacturers. Diode reverse leakage can discharge the output capacitor during LED off times while PWM dimming. If operating at high ambient temperatures, use a low leakage Schottky for the widest PWM dimming range. 3475fb 12 LT3475/LT3475-1 APPLICATIONS INFORMATION Table 4. Schottky Diodes VR (V) On Semiconductor MBR0540 MBRM120E MBRM140 Diodes Inc B120 B130 B140HB DFLS140 B240 International Rectifier 10BQ030 30 1 420 20 30 40 40 40 1 1 1 1.1 2 500 500 530 510 500 40 20 40 0.5 1 1 620 530 550 IAVE(A) (A) VF at 1A (mV) VF at 2A (mV) BOOST Pin Considerations The capacitor and diode tied to the BOOST pin generate a voltage that is higher than the input voltage. In most cases, a 0.22μF capacitor and fast switching diode (such as the CMDSH-3 or MMSD914LT1) will work well. Figure 5 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.5V above the SW D2 pin for full efficiency. For outputs of 3.3V and higher, the standard circuit (Figure 5a) is best. For outputs between 2.8V and 3.3V, use a small Schottky diode (such as the BAT-54). For lower output voltages, the boost diode can be tied to the input (Figure 5b). The circuit in Figure 5a is more efficient because the BOOST pin current comes from a lower voltage source. The anode of the boost diode can be tied to another source that is at least 3V. For example, if you are generating a 3.3V output, and the 3.3V output is on whenever the LED is on, the BOOST pin can be connected to the 3.3V output. For LT3475-1 applications with higher output voltages, an additional Zener diode may be necessary (Figure 5d) to maintain pin voltage below the absolute maximum. In any case, be sure that the maximum voltage at the BOOST pin is both less than 60V and the voltage difference between the BOOST and SW pins is less than 30V. The minimum operating voltage of an LT3475 application is limited by the undervoltage lockout (~3.7V) and by the maximum duty cycle. The boost circuit also limits the minimum input voltage for proper start up. If the input voltage ramps slowly, or the LT3475 turns on when the output is already in regulation, the boost capacitor may not be fully charged. Because the boost capacitor charges D2 BOOST LT3475 VIN VIN GND VBOOST – VSW ≅ VOUT MAX VBOOST ≅ VIN + VOUT SW C3 VOUT VIN VIN BOOST LT3475 SW GND VBOOST – VSW ≅ VIN MAX VBOOST ≅ 2VIN C3 VOUT (5a) D2 VIN2 > 3V BOOST LT3475 VIN VIN GND 3475 F05  (5b) D2 C3 SW VOUT VIN VIN BOOST LT3475 SW GND VBOOST – VSW – VZ MAX VBOOST ≅ VIN + VOUT – VZ C3 VOUT VBOOST – VSW ≅ VIN2 MAX VBOOST ≅ VIN2 + VIN MINIMUM VALUE FOR VIN2 = 3V 3475 F05  (5c) (5d) Figure 5. Generating the Boost Voltage 3475fb 13 LT3475/LT3475-1 APPLICATIONS INFORMATION with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load current generally goes to zero once the circuit has started. The typical performance characteristics section shows a plot of minimum load to start and to run as a function of input voltage. Even without an output load current, in many cases the discharged output capacitor will present a load to the switcher that will allow it to start. The plots show the worst case, where VIN is ramping very slowly. Programming LED Current The LED current can be set by adjusting the voltage on the VADJ pin. For a 1.5A LED current, either tie VADJ to REF or to a 1.25V source. For lower output currents, program the VADJ using the following formula: ILED = 1.5A • VADJ/1.25V. Voltages less than 1.25V can be generated with a voltage divider from the REF pin, as shown in Figure 6. In order to have accurate LED current, precision resistors are preferred (1% or better is recommended). Note that the VADJ pin sources a small amount of bias current, so use the following formula to choose resistors: R2 = VADJ 1.25V – VADJ + 50nA R1 the voltage on the VADJ pin by tying a low on resistance FET to the resistor divider string. This allows the selection of two different LED currents. For reliable operation program an LED current of no less than 50mA. The maximum current dimming ratio (IRATIO) can be calculated from the maximum LED current (IMAX) and the minimum LED current (IMIN) as follows: IMAX/IMIN = IRATIO Another dimming control circuit (Figure 8) uses the PWM pin and an external NFET tied to the cathode of the LED. An external PWM signal is applied to the PWM pin and the gate of the NFET (For PWM dimming ratios of 20 to 1 or less, the NFET can be omitted). The average LED current is proportional to the duty cycle of the PWM signal. When the PWM signal goes low, the NFET turns off, turning off the LED and leaving the output capacitor charged. The PWM pin is pulled low as well, which disconnects the VC pin, storing the voltage in the capacitor tied there. Use the C-RC string shown in Figure 8 and Figure 9 tied to the VC pin for proper operation during startup. When the PWM pin goes high again, the LED current returns rapidly to its previous on state since the compensation and output capacitors are at the correct voltage. This fast settling time allows the REF R1 VADJ R2 DIM GND 3475 F07 LT3475 To minimize the error from variations in VADJ pin current, use resistors with a parallel resistance of less than 4k. Use resistor strings with a high enough series resistance so as not to exceed the 500μA current compliance of the REF pin. Dimming Control There are several different types of dimming control circuits. One dimming control circuit (Figure 7) changes REF Figure 7. Dimming with a MOSFET and Resistor Divider PWM 100Hz TO 10kHz PWM LT3475 LED VC 10k 3.3nF 0.1μF GND 3475 F08 R1 VADJ R2 LT3475 GND 3475 F06 Figure 6. Setting VADJ with a Resistor Divider Figure 8. Dimming Using PWM Signal 3475fb 14 LT3475/LT3475-1 APPLICATIONS INFORMATION LT3475 to maintain diode current regulation with PWM pulse widths as short as 7.5 switching cycles (12.5μs for fSW = 600kHz). Maximum PWM period is determined by the system and is unlikely to be longer than 12ms. Using PWM periods shorter than 100μs is not recommended. The maximum PWM dimming ratio (PWMRATIO) can be calculated from the maximum PWM period (tMAX) and minimum PWM pulse width (tMIN) as follows: tMAX/tMIN = PWMRATIO Total dimming ratio (DIMRATIO) is the product of the PWM dimming ratio and the current dimming ratio. Example: IMAX = 1A, IMIN = 0.1A, tMAX = 9.9ms tMIN = 3.3μs (fSW = 1.4MHz) IRATIO = 1A/0.1A =10:1 PWMRATIO = 9.9ms/3.3μs = 3000:1 DIMRATIO = 10 • 3000 = 30000:1 To achieve the maximum PWM dimming ratio, use the circuit shown in Figure 9. This allows PWM pulse widths as short as 4.5 switching cycles (7.5μs for fSW = 600kHz). Note that if you use the circuit in Figure 9, the rising edge of the two PWM signals must align within 100ns. 220pF RT 1M PWM1 RT LT3475 GND 3475 F09 Layout Hints As with all switching regulators, careful attention must be paid to the PCB layout and component placement. To maximize efficiency, switch rise and fall times are made as short as possible. To prevent electromagnetic interference (EMI) problems, proper layout of the high frequency switching path is essential. The voltage signal of the SW and BOOST pins have sharp rise and fall edges. Minimize the area of all traces connected to the BOOST and SW pins and always use a ground plane under the switching regulator to minimize interplane coupling. In addition, the ground connection for frequency setting resistor RT and capacitors at VC1, VC2 pins (refer to the Block Diagram) should be tied directly to the GND pin and not shared with the power ground path, ensuring a clean, noise-free connection. PWM1 SHDN PWM2 20 19 18 17 16 15 14 13 12 9 VIN VC 10k 3.3nF 0.1μF 10 1 2 3 4 5 6 7 8 11 3475 F10 VIA TO LOCAL GND PLANE Figure 9. Extending the PWM Dimming Range Figure 10. Recommended Component Placement 3475fb 15 LT3475/LT3475-1 TYPICAL APPLICATIONS Dual Step-Down 1A LED Driver VIN 5V TO 36V D3 C1 4.7μF 50V VIN BOOST1 SHDN BOOST2 LT3475 C3 0.22μF 6.3V SW2 D2 D4 L2 10μH C4 0.22μF 6.3V SW1 D1 L1 10μH C5 2.2μF 6.3V LED 1 OUT1 LED1 VC1 C6 0.1μF R2 1k R3 2k REF VADJ1 GND OUT2 LED2 LED 2 VC2 RT VADJ2 R1 24.3k 3475 TA02 C2 2.2μF 6.3V C7 0.1μF C1 TO C5: X5R OR X7R D1, D2: DFLS140 D3, D4: MBR0540 LED CURRENT = 1A fSW = 600kHz Dual Step-Down 1.5A LED Driver with 1200 : 1 True Color PWM Dimming VIN 6V TO 36V D3 C1 4.7μF 50V VIN BOOST1 SHDN BOOST2 LT3475 C3 0.22μF 6.3V SW2 D2 L1 10μH D4 L2 10μH C4 2.2μF 6.3V C2 0.22μF 6.3V SW1 D1 OUT1 LED1 C5 2.2μF 6.3V OUT2 LED2 LED 2 PWM2 VC2 RT VADJ2 GND M2 C9 0.1μF C8 220p 1M R2 R1 24.3k C7 3.3nF R4 10k 1.5A LED CURRENT 1.5A LED CURRENT LED 1 PWM1 VC1 R3 10k M1 C8 0.1μF C6 3.3nF REF VADJ1 M3 fSW = 600kHz PWM1 D1, D2: B260 D3, D4: MBR0540 C1 TO C5: X5R OR X7R M1, M2: Si2302ADS M3: 2n7002L 3475 TA03 PWM2 3475fb 16 LT3475/LT3475-1 TYPICAL APPLICATIONS Step-Down 3A LED Driver VIN 5V TO 36V D3 C1 4.7μF 50V VIN BOOST1 SHDN BOOST2 LT3475 C3 0.22μF 6.3V SW2 D2 L1 10μH D4 L2 10μH C2 0.22μF 6.3V SW1 D1 OUT1 C5 2.2μF 6.3V OUT2 LED1 LED2 C4 2.2μF 6.3V VC1 C6 0.1μF REF VADJ1 GND VC2 RT VADJ2 R1 24.3k C7 0.1μF LED 1 3A LED CURRENT D1, D2: B240A D3, D4: MBR0540 C1 TO C5: X5R OR X7R fSW = 600kHz 3475 TA04 Dual Step-Down LED Driver with Series Connected LEDs VIN 10V TO 36V D3 C1 4.7μF 50V VIN BOOST1 SHDN BOOST2 LT3475 C3 0.22μF 10V SW2 D2 L1 15μH D4 L2 15μH C2 0.22μF 10V SW1 D1 OUT1 OUT2 LED2 C5 2.2μF 10V C4 2.2μF 10V LED1 LED 1 1.5A LED CURRENT C6 0.1μF LED 3 VC1 REF VADJ1 GND VC2 RT VADJ2 R1 24.3k LED 2 C7 0.1μF LED 4 1.5A LED CURRENT D1, D2: B240A D3, D4: MMSD4148T1 C1 TO C5: X5R OR X7R fSW = 600kHz 3475 TA05 3475fb 17 LT3475/LT3475-1 TYPICAL APPLICATIONS Dual Step-Down 1.5A Red LED Driver VIN 5V TO 28V C1 4.7μF 35V D3 VIN BOOST1 SHDN BOOST2 LT3475 C3 0.22μF 35V SW2 D2 D4 L2 10μH C2 0.22μF 35V SW1 D1 OUT1 L1 10μH OUT2 LED2 C5 2.2μF 6.3V C4 2.2μF 6.3V LED1 VC1 C6 0.1μF 1.5A LED CURRENT LED 1 REF VADJ1 GND VC2 RT VADJ2 R1 24.3k C7 0.1μF LED 2 1.5A LED CURRENT D1, D2: B240A D3, D4: MMSD4148T1 C1 TO C5: X5R OR X7R fSW = 600kHz 3475 TA06 3475fb 18 LT3475/LT3475-1 PACKAGE DESCRIPTION FE Package 20-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663) Exposed Pad Variation CB 6.40 – 6.60* (.252 – .260) 3.86 (.152) 20 1918 17 16 15 14 13 12 11 3.86 (.152) 6.60 ± 0.10 4.50 ± 0.10 SEE NOTE 4 2.74 (.108) 0.45 ± 0.05 1.05 ± 0.10 0.65 BSC 6.40 2.74 (.252) (.108) BSC RECOMMENDED SOLDER PAD LAYOUT 1 2 3 4 5 6 7 8 9 10 1.20 (.047) MAX 0° – 8° 4.30 – 4.50* (.169 – .177) 0.25 REF 0.09 – 0.20 (.0035 – .0079) 0.50 – 0.75 (.020 – .030) 0.65 (.0256) BSC NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE20 (CB) TSSOP 0204 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3475fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LT3475/LT3475-1 TYPICAL APPLICATION Dual Step-Down 1.5A LED Driver with Four Series Connected LED Output VIN 21V TO 36V D1 L1 33μH R1 1k D2 C1 4.7μF 50V C2 0.22μF 16V D5 OUT1 LED1 R4 10k D7 C4 2.2μF 25V R6 100k 12V TO 18V LED VOLTAGE VC1 C6 0.1μF REF VADJ1 Q1 1.5A LED CURRENT* GND VC2 RT VADJ2 R3 24.3k C7 0.1μF C5 2.2μF 25V R7 100k D8 OUT2 LED2 12V TO 18V LED VOLTAGE R5 10k VIN BOOST1 SHDN BOOST2 LT3475-1 C3 0.22μF 16V D6 D3 L2 33μH R2 1k D4 SW1 SW2 1.5A LED CURRENT* Q2 fSW = 600kHz 3475 TA08 D1, D4: 7.5V ZENER DIODE D2, D3: MMSD4148 D5, D6: B240A D7, D8: 22V ZENER DIODE R1, R2: USE 0.5W RESISTOR OF TWO 2k 0.25W RESISTORS IN PARALLEL Q1, Q2: MMBT3904 C1 TO C5: X5R or X7R *DERATE LED CURRENT AT ELEVATED AMBIENT TEMPERATURES TO MAINTAIN LT3475-1 JUNCTION TEMPERATURE BELOW 125 °C. RELATED PARTS PART NUMBER LT1618 LT3466 LT3474 LT3477 LT3479 LT3846 DESCRIPTION Constant-Current, 1.4MHz, 1.5A Boost Converter Dual Full Function Step-Up LED Driver 36V, 1A (ILED), 2MHz Step-Down LED Driver 42V, 3A, 3.5MHz Boost, Buck-Boost, Buck LED Driver COMMENTS VIN(MIN) = 1.6V, VIN(MAX) = 18V, VOUT(MAX) = 35V, Analog/PWM, ISD < 1μA, MS10 Package Drivers Up to 20 LEDs, VIN: 2.7V to 24V, VOUT(MAX) = 40V, DFN, TSSOP16E Packages VIN(MIN) = 4V, VIN(MAX) = 36V, 400:1 True Color PWM, ISD < 1μA, TSSOP16E Package VIN(MIN) = 2.5V, VIN(MAX) = 25V, VOUT(MAX) = 40V, Analog/PWM, ISD < 1μA, QFN, TSSOP20E Packages 3A, Full-Featured DC/DC Converter with VIN(MIN) = 2.5V, VIN(MAX) = 24V, VOUT(MAX) = 40V, Analog/PWM, ISD < 1μA, Soft-Start and Inrush Current Protection DFN, TSSOP Packages Dual 1.3A, 2MHz, LED Driver VIN: 2.5V to 24V, VOUT(MAX) = 36V, 1000:1 True Color PWMTM Dimmin, DFN, TSSOP16E Packages 3475fb 20 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● LT 1007 REV B • PRINTED IN USA www.linear.com © LINEAR TECHNOLOGY CORPORATION 2006