LT3574IMS

LT3574 Isolated Flyback Converter Without an Opto-Coupler Features n n n n Description The LT®3574 is a monolithic switching regulator specifically designed for the isolated flyback topology. No third winding or opto-isolator is required for regulation. The part senses the isolated output voltage directly from the primary side flyback waveform. A 0.65A, 60V NPN power switch is integrated along with all control logic into a 16-lead MSOP package. The LT3574 operates with input supply voltages from 3V to 40V, and can deliver output power up to 3W with no external power switch. The LT3574 utilizes boundary mode operation to provide a small magnetic solution with improved load regulation. The output voltage is easily set with two external resistors and the transformer turns ratio. Off-the-shelf transformers are available for many applications. L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks and No RSENSE and ThinSOT are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. n n n n n n 3V to 40V Input Voltage Range 0.65A, 60V Integrated NPN Power Switch Boundary Mode Operation No Transformer Third Winding or Opto-Isolator Required for Regulation Improved Primary-Side Winding Feedback Load Regulation VOUT Set with Two External Resistors BIAS Pin for Internal Bias Supply and Power NPN Driver Programmable Soft-Start Programmable Power Switch Current Limit 16-Lead MSOP Package applications n Industrial, Automotive and Medical Isolated Power Supplies typical application 5V Isolated Flyback Converter VIN 12V TO 24V 10µF 357k SHDN/UVLO 51.1k RFB RREF 6.04k SW VC 28.7k 10k 10nF 59k 1nF 3574 TA01 B340A VIN 0.22µF 2k 50µH PMEG6010 80.6k 3:1 • • OUTPUT VOLTAGE ERROR (%) 5.6µH VOUT+ 5V 0.35A 22µF VOUT– Load Regulation 1.0 VIN = 24V VIN = 12V 0 0.5 LT3574 TC RILIM SS GND –0.5 TEST BIAS –1.0 0 100 200 300 400 IOUT (mA) 500 600 700 4.7µF 3574 TA01b 3574f  LT3574 absolute MaxiMuM ratings SW ............................................................................60V VIN, SHDN/UVLO, RFB, BIAS .....................................40V SS, VC, TC, RREF , RILIM ...............................................5V Maximum Junction Temperature .......................... 125°C Operating Junction Temperature Range (Note 2).................................................. –40°C to 125°C Storage Temperature Range .................. –65°C to 150°C pin conFiguration TOP VIEW GND TEST GND SW VIN BIAS SHDN/UVLO GND 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 GND TC RREF RFB VC RILIM SS GND MS PACKAGE 16-LEAD PLASTIC MSOP TJMAX = 125°C, θJA = 120°C/W, θJC = 21°C/W orDer inForMation LEAD FREE FINISH LT3574EMS#PBF LT3574IMS#PBF TAPE AND REEL LT3574EMS#TRPBF LT3574IMS#TRPBF PART MARKING* 3574 3574 PACKAGE DESCRIPTION 16-Lead Plastic MSOP 16-Lead Plastic MSOP TEMPERATURE RANGE –40°C to 125°C –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ electrical characteristics PARAMETER Input Voltage Range Quiescent Current Soft-Start Current SHDN/UVLO Pin Threshold SHDN/UVLO Pin Hysteresis Current Soft-Start Threshold Maximum Switching Frequency Switch Current Limit Minimum Current Limit Switch VCESAT RREF Voltage RREF Voltage Line Regulation RREF Pin Bias Current IREF Reference Current Error Amplifier Voltage Gain The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, unless otherwise noted. CONDITIONS l MIN 3 TYP 3.5 0 7 MAX 40 1 1.29 3 UNITS V mA µA µA V µA V kHz SS = 0V VSHDN/UVLO = 0V SS = 0.4V UVLO Pin Voltage Rising VUVLO = 1V l 1.15 2 1.22 2.5 0.7 1000 RILIM = 10k VC = 0V ISW = 0.5A VIN = 3V l 0.65 0.9 100 150 1.1 250 1.25 1.25 0.03 600 A mA mV V %/ V nA µA V/V 1.21 1.20 1.23 0.01 3V < VIN < 40V (Note 3) Measured at RFB Pin with RREF = 6.49k VIN = 3V l 100 190 150 3574f  LT3574 electrical characteristics PARAMETER Error Amplifier Transconductance Minimum Switching Frequency TC Current into RREF BIAS Pin Voltage The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, unless otherwise noted. CONDITIONS DI = 10µA, VIN = 3V VC = 0.35V RTC = 20.1k IBIAS = 30mA 2.9 MIN TYP 150 40 27.5 3 3.1 MAX UNITS µmhos kHz µA V Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3574E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design characterization and correlation with statistical process controls. The LT3574I is guaranteed over the full –40°C to 125°C operating junction temperature range. Note 3: Current flows out of the RREF pin. typical perForMance characteristics Output Voltage 5.20 5.15 5.10 VOUT (V) 5.05 5.00 4.95 4.90 4.85 4.80 –50 –25 50 0 25 75 TEMPERATURE (°C) 100 125 QUIESCENT CURRENT (mA) 6 5 4 3 2 1 0 –50 –25 VIN = 40V BIAS = 20V TA = 25°C, unless otherwise noted. Bias Pin Voltage 3.2 3.0 BIAS VOLTAGE (V) 2.8 2.6 2.4 2.2 2.0 –50 VIN = 40V VIN = 12V Quiescent Current VIN = 5V BIAS = 5V 50 25 75 0 TEMPERATURE (°C) 100 125 –25 50 25 0 75 TEMPERATURE (°C) 100 125 3574 G01 3574 G02 3574 G03 3574f  LT3574 typical perForMance characteristics Switch Saturation Voltage 300 250 200 150 125°C 100 50 0 25°C –50°C 1200 1000 CURRENT LIMIT (mA) 800 600 400 200 0 –50 –25 MINIMUM CURRENT LIMIT 50 25 75 0 TEMPERATURE (°C) 100 125 MAXIMUM CURRENT LIMIT TA = 25°C, unless otherwise noted. Switch Current Limit vs RILIM Switch Current Limit RILIM = 10k SWITCH CURRENT LIMIT (A) 1.2 1.0 0.8 0.6 0.4 0.2 0 SWITCH VCESAT VOLTAGE (mV) 0 100 200 300 400 500 600 700 800 900 SWITCH CURRENT (mA) 3574 G04 1 10 20 30 40 RILIM RESISTANCE (k ) 50 3574 G06 3574 G05 SHDN/UVLO Falling Threshold 1.28 12 10 SS PIN CURRENT (µA) 3574 G07 SS Pin Current SHDN/UVLO VOLTAGE (V) 1.26 8 6 4 2 0 –60 –40 –20 0 20 40 60 80 100 120 140 TEMPERATURE (°C) 3574 G08 1.24 1.22 1.20 1.18 –60 –40 –20 0 20 40 60 80 100 120 140 TEMPERATURE (°C) 3574f  LT3574 pin Functions BIAS: Bias Voltage. This pin supplies current to the switch driver and internal circuitry of the LT3574. This pin must be locally bypassed with a capacitor. This pin may also be connected to VIN if a third winding is not used and if VIN ≤ 15V. If a third winding is used, the BIAS voltage should be lower than the input voltage for proper operation. GND: Ground. RFB: Input Pin for External Feedback Resistor. This pin is connected to the transformer primary (VSW). The ratio of this resistor to the RREF resistor, times the internal bandgap reference, determines the output voltage (plus the effect of any non-unity transformer turns ratio). The average current through this resistor during the flyback period should be approximately 200µA. For nonisolated applications, this pin should be connected to VIN. RILIM: Maximum Current Limit Adjust Pin. A resistor should be tied to this pin to ground to set the current limit. Use a 10k resistor for the full current capabilities of the switch. RREF : Input Pin for External Ground-Referred Reference Resistor. This resistor should be in the range of 6k, but for convenience, need not be precisely this value. For nonisolated applications, a traditional resistor voltage divider may be connected to this pin. SHDN/UVLO: Shutdown/Undervoltage Lockout. A resistor divider connected to VIN is tied to this pin to program the minimum input voltage at which the LT3574 will operate. At a voltage below ~0.7V, the part draws no quiescent current. When below 1.22V and above ~0.7V, the part will draw 10µA of current, but internal circuitry will remain off. Above 1.22V, the internal circuitry will start and a 7µA current will be fed into the SS pin. When this pin falls below 1.22V, 2.5µA will be pulled from the pin to provide programmable hysteresis for UVLO. SS: Soft-Start Pin. Place a soft-start capacitor here to limit start-up inrush current and output voltage ramp rate. Switching starts when the voltage at this pin reaches ~0.7V. SW: Collector Node of the Output Switch. This pin has large currents flowing through it. Keep the traces to the switching components as short as possible to minimize electromagnetic radiation and voltage spikes. TC: Output Voltage Temperature Compensation. Connect a resistor to ground to produce a current proportional to absolute temperature to be sourced into the RREF node. ITC = 0.55V/RTC . TEST: This pin is used for testing purposes only and must be connected to ground for the part to operate properly. VC: Compensation Pin for Internal Error Amplifier. Connect a series RC from this pin to ground to compensate the switching regulator. A 100pF capacitor in parallel helps eliminate noise. VIN : Input Voltage. This pin supplies current to the internal start-up circuitry and as a reference voltage for the DCM comparator and feedback circuitry. This pin must be locally bypassed with a capacitor. 3574f  LT3574 block DiagraM VIN C1 R3 L1A T1 N:1 D1 VOUT + L1B C2 VOUT – SW FLYBACK ERROR AMP 1.23V ONE SHOT • • TC CURRENT TC R6 RREF R4 BIAS C5 R1 SHDN/UVLO VIN Q3 Q2 RFB I2 20µA – gm + CURRENT COMPARATOR A2 – + – A1 + S S R + VIN V1 120mV DRIVER BIAS Q – Q1 MASTER LATCH A4 1.22V + A5 – INTERNAL REFERENCE AND REGULATORS + – RSENSE 0.036 GND I1 7µA OSCILLATOR VC R7 C4 R2 Q4 SS C3 2.5µA RILIM 3574 BD R5 3574f  LT3574 operation The LT3574 is a current mode switching regulator IC designed specifically for the isolated flyback topology. The special problem normally encountered in such circuits is that information relating to the output voltage on the isolated secondary side of the transformer must be communicated to the primary side in order to maintain regulation. Historically, this has been done with opto-isolators or extra transformer windings. Opto-isolator circuits waste output power and the extra components increase the cost and physical size of the power supply. Optoisolators can also exhibit trouble due to limited dynamic response, nonlinearity, unit-to-unit variation and aging over life. Circuits employing extra transformer windings also exhibit deficiencies. Using an extra winding adds to the transformer’s physical size and cost, and dynamic response is often mediocre. The LT3574 derives its information about the isolated output voltage by examining the primary side flyback pulse waveform. In this manner, no opto-isolator nor extra transformer winding is required for regulation. The output voltage is easily programmed with two resistors. Since this IC operates in boundary control mode, the output voltage is calculated from the switch pin when the secondary current is almost zero. This method improves load regulation without external resistors and capacitors. The Block Diagram shows an overall view of the system. Many of the blocks are similar to those found in traditional switching regulators including: internal bias regulator, oscillator, logic, current amplifier and comparator, driver, and output switch. The novel sections include a special flyback error amplifier and a temperature compensation circuit. In addition, the logic system contains additional logic for boundary mode operation, and the sampling error amplifier. The LT3574 features a boundary mode control method, where the part operates at the boundary between continuous conduction mode and discontinuous conduction mode. The VC pin controls the current level just as it does in normal current mode operation, but instead of turning the switch on at the start of the oscillator period, the part detects when the secondary-side winding current is zero. Boundary Mode Operation Boundary mode is a variable frequency, current mode switching scheme. The switch turns on and the inductor current increases until a VC pin controlled current limit. The voltage on the SW pin rises to the output voltage divided by the secondary-to-primary transformer turns ratio plus the input voltage. When the secondary current through the diode falls to zero, the SW pin voltage falls below VIN . A discontinuous conduction mode (DCM) comparator detects this event and turns the switch back on. Boundary mode returns the secondary current to zero every cycle, so the parasitic resistive voltage drops do not cause load regulation errors. Boundary mode also allows the use of a smaller transformer compared to continuous conduction mode and no subharmonic oscillation. At low output currents the LT3574 delays turning on the switch, and thus operates in discontinuous mode. Unlike a traditional flyback converter, the switch has to turn on to update the output voltage information. Below 0.6V on the VC pin, the current comparator level decreases to its minimum value, and the internal oscillator frequency decreases in frequency. With the decrease of the internal oscillator, the part starts to operate in DCM. The output current is able to decrease while still allowing a minimum switch off-time for the error amp sampling circuitry. The typical minimum internal oscillator frequency with VC equal to 0V is 40kHz. 3574f  LT3574 applications inForMation ERROR AMPLIFIER—PSEUDO DC THEORY In the Block Diagram, the RREF (R4) and RFB (R3) resistors can be found. They are external resistors used to program the output voltage. The LT3574 operates much the same way as traditional current mode switchers, the major difference being a different type of error amplifier which derives its feedback information from the flyback pulse. Operation is as follows: when the output switch, Q1, turns off, its collector voltage rises above the VIN rail. The amplitude of this flyback pulse, i.e., the difference between it and VIN, is given as: VFLBK = (VOUT + VF + ISEC • ESR) • NPS VF = D1 forward voltage ISEC = Transformer secondary current ESR = Total impedance of secondary circuit NPS = Transformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current by the action of RFB and Q2. Nearly all of this current flows through resistor RREF to form a ground-referred voltage. This voltage is fed into the flyback error amplifier. The flyback error amplifier samples this output voltage information when the secondary side winding current is zero. The error amplifier uses a bandgap voltage, 1.23V, as the reference voltage. The relatively high gain in the overall loop will then cause the voltage at the RREF resistor to be nearly equal to the bandgap reference voltage VBG. The relationship between VFLBK and VBG may then be expressed as: V V a  FLBK  = BG  RFB  RREF VFLBK or, In combination with the previous VFLBK expression yields an expression for VOUT, in terms of the internal reference, programming resistors, transformer turns ratio and diode forward voltage drop:  R  1  VOUT = VBG  FB    − VF − ISEC (ES R)  RREF   a NPS  Additionally, it includes the effect of nonzero secondary output impedance (ESR). This term can be assumed to be zero in boundary control mode. More details will be discussed in the next section. Temperature Compensation The first term in the VOUT equation does not have a temperature dependence, but the diode forward drop has a significant negative temperature coefficient. To compensate for this, a positive temperature coefficient current source is connected to the RREF pin. The current is set by a resistor to ground connected to the TC pin. To cancel the temperature coefficient, the following equation is used: d VF R 1 = − FB • • dT R TC NPS −RFB 1 R TC = • NPS d VF / d T d VTC or, dT dV R • TC ≈ FB dT NPS (dVF / dT) = Diode’s forward voltage temperature coefficient (dVTC / dT) = 2mV VTC = 0.55V The resistor value given by this equation should also be verified experimentally, and adjusted if necessary to achieve optimal regulation over temperature. The revised output voltage is as follows:  R  1  VOUT = VBG  FB    − VF  RREF   NPS a  V  R −  TC  • FB – ISEC (ESR)  R TC  NPS a 3574f  R   1 = VBG  FB     RREF   a  a = Ratio of Q1 IC to IE, typically ≈ 0.986 VBG = Internal bandgap reference  LT3574 applications inForMation ERROR AMPLIFIER—DYNAMIC THEORY Due to the sampling nature of the feedback loop, there are several timing signals and other constraints that are required for proper LT3574 operation. Minimum Current Limit The LT3574 obtains output voltage information from the SW pin when the secondary winding conducts current. The sampling circuitry needs a minimum amount of time to sample the output voltage. To guarantee enough time, a minimum inductance value must be maintained. The primary side magnetizing inductance must be chosen above the following value: L PRI ≥ VOUT • t MIN  2µH • NPS = VOUT • NPS •  V  IMIN Selecting RFB and RREF Resistor Values The expression for VOUT, developed in the Operation section, can be rearranged to yield the following expression for RFB: RFB = where, VOUT = Output voltage VF = Switching diode forward voltage a = Ratio of Q1, IC to IE, typically 0.986 NPS = Effective primary-to-secondary turns ratio VTC = 0.55V The equation assumes the temperature coefficients of the diode and VTC are equal, which is a good first-order approximation. Strictly speaking, the above equation defines RFB not as an absolute value, but as a ratio of RREF. So, the next question is, “What is the proper value for RREF?” The answer is that RREF should be approximately 6.04k. The LT3574 is trimmed and specified using this value of RREF. If the impedance of RREF varies considerably from 6.04k, additional errors will result. However, a variation in RREF of several percent is acceptable. This yields a bit of freedom in selecting standard 1% resistor values to yield nominal RFB /RREF ratios. The RFB resistor given by this equation should also be verified experimentally, and adjusted if necessary for best output accuracy. Tables 1-4 are useful for selecting the resistor values for RREF and RFB with no equations. The tables provide RFB, RREF and RTC values for common output voltages and common winding ratios. Table 1. Common Resistor Values for 1:1 Transformers VOUT (V) 3.3 5 12 15 20 NPS 1.00 1.00 1.00 1.00 1.00 RFB (kΩ) 18.7 27.4 64.9 80.6 107 RREF (kΩ) 6.04 6.04 6.04 6.04 6.04 RTC (kΩ) 19.1 28 66.5 80.6 105 RREF • NPS ( VOUT + VF ) a + VTC    VBG tMIN = minimum off-time, 350ns IMIN = minimum current limit, 175mA The minimum current limit is higher than that on the Electrical Characteristics table due to the overshoot caused by the comparator delay. Leakage Inductance Blanking When the output switch first turns off, the flyback pulse appears. However, it takes a finite time until the transformer primary-side voltage waveform approximately represents the output voltage. This is partly due to the rise time on the SW node, but more importantly due to the transformer leakage inductance. The latter causes a very fast voltage spike on the primary side of the transformer that is not directly related to output voltage (some time is also required for internal settling of the feedback amplifier circuitry). The leakage inductance spike is largest when the power switch current is highest. In order to maintain immunity to these phenomena, a fixed delay is introduced between the switch turn-off command and the beginning of the sampling. The blanking is internally set to 150ns. In certain cases, the leakage inductance may not be settled by the end of the blanking period, but will not significantly affect output regulation. 3574f  LT3574 applications inForMation Table 2. Common Resistor Values for 2:1 Transformers VOUT (V) 3.3 5 12 15 NPS 2.00 2.00 2.00 2.00 RFB (kΩ) 37.4 56 130 162 RREF (kΩ) 6.04 6.04 6.04 6.04 RTC (kΩ) 18.7 28 66.5 80.6 Table 3. Common Resistor Values for 3:1 Transformers VOUT (V) 3.3 5 10 NPS 3.00 3.00 3.00 RFB (kΩ) 56.2 80.6 165 RREF (kΩ) 6.04 6.04 6.04 RTC (kΩ) 20 28.7 54.9 relatively constant maximum output current regardless of input voltage. This is due to the continuous nonswitching behavior of the two currents. A flyback converter has both discontinuous input and output currents which makes it similar to a nonisolated buck-boost. The duty cycle will affect the input and output currents, making it hard to predict output power. In addition, the winding ratio can be changed to multiply the output current at the expense of a higher switch voltage. The graphs in Figures 1-3 show the maximum output power possible for the output voltages 3.3V, 5V and 12V. The maximum power output curve is the calculated output power if the switch voltage is 50V during the off-time. To achieve this power level at a given input, a winding ratio value must be calculated to stress the switch to 50V, resulting in some odd ratio values. The curves below are examples of common winding ratio values and the amount of output power at given input voltages. One design example would be a 5V output converter with a minimum input voltage of 20V and a maximum input voltage of 30V. A three-to-one winding ratio fits this design example perfectly and outputs close to 2.5W at 30V but lowers to 2W at 20V. 3.5 3.0 OUTPUT POWER (W) 2.5 2.0 1.5 1.0 0.5 Table 4. Common Resistor Values for 4:1 Transformers VOUT (V) 3.3 5 NPS 4.00 4.00 RFB (kΩ) 76.8 113 RREF (kΩ) 6.04 6.04 RTC (kΩ) 19.1 28 Output Power A flyback converter has a complicated relationship between the input and output current compared to a buck or a boost. A boost has a relatively constant maximum input current regardless of input voltage and a buck has a 3.5 3.0 OUTPUT POWER (W) OUTPUT POWER (W) 2.5 2.0 1.5 1.0 0.5 0 0 5 10 15 20 25 30 35 INPUT VOLTAGE (V) MAX POWER OUTPUT 5:1 1:1 7:1 2:1 10:1 3:1 4:1 40 45 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 0 5 10 3574 F01 15 20 25 30 35 INPUT VOLTAGE (V) MAX POWER OUTPUT 4:1 1:1 5:1 2:1 7:1 3:1 40 45 0 0 5 10 3574 F02 15 20 25 30 35 INPUT VOLTAGE (V) MAX POWER OUTPUT 1:1 2:1 3:1 40 45 3574 F03 Figure 1. Output Power for 3.3V Output Figure 2. Output Power for 5V Output Figure 3. Output Power for 12V Output 3574f 0 LT3574 applications inForMation TRANSFORMER DESIGN CONSIDERATIONS Transformer specification and design is perhaps the most critical part of successfully applying the LT3574. In addition to the usual list of caveats dealing with high frequency isolated power supply transformer design, the following information should be carefully considered. Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed flyback transformers for use with the LT3574. Table 5 shows the details of several of these transformers. Table 5. Predesigned Transformers—Typical Specifications Unless Otherwise Noted TRANSFORMER PART NUMBER PA3018NL PA2626NL PA2627NL PA3019NL PA3020NL PA3021NL 750311304 750310564 750370040 750370041 750370047 750311307 750311308 L10-1022 SIZE (W × L × H) mm 12.70 × 10.67 × 9.14 12.70 × 10.67 × 9.14 15.24 × 13.1 × 11.45 12.70 × 10.67 × 9.14 12.70 × 10.67 × 9.14 12.70 × 10.67 × 9.14 15.24 × 13.3 × 11.43 15.24 × 13.3 × 11.43 9.14 × 9.78 × 10.54 9.14 × 9.78 × 10.54 13.35 × 10.8 × 9.14 15.24 × 13.3 × 11.43 15.24 × 13.3 × 11.43 9.52 × 9.52 × 4.95 LPRI (µH) 50 30 50 50 60 50 50 63 30 50 30 100 100 30 LLEAKAGE (nH) 700 403 766 700 680 195 825 450 150 450 150 2000 2090 NP:NS:NB 4:1:1 3:1:1 3:1:1 3:1:1 2:1:0.33 1:1:0.33 4:1:1.5 3:1:1 3:1:1 3:1:1 3:1:1 2:1:0.33 1:1:0.33 1:1 RPRI (mΩ) 250 240 420 250 400 100 146 115 60 190 60 173 325 0.142 RSEC (mΩ) 32 66 44 72 200 200 17 50 12.5 26 12.5 104 480 0.142 VENDOR Pulse Engineering Pulse Engineering Pulse Engineering Pulse Engineering Pulse Engineering Pulse Engineering Würth Elektronik Würth Elektronik Würth Elektronik Würth Elektronik Würth Elektronik Würth Elektronik Würth Elektronik BH Electronics TARGET APPLICATIONS 3.3V, 0.7A 5V, 0.5A 5V, 0.5A 5V, 0.5A 12V, 0.25A 15V, 0.15A 3.3V, 0.7A ±5V, 0.5A 5V, 0.5A 5V, 0.5A 5V, 0.5A 12V, 0.25A 15V, 0.15A 5V, 0.1A 3574f  LT3574 applications inForMation Turns Ratio Note that when using an RFB /RREF resistor ratio to set output voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application. In contrast, simpler ratios of small integers, e.g., 1:1, 2:1, 3:2, etc., can be employed to provide more freedom in setting total turns and mutual inductance. Typically, the transformer turns ratio is chosen to maximize available output power. For low output voltages (3.3V or 5V), a N:1 turns ratio can be used with multiple primary windings relative to the secondary to maximize the transformer’s current gain (and output power). However, remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. This quantity needs to remain below the abs max rating of the SW pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio, N, for a given application. Choose a turns ratio low enough to ensure: N< 50 V – VIN(MAX ) VOUT + VF Leakage Inductance Transformer leakage inductance (on either the primary or secondary) causes a voltage spike to appear at the primary after the output switch turns off. This spike is increasingly prominent at higher load currents where more stored energy must be dissipated. In most cases, a snubber circuit will be required to avoid overvoltage breakdown at the output switch node. Transformer leakage inductance should be minimized. An RCD (resistor capacitor diode) clamp, shown in Figure 4, is required for most designs to prevent the leakage inductance spike from exceeding the breakdown voltage of the power device. The flyback waveform is depicted in Figure 5. In most applications, there will be a very fast voltage spike caused by a slow clamp diode that may not exceed 60V. Once the diode clamps, the leakage inductance current is absorbed by the clamp capacitor. This period should not last longer than 150ns so as not to interfere with the output regulation, and the voltage during this clamp period must not exceed 55V. The clamp diode turns off after the leakage inductance energy is absorbed and the switch voltage is then equal to: VSW(MAX) = VIN(MAX) + N(VOUT + VF) This voltage must not exceed 50V. This same equation also determines the maximum turns ratio. When choosing the snubber network diode, careful attention must be paid to maximum voltage seen by the SW pin. Schottky diodes are typically the best choice to be used in the snubber, but some PN diodes can be used if they turn on fast enough to limit the leakage inductance spike. The leakage spike must always be kept below 60V. Figures 6 and 7 show the SW pin waveform for a 24VIN, 5VOUT application at a 0.5A load current. Notice that the leakage spike is very high (more than 65V) with the bad diode, while the good diode effectively limits the spike to less than 55V. For larger N:1 values, a transformer with a larger physical size is needed to deliver additional current and provide a large enough inductance value to ensure that the off-time is long enough to accurately measure the output voltage. For lower output power levels, a 1:1 or 1:N transformer can be chosen for the absolute smallest transformer size. A 1:N transformer will minimize the magnetizing inductance (and minimize size), but will also limit the available output power. A higher 1:N turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch. 3574f  LT3574 applications inForMation LS C R VSW < 60V • D • < 55V < 50V 3574 F04 t OFF > 350ns tSP < 150ns 3574 F05 Figure 4. RCD Clamp TIME Figure 5. Maximum Voltages for SW Pin Flyback Waveform 10V/DIV 10V/DIV 100ns/DIV 3574 F06 100ns/DIV 3574 F07 Figure 6. Good Snubber Diode Limits SW Pin Voltage Figure 7. Bad Snubber Diode Does Not Limit SW Pin Voltage Secondary Leakage Inductance In addition to the previously described effects of leakage inductance in general, leakage inductance on the secondary in particular exhibits an additional phenomenon. It forms an inductive divider on the transformer secondary that effectively reduces the size of the primary-referred flyback pulse used for feedback. This will increase the output voltage target by a similar percentage. Note that unlike leakage spike behavior, this phenomenon is load independent. To the extent that the secondary leakage inductance is a constant percentage of mutual inductance (over manufacturing variations), this can be accommodated by adjusting the RFB /RREF resistor ratio. Winding Resistance Effects Resistance in either the primary or secondary will reduce overall efficiency (POUT /PIN). Good output voltage regulation will be maintained independent of winding resistance due to the boundary mode operation of the LT3574. Bifilar Winding A bifilar, or similar winding technique, is a good way to minimize troublesome leakage inductances. However, remember that this will also increase primary-to-secondary capacitance and limit the primary-to-secondary breakdown voltage, so, bifilar winding is not always practical. The Linear Technology applications group is available and extremely qualified to assist in the selection and/or design of the transformer. 3574f  LT3574 applications inForMation Setting the Current Limit Resistor The maximum current limit can be set by placing a resistor between the RILIM pin and ground. This provides some flexibility in picking standard off-the-shelf transformers that may be rated for less current than the LT3574’s internal power switch current limit. If the maximum current limit is needed, use a 10k resistor. For lower current limits, the following equation sets the approximate current limit: RILIM = 65 • 10 3(0 . 9 A − ILIM ) + 10k The Switch Current Limit vs RILIM plot in the Typical Performance Characteristics section depicts a more accurate current limit. Undervoltage Lockout (UVLO) The SHDN/UVLO pin is connected to a resistive voltage divider connected to VIN as shown in Figure 8. The voltage threshold on the SHDN/UVLO pin for VIN rising is 1.22V. To introduce hysteresis, the LT3574 draws 2.5µA from the SHDN/UVLO pin when the pin is below 1.22V. The hysteresis is therefore user-adjustable and depends on the value of R1. The UVLO threshold for VIN rising is: VIN(UVLO,RISING) = 1 . 22V • (R1 + R2) + 2 . 5µA • R1 R2 To implement external run/stop control, connect a small NMOS to the UVLO pin, as shown in Figure 8. Turning the NMOS on grounds the UVLO pin and prevents the LT3574 from operating, and the part will draw less than a 1µA of quiescent current. Minimum Load Requirement The LT3574 obtains output voltage information through the transformer while the secondary winding is conducting current. During this time, the output voltage (multiplied times the turns ratio) is presented to the primary side of the transformer. The LT3574 uses this reflected signal to regulate the output voltage. This means that the LT3574 must turn on every so often to sample the output voltage, which delivers a small amount of energy to the output. This sampling places a minimum load requirement on the output of 1% to 2% of the maximum load. BIAS Pin Considerations For applications with an input voltage less than 15V, the BIAS pin is typically connected directly to the VIN pin. For input voltages greater than 15V, it is preferred to leave the BIAS pin separate from the VIN pin. In this condition, the BIAS pin is regulated with an internal LDO to a voltage of 3V. By keeping the BIAS pin separate from the input voltage at high input voltages, the physical size of the capacitors can be minimized (the BIAS pin can then use a 6.3V or 10V rated capacitor). The UVLO threshold for VIN falling is: VIN(UVLO,FALLING) = 1 . 22V • (R1 + R2) R2 VIN R1 SHDN/UVLO LT3574 R2 RUN/STOP CONTROL (OPTIONAL) GND 3574 F08 Figure 8. Undervoltage Lockout (UVLO) 3574f  LT3574 applications inForMation Overdriving the BIAS Pin with a Third Winding The LT3574 provides excellent output voltage regulation without the need for an opto-coupler, or third winding, but for some applications with higher input voltages (>20V), it may be desirable to add an additional winding (often called a third winding) to improve the system efficiency. For proper operation of the LT3574, if a winding is used as a supply for the BIAS pin, ensure that the BIAS pin voltage is at least 3.15V and always less than the input voltage. For a typical 24VIN application, overdriving the BIAS pin will improve the efficiency gain 4% to 5%. Loop Compensation The LT3574 is compensated using an external resistorcapacitor network on the VC pin. Typical values are in the range of RC = 50k and CC = 1nF (see the numerous schematics in the Typical Applications section for other possible values). If too large of an RC value is used, the part will be more susceptible to high frequency noise and jitter. If too small of an RC value is used, the transient performance will suffer. The value choice for CC is somewhat the inverse of the RC choice: if too small a CC value is used, the loop may be unstable, and if too large a CC value is used, the transient performance will also suffer. Transient response plays an important role for any DC/DC converter. Design Example The following example illustrates the converter design process using LT3574. Given the input voltage of 20V to 28V, the required output is 5V, 0.5A. VIN(MIN) = 20V, VIN(MAX) = 28V, VOUT = 5V, VF = 0.5V and IOUT = 0.5A 1. Select the transformer turns ratio to accommodate the output. The output voltage is reflected to the primary side by a factor of turns ratio N. The switch voltage stress VSW is expressed as: N= NP NS VSW(MAX ) = VIN + N( VOUT + VF ) < 50 V or rearranged to: N< 50 − VIN(MAX ) ( VOUT + VF ) On the other hand, the primary-side current is multiplied by the same factor of N. The converter output capability is: IOUT(MAX ) = 0 . 8 • (1 − D) • D= N( VOUT + VF ) VIN + N( VOUT + VF ) 1 NI 2 PK The transformer turns ratio is selected such that the converter has adequate current capability and a switch stress below 50V. Table 6 shows the switch voltage stress and output current capability at different transformer turns ratio. Table 6. Switch Voltage Stress and Output Current Capability vs Turns Ratio N 1:1 2:1 3:1 4:1 VSW(MAX) AT VIN(MAX) ( V) 33.5 39 44.5 50 IOUT(MAX) AT VIN(MIN) ( A) 0.34 0.57 0.73 0.84 DUTY CYCLE (%) 16~22 28~35 37~45 44~52 3574f  LT3574 applications inForMation BIAS winding turns ratio is selected to program the BIAS voltage to 3~5V. The BIAS voltage shall not exceed the input voltage. The turns ratio is then selected as primary: secondary: BIAS = 3:1:1. 2. Select the transformer primary inductance for target switching frequency. The LT3574 requires a minimum amount of time to sample the output voltage during the off-time. This off-time, tOFF(MIN), shall be greater than 350ns over all operating conditions. The converter also has a minimum current limit, IMIN, of 175mA to help create this off-time. This defines the minimum required inductance as defined as: L MIN = N( VOUT + VF ) • t OFF(MIN) IMIN In this design example, the minimum primary inductance is used to achieve a nominal switching frequency of 350kHz at full load. The PA2627NL from Pulse Engineering is chosen as the flyback transformer. Given the turns ratio and primary inductance, a customized transformer can be designed by magnetic component manufacturer or a multi-winding transformer such as a Coiltronics Versa-Pac may be used. 3. Select the output diodes and output capacitor. The output diode voltage stress VD is the summation of the output voltage and reflection of input voltage to the secondary side. The average diode current is the load current. VD = VOUT + VIN N The transformer primary inductance also affects the switching frequency which is related to the output ripple. If above the minimum inductance, the transformer’s primary inductance may be selected for a target switching frequency range in order to minimize the output ripple. The following equation estimates the switching frequency. fSW = 1 1 = IPK IPK t ON + t OFF + VIN NPS ( VOUT + VF ) L L The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor. The following equation calculates the output voltage ripple. D VMAX = LI 2PK 2 CVOUT 4. Select the snubber circuit to clamp the switch voltage spike. A flyback converter generates a voltage spike during switch turn-off due to the leakage inductance of the transformer. In order to clamp the voltage spike below the maximum rating of the switch, a snubber circuit is used. There are many types of snubber circuits, for example R-C, R-C-D and Zener clamps. Among them, RCD is widely used. Figure 9 shows the RCD snubber in a flyback converter. A typical switch node waveform is shown in Figure 10. Table 7. Switching Frequency at Different Primary Inductance at IPK L (µH) 30 60 120 fSW AT VIN(MIN) (kHz) 317 159 79 fSW AT VIN(MAX) (kHz) 373 187 93 Note: The switching frequency is calculated at maximum output. 3574f  LT3574 applications inForMation LS C R D VC NVOUT • • 3574 F09 VIN tSP 3574 F10 Figure 9. RCD Snubber in a Flyback Converter Figure 10. Typical Switch Node Waveform During switch turn-off, the energy stored in the leakage inductance is transferred to the snubber capacitor, and eventually dissipated in the snubber resistor. V ( V − N • VOUT ) 1 L S I2PK fSW = C C 2 R The snubber resistor affects the spike amplitude VC and duration tSP, the snubber resistor is adjusted such that tSP is about 150ns. Prolonged tSP may cause distortion to the output voltage sensing. The previous steps finish the flyback power stage design. 5. Select the feedback resistor for proper output voltage. Using the resistor Tables 1-4, select the feedback resistor RFB, and program the output voltage to 5V. Adjust the RTC resistor for temperature compensation of the output voltage. RREF is selected as 6.04k. A small capacitor in parallel with RREF filters out the noise during the voltage spike, however, the capacitor should limit to 10pF A large capacitor causes distortion on volt. age sensing. 6. Optimize the compensation network to improve the transient performance. The transient performance is optimized by adjusting the compensation network. For best ripple performance, select a compensation capacitor not less than 1nF and select a , compensation resistor not greater than 50k. 7. Current limit resistor, soft-start capacitor and UVLO resistor divider Use the current limit resistor RLIM to lower the current limit if a compact transformer design is required. Soft-start capacitor helps during the start-up of the flyback converter. Select the UVLO resistor divider for intended input operation range. These equations are aforementioned. 3574f  LT3574 typical applications Low Input Voltage 5V Isolated Flyback Converter VIN 5V 3:1 C1 10µF R1 200k R2 90.9k SHDN/UVLO D2 RFB RREF R3 80.6k VIN C6 0.22µF R8 T1 2k 30µH D1 C5 22µF VOUT+ 5V 175mA 3.3µH VOUT– LT3574 R4 6.04k TC RILIM SS VC R6 28.7k R5 10k C2 10nF GND TEST BIAS R7 59k C3 1500pF VIN SW 3574 TA02 T1: PULSE PA2626NL OR WÜRTH ELEKTRONIK 750370040 D1: B340A D2: PMEG6010 C5: MURATA, GRM32ER71A226K ±12V Isolated Flyback Converter VIN 5V 2:1:1 C1 10µF R1 200k R2 90.9k SHDN/UVLO R3 118k VIN C6 0.22µF R8 T1 2k 43.6µH D3 RFB RREF D1 C5 22µF VOUT1+ 12V 40mA • • 10.9µH D2 C6 22µF VOUT 1– VOUT2+ LT3574 10.9µH R4 6.04k TC RILIM SS VC R6 59k R5 10k C2 10nF GND TEST BIAS R7 56.2k C3 0.01µF 3574 TA03 VOUT 2– –12V 40mA SW VIN T1: COILTRONICS VPH1-0076-R D1, D2: B240A D3: PMEG6010 C5, C6: MURATA, GRM32ER71A226K 3574f  LT3574 typical applications 5V Isolated Flyback Converter VIN 12V TO 24V (30V*) 3:1:1 C1 10µF R1 499k R2 71.5k SHDN/UVLO R3 80.6k T1: PULSE PA3019NL OR WÜRTH ELEKTRONIK 750370041 D1: B340A D3: PMEG6010 C5: MURATA, GRM32ER71A226K VIN C6 0.22µF R8 T1 4.02k 50µH D3 RFB RREF D1 C5 22µF VOUT + 5V 350mA • • 5.6µH VOUT – LT3574 R4 6.04k TC RILIM SS VC R6 28.7k R5 10k C2 10nF R7 59k C3 1000pF C4 4.7µF GND TEST BIAS D2 L1C 5.6µH SW • *OPTIONAL THIRD WINDING FOR 30V OPERATION 3574 TA04 Efficiency 90 80 70 EFFICIENCY (%) 60 50 40 30 20 10 0 0 100 200 300 400 500 VIN = 24V VIN = 12V 600 700 IOUT (mA) 3574 TA04b 3574f  LT3574 typical applications 3.3V Isolated Flyback Converter VIN 12V TO 24V (36V*) 4:1:1 C1 10µF R1 499k R2 71.5k SHDN/UVLO R3 76.8k VIN C6 0.22µF R8 T1 2k 50µH D3 D1 VOUT + 3.3V 0.5A • • 3.1µH C5 47µF VOUT – LT3574 RFB RREF R4 6.04k TC RILIM SS VC R6 19.1k R5 10k C2 10nF GND TEST BIAS R7 25.5k C3 1500pF D2 L1C 3.1µH SW T1: PULSE PA3018NL OR WÜRTH ELEKTRONIK 750311304 D1: B340A D3: PMEG6010 C4 4.7µF *OPTIONAL THIRD WINDING FOR 36V OPERATION 3574 TA05 12V Isolated Flyback Converter VIN 12V D1 C1 10µF 3:1 R1 499k R2 71.5k SHDN/UVLO VIN C6 0.22µF R8 T1 2k 58.5µH D2 LT3574 RFB RREF TC RILIM SS VC R6 59k R5 10k C2 10nF R3 178k R4 6.04k VOUT 12V 150mA • • 6.5µH C5 22µF VOUT– SW GND TEST BIAS R7 40.2k C3 4700pF VIN 3574 TA06 T1: COILTRONICS VP1-0102-R D1: B340A D2: PMEG6010 3574f 0 LT3574 typical applications Four Output 12V Isolated Flyback Converter D1 C5 22µF VOUT1+ 12V 40mA VOUT 1– VOUT2+ 12V 40mA 2:1:1:1:1 C1 10µF R1 499k SHDN/UVLO LT3574 TC RILIM SS VC R6 59k R5 10k C2 10nF R7 20k C3 0.1µF SW GND TEST BIAS VIN R3 118k VIN C6 0.22µF R8 2k D5 T1 43.6µH VIN 12V TO 24V • • 10.9µH D2 R2 71.5k • RFB RREF 10.9µH D3 C6 22µF VOUT 2– VOUT3+ 12V 40mA R4 6.04k • 10.9µH D4 C7 22µF VOUT 3– VOUT4+ 12V 40mA T1: COILTRONICS VPH1-0076-R D1-D4: B240A D5: PMEG6010 • 10.9µH C8 22µF VOUT 4– 3574 TA07 3574f  LT3574 typical applications 5V Isolated Flyback Converter Using Coupling Inductor VIN 5V D1 C1 10µF 1:1 R1 200k R2 90.9k SHDN/UVLO R3 26.1k R4 6.04k 3574 TA09 VIN C6 0.22µF R8 2k T1 30µH • 30µH • C5 47µF VOUT+ 5V 0.1A D2 VOUT– LT3574 RFB RREF TC RILIM SS VC R6 26.1k R5 10k C2 10nF SW GND TEST BIAS R7 54.9k C3 3300pF VIN T1: BH ELECTRONICS, L10-1022 D1: B220A D2: CMDSH-3 3574f  LT3574 package Description (Reference LTC DWG # 05-08-1669 Rev Ø) 0.889 (.035 0.127 .005) MS Package 16-Lead Plastic MSOP 5.23 (.206) MIN 3.20 – 3.45 (.126 – .136) 0.305 0.038 (.0120 .0015) TYP 0.50 (.0197) BSC 4.039 0.102 (.159 .004) (NOTE 3) 16151413121110 9 RECOMMENDED SOLDER PAD LAYOUT DETAIL “A” 0 – 6 TYP 0.280 0.076 (.011 .003) REF 0.254 (.010) GAUGE PLANE 4.90 0.152 (.193 .006) 3.00 0.102 (.118 .004) (NOTE 4) 0.53 0.152 (.021 .006) DETAIL “A” 0.18 (.007) SEATING PLANE 1.10 (.043) MAX 12345678 0.86 (.034) REF 0.50 NOTE: (.0197) 1. DIMENSIONS IN MILLIMETER/(INCH) BSC 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 0.17 – 0.27 (.007 – .011) TYP 0.1016 (.004 0.0508 .002) MSOP (MS16) 1107 REV Ø 3574f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.  LT3574 typical application 10V to 30VIN, +5V/–5VOUT Isolated Flyback Converter VIN 10V TO 30V T1 3:1:1:1 C1 10µF VIN SHDN/UVLO LT3574 R3 80.6k C6 0.22µF R8 2k L1A 63µH D1 VOUT + 5V 175mA • • R1 357k R2 51.1k L1B 7µH C5 47µF D4 RFB RREF COM L1C 7µH D2 C6 47µF • TC RILIM SS VC R5 28.7k R6 10k C2 10nF R7 40.2k C3 2700pF SW GND TEST BIAS R4 6.04k VOUT – –5V 175mA D3 L1D 7µH 3574 TA11 *OPTIONAL THIRD WINDING FOR >24V OPERATION D1, D2: B340A D4: PMEG6010 T1: WÜRTH ELEKTRONIK 750310564 C4 4.7µF relateD parts PART NUMBER LT3573 LT3757 LT3758 LT3837 LT3825 LT1725 LT1737 DESCRIPTION Isolated Flyback Switching Regulator with 60V Integrated Switch Boost, Flyback, SEPIC and Inverting Controller Boost, Flyback, SEPIC and Inverting Controller Isolated No-Opto Synchronous Flyback Controller Isolated No-Opto Synchronous Flyback Controller Isolated No-Opto Flyback Controller Isolated No-Opto Flyback Controller COMMENTS 3V ≤ VIN ≤ 40V, No Opto-Isolator or “Third Winding” Required, Up to 7W, MSOP-16E 2.9V ≤ VIN ≤ 40V, Current Mode Control, 100kHz to 1MHz Programmable Operation Frequency, 3mm × 3mm DFN-10 and MSOP-10E Package 5.5V ≤ VIN ≤ 100V, Current Mode Control, 100kHz to 1MHz Programmable Operation Frequency, 3mm × 3mm DFN-10 and MSOP-10E Package Ideal for VIN from 4.5V to 36V Limited by External Components, Up to 60W, Current Mode Control VIN from 16V to 75V Limited by External Components, Up to 60W, Current Mode Control VIN and VOUT Limited Only by External Components, Ideal for 48V Nominal Input Voltage VIN and VOUT Limited Only by External Components, Ideal for 24V Nominal Input Voltage Adjustable Switching Frequency, 2.5V ≤ VIN ≤ 36V, Burst Mode® Operation at Light Loads VIN and VOUT Limited Only by External Components, 6-Pin ThinSOT™ Package VIN and VOUT Limited Only by External Components, 2mm × 3mm DFN-8 or 8-Pin ThinSOT Packages VIN and VOUT Limited Only by External Components, 3mm × 3mm DFN-10, MSOP-10E LTC®1871/LTC1871-1 No RSENSE™ Low Quiescent Current Flyback, Boost LTC1871-7 and SEPIC Controllers LTC3803/LTC3803-3 LTC3803-5 LTC3873/LTC3873-5 LTC3805/LTC3805-5 200kHz or 300kHz Flyback DC/DC Controllers No RSENSE Constant Frequency Flyback, Boost, SEPIC Controllers Adjustable Fixed 70kHz to 700kHz Operating Frequency Flyback Controllers 3574f  Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● LT 0110 • PRINTED IN USA www.linear.com  LINEAR TECHNOLOGY CORPORATION 2010