LT3956EUHE-PBF

FeaTures n LT3956 80VIN, 80VOUT Constant-Current, Constant-Voltage Converter DescripTion The LT®3956 is a DC/DC converter designed to operate as a constant-current source and constant-voltage regulator. It is ideally suited for driving high current LEDs. It features an internal low side N-channel power MOSFET rated for 84V at 3.3A and driven from an internal regulated 7.15V supply. The fixed frequency, current-mode architecture results in stable operation over a wide range of supply and output voltages. A ground referenced voltage FB pin serves as the input for several LED protection features, and also makes it possible for the converter to operate as a constant-voltage source. A frequency adjust pin allows the user to program the frequency from 100kHz to 1MHz to optimize efficiency, performance or external component size. The LT3956 senses output current at the high side of the LED string. High side current sensing is the most flexible scheme for driving LEDs, allowing boost, buck mode or buck-boost mode configuration. The PWM input provides LED dimming ratios of up to 3000:1, and the CTRL input provides additional analog dimming capability. L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and True Color PWM is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 7199560 and 7321203. 3000:1 True Color PWMTM Dimming n Wide Input Voltage Range: 4.5V to 80V n Output Voltage Up to 80V n Internal 3.3A/84V Switch n Constant-Current and Constant-Voltage Regulation n 250mV High Side Current Sense n Drives LEDs in Boost, Buck Mode, Buck-Boost Mode, SEPIC or Flyback Topology n Adjustable Frequency: 100kHz to 1MHz n Open LED Protection n Programmable Undervoltage Lockout with Hysteresis n Constant-Voltage Loop Status Pin n PWM Disconnect Switch Driver n CTRL Pin Adjusts High Side Current Sense Threshold n Low Shutdown Current: 1.2V or ILED = (VCTRL –100mV)/(4 • RLED) if VCTRL < 1V. Input bias current is typically 20µA. Below 3V, ISN is an input to the short-circuit protection feature that forces GATE to 0V if ISP exceeds ISN by more than 350mV (typ). ISP: Connection point for the positive terminal of the current feedback resistor. Input bias current for this pin depends on CTRL pin voltage, as shown in the Typical Performance Characteristics. ISP is an input to the short-circuit protection feature when ISN is less than 3V. VC: Transconductance Error Amplifier Output Pin. This pin is used to stabilize the voltage loop with an RC network. This pin is high impedance when PWM is low, a feature that stores the demand current state variable for the next PWM high transition. Connect a capacitor between this pin and GND; a resistor in series with the capacitor is recommended for fast transient response. CTRL: Current Sense Threshold Adjustment Pin. Regulating threshold V(ISP – ISN) is 0.25 • VCTRL plus an offset for 0V < VCTRL < 1V. For VCTRL > 1.2V the current sense threshold is constant at the full-scale value of 250mV. For 1V < VCTRL < 1.2V, the dependence of the current sense threshold upon VCTRL transitions from a linear function to a constant value, reaching 98% of full-scale value by VCTRL = 1.1V. Connect CTRL to VREF for the 250mV default threshold. Do not leave this pin open. 3956f  LT3956 pin FuncTions VREF: Voltage Reference Output Pin (typically 2V). This pin drives a resistor divider for the CTRL pin, either for analog dimming or for temperature limit/compensation of LED load. Can supply up to 100μA. PWM: A signal low turns off switcher, idles oscillator and disconnects VC pin from all internal loads. PWMOUT pin follows PWM pin. PWM has an internal pull-down resistor. If not used, connect to INTVCC. VMODE: An open-collector pull-down on VMODE asserts if the FB input is greater than the FB regulation threshold minus 50mV (typical). To function, the pin requires an external pull-up resistor. When the PWM input is low and the DC/DC converter is idle, the VMODE condition is latched to the last valid state when the PWM input was high. When PWM input goes high again, the VMODE pin will be updated. This pin may be used to report an open LED fault. Use a pull-up current less than 1mA. SS: Soft-Start Pin. This pin modulates oscillator frequency and compensation pin voltage (VC) clamp. The soft-start interval is set with an external capacitor. The pin has a 10µA (typical) pull-up current source to an internal 2.5V rail. The soft-start pin is reset to GND by an undervoltage condition (detected by EN/UVLO pin) or thermal limit. RT: Switching Frequency Adjustment Pin. Set the frequency using a resistor to GND (for resistor values, see the Typical Performance curve or Table 1). Do not leave the RT pin open. block DiagraM + – EN/UVLO 1.22V 2.1µA A6 SHDN 1.31V FB VC PWMOUT PWM LDO 7.15V INTVCC VIN 1.25V 1.25V SHORT-CIRCUIT DETECT + gm – SCILMB A5 10µA AT FB = 1.25V SCILMB + – ISN ISP CTRL 350mV + A10 – 5k 10µA 1.1V + + A3 – CTRL BUFFER Q2 + A1 – gm EAMP 10µA AT A1+ = A1– + A2 – R S PWM COMPARATOR VC SSCLAMP RAMP GENERATOR 100kHz TO 1MHz OSCILLATOR 20k FAULT LOGIC TSD 165°C 10µA VREF 1.25V 1.2V FREQ PROG FB 2V A7 170k 3956 BD SS RT  + – 140µA + + – + – OVFB COMPARATOR + – A8 SW Q DRIVER ISENSE + A4 – PGND GND VMODE 1mA (MAX) + – 3956f LT3956 operaTion The LT3956 is a constant-frequency, current mode converter with a low side N-channel MOSFET switch. The switch and PWMOUT pin drivers, and other chip loads, are powered from INTVCC, which is an internally regulated supply. In the discussion that follows, it will be helpful to refer to the Block Diagram of the IC. In normal operation, with the PWM pin low, the power switch is turned off and the PWMOUT pin is driven to GND, the VC pin is high impedance to store the previous switching state on the external compensation capacitor, and the ISP and ISN pin bias currents are reduced to leakage levels. When the PWM pin transitions high, the PWMOUT pin transitions high after a short delay. At the same time, the internal oscillator wakes up and generates a pulse to set the PWM latch, turning on the internal power MOSFET switch. A voltage input proportional to the switch current, sensed by an internal current sense resistor, is added to a stabilizing slope compensation ramp and the resulting switch-current sense signal is fed into the positive terminal of the PWM comparator. The current in the external inductor increases steadily during the time the switch is on. When the switch-current sense voltage exceeds the output of the error amplifier, labeled VC , the latch is reset and the switch is turned off. During the switch off phase, the inductor current decreases. At the completion of each oscillator cycle, internal signals such as slope compensation return to their starting points and a new cycle begins with the set pulse from the oscillator. Through this repetitive action, the PWM control algorithm establishes a switch duty cycle to regulate a current or voltage in the load. The VC signal is integrated over many switching cycles and is an amplified version of the difference between the LED current sense voltage, measured between ISP and ISN, and the target difference voltage set by the CTRL pin. In this manner, the error amplifier sets the correct peak switch-current level to keep the LED current in regulation. If the error amplifier output increases, more current is demanded in the switch; if it decreases, less current is demanded. The switch current is monitored during the on-phase and is not allowed to exceed the current limit threshold of 3.9A (typical). If the SW pin exceeds the current limit threshold, the SR latch is reset regardless of the output state of the PWM comparator. Likewise, at an ISP/ISN common mode voltage less than 3V, the difference between ISP and ISN is monitored to determine if the output is in a short-circuit condition. If the difference between ISP and ISN is greater than 335mV (typical), the SR latch will be reset regardless of the PWM comparator. These functions are intended to protect the power switch, as well as various external components in the power path of the DC/DC converter. In voltage feedback mode, the operation is similar to that described above, except the voltage at the VC pin is set by the amplified difference of the internal reference of 1.25V (nominal) and the FB pin. If FB is lower than the reference voltage, the switch current will increase; if FB is higher than the reference voltage, the switch demand current will decrease. The LED current sense feedback interacts with the FB voltage feedback so that FB will not exceed the internal reference and the voltage between ISP and ISN will not exceed the threshold set by the CTRL pin. For accurate current or voltage regulation, it is necessary to be sure that under normal operating conditions, the appropriate loop is dominant. To deactivate the voltage loop entirely, FB can be connected to GND. To deactivate the LED current loop entirely, the ISP and ISN should be tied together and the CTRL input tied to VREF . Two LED specific functions featured on the LT3956 are controlled by the voltage feedback pin. First, when the FB pin exceeds a voltage 50mV lower (–4%) than the FB regulation voltage, the pull-down driver on the VMODE pin is activated. This function provides a status indicator that the load may be disconnected and the constant-voltage feedback loop is taking control of the switching regulator. When the FB pin exceeds the FB regulation voltage by 60mV (5% typical), the PWMOUT pin is driven low, ignoring the state of the PWM input. In the case where the PWMOUT pin drives a disconnect NFET, this action isolates the LED load from GND, preventing excessive current from damaging the LEDs. If the FB input exceeds the overvoltage threshold (1.31V typical), then an externally driven overvoltage event may have caused the FB pin to be too high and the VMODE pull-down will be deactivated until the FB pin drops below the overvoltage threshold. 3956f  LT3956 applicaTions inForMaTion INTVCC Regulator Bypassing and Operation The INTVCC pin requires a capacitor for stable operation and to store the charge for the switch driver and PWMOUT pin switching currents. Choose a 10V rated low ESR, X7R or X5R ceramic capacitor for best performance. A 4.7µF capacitor will be adequate for many applications. Place the capacitor close to the IC to minimize the trace length to the INTVCC pin and also to the IC ground. An internal current limit on the INTVCC output protects the LT3956 from excessive on-chip power dissipation. The INTVCC pin has its own undervoltage disable (UVLO) set to 4.1V (typical) to protect the internal MOSFET from excessive power dissipation caused by not being fully enhanced. If the INTVCC pin drops below the UVLO threshold, the PWMOUT pin will be forced to 0V, the power switch turned off and the soft-start pin will be reset. If the input voltage, VIN, will not exceed 7V, then the INTVCC pin could be connected to the input supply. This action allows the LT3956 to operate from as low as 4.5V. Be aware that a small current (less than 12μA) will load the INTVCC in shutdown. Otherwise, the minimum operating VIN value is determined by the dropout voltage of the linear regulator and the 4.4V (4.1V typical) INTVCC undervoltage lockout threshold mentioned above. Programming the Turn-On and Turn-Off Thresholds With the EN/UVLO Pin The falling UVLO value can be accurately set by the resistor divider. A small 2.1µA pull-down current is active when EN/UVLO is below the falling threshold. The purpose of this current is to allow the user to program the rising hysteresis. The following equations should be used to determine the values of the resistors: VIN,FALLING = 1.22 • R1 + R2 R2 LED Current Programming The LED current is programmed by placing an appropriate value current sense resistor, RLED , between the ISP and ISN pins. Typically, sensing of the current should be done at the top of the LED string. If this option is not available, then the current may be sensed at the bottom of the string, but take caution that the minimum ISN value does not fall below 3V, which is the lower limit of the LED current regulation function. The CTRL pin should be tied to a voltage higher than 1.2V to get the full-scale 250mV (typical) threshold across the sense resistor. The CTRL pin can also be used to dim the LED current to zero, although relative accuracy decreases with the decreasing voltage sense threshold. When the CTRL pin voltage is less than 1V, the LED current is: ILED = VCTRL − 100mV RLED • 4 When the CTRL pin voltage is between 1V and 1.2V the LED current varies with CTRL, but departs from the previous equation by an increasing amount as the CTRL voltage increases. Ultimately, above CTRL = 1.2V, the LED current no longer varies with CTRL. At CTRL = 1.1V, the actual value of ILED is ~98% of the equation’s estimate. When VCTRL is higher than 1.2V, the LED current is regulated to: ILED = 250mV RLED VIN,RISING = 2.1µA • R1 + VIN,FALLING VIN LT3956 EN/UVLO R2 3956 F01 R1 Figure 1 The CTRL pin should not be left open (tie to VREF if not used). The CTRL pin can also be used in conjunction with a thermistor to provide overtemperature protection for the LED load, or with a resistor divider to VIN to reduce output power and switching current when VIN is low. The presence of a time varying differential voltage signal (ripple) across ISP and ISN at the switching frequency is expected. The amplitude of this signal is increased by high LED load current, low switching frequency and/or a smaller value output filter capacitor. Some level of ripple signal is acceptable: the compensation capacitor on the VC pin filters the signal so the average difference between ISP and ISN is regulated to the user-programmed value. Ripple voltage amplitude (peak-to-peak) in excess of 3956f 0 LT3956 applicaTions inForMaTion 20mV should not cause misoperation, but may lead to noticeable offset between the average value and the userprogrammed value. Output Current Capability An important consideration when using a switch with a fixed current limit is whether the regulator will be able to supply the load at the extremes of input and output voltage range. Several equations are provided to help determine this capability. Some margin to data sheet limits is included. For boost converters: IOUT(MAX ) ≤ 2.5A VIN(MIN) VOUT(MAX ) Programming Output Voltage (Constant-Voltage Regulation) or Open LED/Overvoltage Threshold For a boost or SEPIC application, the output voltage can be set by selecting the values of R3 and R4 (see Figure 2) according to the following equation: VOUT = 1.25 • R3 + R4 R4 For a boost type LED driver, set the resistor from the output to the FB pin such that the expected voltage level during normal operation will not exceed 1.1V. For an LED driver of buck mode or a buck-boost mode configuration, the output voltage is typically level-shifted to a signal with respect to GND as illustrated in Figure 3. The output can be expressed as: VOUT = VBE + 1.25 • R3 R4 + VOUT LT3956 FB R4 3956 F03 For buck mode converters: IOUT(MAX) ≤ 2.5A For SEPIC and buck-boost mode converters: VIN(MIN) IOUT(MAX ) ≤ 2.5A ( VOUT(MAX ) + VIN(MIN) ) These equations assume the inductor value and switching frequency have been selected so that inductor ripple current is ~600mA. Ripple current higher than this value will reduce available output current. Be aware that current limited operation at high duty cycle can greatly increase inductor ripple current, so additional margin may be required at high duty cycle. If some level of analog dimming is acceptable at minimum supply levels, then the CTRL pin can be used with a resistor divider to VIN (as shown on page 1) to provide a higher output current at nominal VIN levels. VOUT LT3956 FB R4 3956 F02 R3 RLED LED ARRAY COUT – 100k Figure 3. Feedback Resistor Connection for Buck Mode or Buck-Boost Mode LED Driver ISP/ISN Short-Circuit Protection Feature for SEPIC The ISP and ISN pins have a protection feature independent of the LED current sense feature that operates at ISN below 3V. The purpose of this feature is to provide continuous current sensing when ISN is below the LED current sense common mode range (during start-up or an output short-circuit fault) to prevent the development of excessive switching currents that could damage the power components in a SEPIC converter. The action threshold (335mV, typ) is above the default LED current sense threshold, so that no interference will occur over the ISN voltage range where these two functions overlap. This feature acts in the same manner as switch-current limit — it prevents switch turn-on until the ISP/ISN difference falls below the threshold. 3956f R3 Figure 2. Feedback Resistor Connection for Boost or SEPIC LED Drivers  LT3956 applicaTions inForMaTion Dimming Control There are two methods to control the current source for dimming using the LT3956. One method uses the CTRL pin to adjust the current regulated in the LEDs. A second method uses the PWM pin to modulate the current source between zero and full current to achieve a precisely programmed average current. To make this method of current control more accurate, the switch demand current is stored on the VC node during the quiescent phase when PWM is low. This feature minimizes recovery time when the PWM signal goes high. To further improve the recovery time, a disconnect switch may be used in the LED current path to prevent the ISP node from discharging during the PWM signal low phase. The minimum PWM on or off time will depend on the choice of operating frequency through the RT input. For best overall performance, the minimum PWM low or high time should be at least six switching cycles (6μs for fSW = 1MHz). Programming the Switching Frequency The RT frequency adjust pin allows the user to program the switching frequency from 100kHz to 1MHz to optimize efficiency/performance or external component size. Higher frequency operation yields smaller component size but increases switching losses and gate driving current, and may not allow sufficiently high or low duty cycle operation. Lower frequency operation gives better performance at the cost of larger external component size. For an appropriate RT resistor value see Table 1. An external resistor from the RT pin to GND is required—do not leave this pin open. Table 1. Switching Frequency vs RT Value fOSC (kHz) 1000 900 800 700 600 500 400 300 200 100 RT (k) 10 11.8 13 15.4 17.8 21 26.7 35.7 53.6 100 300 250 200 TIME (ns) 150 100 50 0 –50 MINIMUM ON-TIME MINIMUM OFF-TIME Duty Cycle Considerations Switching duty cycle is a key variable defining converter operation, therefore, its limits must be considered when programming the switching frequency for a particular application. The fixed minimum on-time and minimum off-time (see Figure 4) and the switching frequency define the minimum and maximum duty cycle of the switch, respectively. The following equations express the minimum/maximum duty cycle: Min Duty Cycle = (minimum on-time) • switching frequency Max Duty Cycle = 1 – (minimum off-time) • switching frequency When calculating the operating limits, the typical values for on/off-time in the data sheet should be increased by at least 60ns to allow margin for PWM control latitude and SW node rise/fall times. –25 50 25 0 75 TEMPERATURE (°C) 100 125 3956 F04 Figure 4. Typical Switch Minimum On and Off Pulse Width vs Temperature Thermal Considerations The LT3956 is rated to a maximum input voltage of 80V. Careful attention must be paid to the internal power dissipation of the IC at higher input voltages to ensure that a junction temperature of 125°C is not exceeded. This junction limit is especially important when operating at high ambient temperatures. If the LT3956’s junction temperature reaches 165°C (typ), the power switch will be turned off and the soft-start (SS) pin will be discharged to GND. Switching 3956f  LT3956 applicaTions inForMaTion will be enabled after the device temperature drops 10°C. This function is intended to protect the device during momentary overload conditions. The major contributors to internal power dissipation are the current in the linear regulator to drive the switch, and the ohmic losses in the switch. The linear regulator power is proportional to VIN and switching frequency, so at high VIN the switching frequency should be chosen carefully to ensure that the IC does not exceed a safe junction temperature. The internal junction temperature of the IC can be estimated by: TJ = TA + [VIN • (IQ + fSW • 7nC) + ISW2 • 0.14Ω • DSW] • θJA where TA is the ambient temperature, IQ is the quiescent current of the part (maximum 1.7mA) and θJA is the package thermal impedance (43°C/W for the 5mm × 6mm QFN package). For example, an application with TA(MAX) = 85°C, VIN(MAX) = 60V, fSW = 400kHz, and having an average switching current of 2.5A at 70% duty cycle, the maximum IC junction temperature will be approximately: TJ = 85°C + [(2.5A)2 • 0.14Ω • 0.7 + 60V • (1.7mA + 400kHz • 7nC)] • 43°C/W= 123°C The Exposed Pads on the bottom of the package must be soldered to a plane. These should then be connected to internal copper planes with thermal vias placed directly under the package to spread out the heat dissipated by the IC. Open LED Detection The LT3956 provides an open-drain status pin, VMODE, that pulls low when the FB pin is within ~50mV of its 1.25V regulated voltage. If the open LED clamp voltage is programmed correctly using the FB pin, then the FB pin should never exceed 1.1V when LEDs are connected, therefore, the only way for the FB pin to be within 50mV of the regulation voltage is for an open LED event to have occurred. Input Capacitor Selection The input capacitor supplies the transient input current for the power inductor of the converter and must be placed and sized according to the transient current requirements. The switching frequency, output current and tolerable input voltage ripple are key inputs to estimating the capacitor value. An X7R type ceramic capacitor is usually the best choice since it has the least variation with temperature and DC bias. Typically, boost and SEPIC converters require a lower value capacitor than a buck mode converter. Assuming that a 100mV input voltage ripple is acceptable, the required capacitor value for a boost converter can be estimated as follows: V 1µF CIN(µF ) = ILED( A ) • OUT • TSW(µs) • VIN A • µs Therefore, a 4.7µF capacitor is an appropriate selection for a 400kHz boost regulator with 12V input, 48V output and 1A load. With the same VIN voltage ripple of 100mV, the input capacitor for a buck converter can be estimated as follows: CIN(µF ) = ILED( A ) • TSW (µ s) • 4 . 7 µF A • µs A 10µF input capacitor is an appropriate selection for a 400kHz buck mode converter with a 1A load. In the buck mode configuration, the input capacitor has large pulsed currents due to the current returned through the Schottky diode when the switch is off. In this buck converter case it is important to place the capacitor as close as possible to the Schottky diode and to the PGND return of the switch. It is also important to consider the ripple current rating of the capacitor. For best reliability, this capacitor should have low ESR and ESL and have an adequate ripple current rating. The RMS input current for a buck mode LED driver is: IIN(RMS) = ILED • ( 1 – D) • D where D is the switch duty cycle. Table 2. Recommended Ceramic Capacitor Manufacturers MANUFACTURER TDK Kemet Murata Taiyo Yuden WEB SITE www.tdk.com www.kemet.com www.murata.com www.t-yuden.com 3956f  LT3956 applicaTions inForMaTion Output Capacitor Selection The selection of the output capacitor depends on the load and converter configuration, i.e., step-up or step-down and the operating frequency. For LED applications, the equivalent resistance of the LED is typically low and the output filter capacitor should be sized to attenuate the current ripple. Use of an X7R type ceramic capacitor is recommended. To achieve the same LED ripple current, the required filter capacitor is larger in the boost and buck-boost mode applications than that in the buck mode applications. Lower operating frequencies will require proportionately higher capacitor values. Soft-Start Capacitor Selection For many applications, it is important to minimize the inrush current at start-up. The built-in soft-start circuit significantly reduces the start-up current spike and output voltage overshoot. The soft-start interval is set by the softstart capacitor selection according to the equation: TSS = CSS • 2V 10µA it is important to consider diode leakage, which increases with the temperature, from the output during the PWM low interval. Therefore, choose the Schottky diode with sufficiently low leakage current. Table 3 has some recommended component vendors. Table 3. Schottky Rectifier Manufacturers VENDOR On Semiconductor Diodes, Inc. Central Semiconductor WEB SITE www.onsemi.com www.diodes.com www.centralsemi.com Inductor Selection The inductor used with the LT3956 should have a saturation current rating appropriate to the maximum switch current of 4.6A. Choose an inductor value based on operating frequency, input and output voltage to provide a current mode signal of approximately 0.6A magnitude. The following equations are useful to estimate the inductor value (TSW = 1/fOSC): LBUCK = TSW • VLED VIN – VLED VIN • 0.6 A ( ) A typical value for the soft-start capacitor is 0.01µF The . soft-start pin reduces the oscillator frequency and the maximum current in the switch. The soft-start capacitor is discharged when EN/UVLO falls below its threshold, during an overtemperature event or during an INTVCC undervoltage event. During start-up with EN/UVLO, charging of the soft-start capacitor is enabled after the first PWM high period. Schottky Rectifier Selection The power Schottky diode conducts current during the interval when the switch is turned off. Select a diode rated for the maximum SW voltage of the application and the RMS diode current. If using the PWM feature for dimming, LBUCK-BOOST = LBOOST = ( TSW • VLED • VIN VLED + VIN • 0.6 A ) TSW • VIN VLED – VIN VLED • 0.6 A ( ) Table 4 provides some recommended inductor vendors. Table 4. Inductor Manufacturers VENDOR Sumida Würth Elektronik Coiltronics Renco Coilcraft WEB SITE www.sumida.com www.we-online.com www.cooperet.com www.rencousa.com www.coilcraft.com 3956f  LT3956 applicaTions inForMaTion Loop Compensation The LT3956 uses an internal transconductance error amplifier whose VC output compensates the control loop. The external inductor, output capacitor and the compensation resistor and capacitor determine the loop stability. The inductor and output capacitor are chosen based on performance, size and cost. The compensation resistor and capacitor at VC are selected to optimize control loop response and stability. For typical LED applications, a 4.7nF compensation capacitor at VC is adequate, and a series resistor should always be used to increase the slew rate on the VC pin to maintain tighter regulation of LED current during fast transients on the input supply to the converter. Board Layout The high speed operation of the LT3956 demands careful attention to board layout and component placement. The exposed pads of the package are important for thermal management of the IC. It is crucial to achieve a good electrical and thermal contact between the GND exposed pad and the ground plane of the board. To reduce electromagnetic CSS RT 36 35 34 33 32 31 30 1 2 CVCC VIAS TO GND PLANE VIAS TO SW PLANE VIN R1 R2 3 4 GND 6 8 9 10 12 13 14 15 16 17 PGND VIAS L1 COUT SW 25 24 23 21 20 R4 R3 1 LT3956 28 27 VIA FROM LED+ LED– 3 M1 2 VIAS FROM PGND interference (EMI), it is important to minimize the area of the high dV/dt switching node between the inductor, SW pin and anode of the Schottky rectifier. Use a ground plane under the switching node to eliminate interplane coupling to sensitive signals. The lengths of the high dI/dt traces: 1) from the switch node through the switch to PGND, and 2) from the switch node through the Schottky rectifier and filter capacitor to PGND, should be minimized. The ground points of these two switching current traces should come to a common point then connect to the ground plane at the PGND pin of the LT3956 through a separate via to Pin 12, as shown in the suggested layout (Figure 5). Likewise, the ground terminal of the bypass capacitor for the INTVCC regulator should be placed near the GND of the IC. The ground for the compensation network and other DC control signals should be star connected to the GND Exposed Pad of the IC. Do not extensively route high impedance signals such as FB and VC, as they may pick up switching noise. Since there is a small variable DC input bias current to the ISN and ISP inputs, resistance in series with these pins should be minimized to avoid creating an offset in the current sense threshold. VMODE PWM CTRL CC RC VIA FROM VOUT D1 COUT RS CVIN VIN PGND VOUT LED+ VIA VIA LED+ 3956 F05 Figure 5. Boost Converter Suggested Layout 3956f  LT3956 Typical applicaTions 94% Efficient 25W White LED Headlamp Driver VIN 6V TO 60V (80V TRANSIENT) CVIN 2.2µF 2 R1 332k R2 100k VIN EN/UVLO 332k 40.2k VREF LT3956 CTRL INTVCC 100k VMODE PWM SS RT VC RC 20k CC 4.7nF ISN FB R3 1M R4 16.2k PWMOUT GND INTVCC INTVCC CVCC 4.7µF 25W LED STRING (CURRENT DERATED FOR VIN < 11V) L1 22µH SW PGND ISP RS 0.68 370mA D1 COUT 2.2µF 5 M1: VISHAY SILICONIX Si2328DS D1: DIODES INC PDS5100 L1: COILTRONICS DR125-220 C1, C2: MURATA GRM42-2x7R225 RT 28.7k 375kHz CSS 47nF M1 3956 TA02a SEE SUGGESTED LAYOUT (FIGURE 5) PWM Waveforms for 25W Headlamp Driver PWM ILED 200mA/DIV ILI 1A/DIV VOUT = 68V VIN = 15V 5µs/DIV 3956 TA02b 3956f  LT3956 Typical applicaTions Buck-Boost Mode LED Driver VIN 9V TO 45V L1 68µH C1 4.7µF 1M VIN EN/UVLO 187k VREF LT3956 INTVCC 100k CTRL ISN 619k FB VMODE PGND PWM SS RT PWMOUT VC GND INTVCC 3.4k 10nF INTVCC 4.7µF Q1 1k 3956 TA03a Efficiency vs VIN VOUT 4.7µF 35V VIN EFFICIENCY (%) 680m 100 96 D1 SW ISP 1µF 100V 92 10k 24V LED STRING 350mA 88 84 80 0 10 35.7k 300kHz 0.1µF 750 M1 VIN 20 30 VIN (V) 40 50 3956 TA03b L1: COILCRAFT MSS1038-683 D1: ON SEMICONDUCTOR MBRS3100T3 M1: ZETEX ZXM6IP03F Q1: ZETEX FMMT493 28VIN /0V to 28V SEPIC SuperCap Charger with Input Current Limit VIN 28V ≤ 1.2A 1µF VIN EN/UVLO LT3956 PWMOUT VMODE 10k INTVCC PWM VC C2 4.7µF 3956 TA04a 200m L1A 33µH C1 10µF 1:1 C4 10µF Input and Output Current vs Output Voltage VOUT 0V TO 28V 3.0 2.5 2.0 1.5 1.0 INPUT 0.5 0 OUTPUT D1 ISP ISN SW PGND FB C3 10µF 536k 25k 1M 40.2k INTVCC 2k Q1 59k 1M CTRL VREF SS RT GND 28.7k 375kHz 14k INPUT/OUTPUT CURRENT (A) L1B 0 5 10 15 20 VOUT (V) 25 30 3956 TA04b 10nF 30.1k L1: WÜRTH ELEKTRONIK 744871330 D1: ON SEMI MBRS36OT Q1: MMBTA42 C1, C3, C4: TAIYO-YUDEN GMK 3I6BJ106 3956f  LT3956 Typical applicaTions 90% Efficient, 20W SEPIC LED Driver L1A 33µH C1 4.7µF 50V 1:1 VIN EN/UVLO 185k VREF CTRL INTVCC 100k VMODE PWM SS RT VC 28.7k 375kHz 0.01µF 15k 10nF 25k LT3956 ISN 1M FB PWMOUT GND INTVCC C2 4.7µF 10V 56.2k 20W LED STRING CURRENT DERATED FOR VIN < 13V M1 3956 TA05a Efficiency vs VIN D1 C3 10µF 2 35V 100 96 EFFICIENCY (%) VIN 8V TO 50V C4 2.2µF (50V) 1M 250k SW PGND ISP L1B 92 88 0.25 1A 84 80 0 10 20 30 VIN (V) 40 50 3956 TA05b L1: COILTRONICS DRQ127-330 D1: VISHAY PDS5100 M1: ZETEX ZXM61N03F 90W Buck Mode LED Driver, 80VIN /60VOUT VIN 64V TO 80V 100 1M EN/UVLO INTVCC 100k VMODE VREF 24.3k CTRL 13k PWM SS RT VC 1k 20k LT3956 ISN FB M1 PWMOUT Q2 10k 16 WHITE LEDs, 90W L1 33µH SW PGND GND INTVCC INTVCC 4.7µF D1 VIN C1 2.2µF 4 3956 TA06a Efficiency vs VIN VIN ISP 470 0.1 1.5A Q1 267k 200k 200k C2 2.2µF 3 EFFICIENCY (%) 98 96 94 92 90 64 68 72 VIN (V) 76 80 3956 TA05b 28.7k 375kHz 0.1µF 0.01µF D1: VISHAY 10MQ100N L1: WÜRTH ELEKTRONIK 744066330 M1: VISHAY SILICONIX Si7113DN Q1: ZETEX FMMT593 Q2: ZETEX FMMT493 C1, C2: MURATA GRM42-2x7R225 3956f  LT3956 package DescripTion (Reference LTC DWG # 05-08-1836 Rev C) 28 27 25 24 23 21 20 0.70 0.05 UHE Package Variation: UHE28MA 36-Lead Plastic QFN (5mm 6mm) 30 5.50 0.05 4.10 0.05 31 32 33 34 35 36 3.00 1.88 0.05 0.05 0.12 0.05 3.00 1.53 0.05 0.05 17 16 15 14 PACKAGE OUTLINE 13 12 1.50 REF 0.48 0.05 1 2 3 4 6 0.50 BSC 5.10 6.50 0.05 0.05 8 9 0.25 0.05 2.00 REF 10 RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 0.75 5.00 0.10 0.05 PIN 1 NOTCH R = 0.30 OR 0.35 45 CHAMFER 1 1.88 0.10 3.00 0.12 0.10 2 0.10 3 4 R = 0.10 TYP 28 27 2.00 REF 25 24 30 31 32 1.50 REF 33 34 35 36 PIN 1 TOP MARK (NOTE 6) 6.00 0.10 23 0.48 21 20 1.53 0.10 3.00 0.10 0.10 6 8 R = 0.125 TYP 9 10 0.40 0.10 17 16 15 0.25 0.05 0.50 BSC 14 13 12 (UHE28MA) QFN 0110 REV C 0.200 REF 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 5. EXPOSED PAD SHALL BE SOLDER PLATED 2. DRAWING NOT TO SCALE 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION 3. ALL DIMENSIONS ARE IN MILLIMETERS ON THE TOP AND BOTTOM OF PACKAGE 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE 3956f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.  LT3956 Typical applicaTion Buck Mode 1A LED Driver with High Dimming Ratio and Open LED Reporting VIN 24V TO 80V 100 1M EN/UVLO 61.9k ISN FB VREF 30.1k CTRL 10k INTVCC 100k VMODE SW PWMOUT LT3956 Q1 1k 6 WHITE LEDs 20W M1 VIN ISP 750 0.1 1A Q2 20k 200k 200k 200k C2 4.7µF 5 EFFICIENCY (%) 96 Efficiency vs VIN 92 88 84 L1 33µH D1 C1 1µF 2 VIN 80 20 30 40 50 VIN (V) 60 70 80 0.1µF 28.7k 375kHz 47k 2.2nF PWM SS RT VC GND INTVCC PGND INTVCC 4.7µF 3956 TA07a 3956 TA06b D1: DIODES INC B1100/B L1: WÜRTH 74456133 M1: VISHAY SILICONIX Si5435BDC Q1: ZETEX FMMT493 Q2: ZETEX FMMT593 C1: TDKC3226X7R2A105K C2: TDKC3225X7RIE475K relaTeD parTs PART NUMBER LT3756/LT3756-1/ LT3756-2 LT3755/LT3755-1/ LT3755-2 LT3474 LT3475 LT3476 LT3477 LT3478/LT3478-1 DESCRIPTION 100VIN , 100VOUT, Full Featured LED Controller 40VIN , 75VOUT, Full Featured LED Controller 36V, 1A (ILED), 2MHz, Step-Down LED Driver Dual 1.5A (ILED), 36V, 2MHz Step-Down LED Driver COMMENTS VIN: 6V to 100V, VOUT(MAX) = 100V, True Color PWM Dimming = 3000:1, ISD < 1µA, 3mm × 3mm QFN-16 and MS16E Packages VIN: 4.5V to 40V, VOUT(MAX) = 60V, True Color PWM Dimming = 3000:1, ISD < 1µA, 3mm × 3mm QFN-16 and MS16E Packages VIN: 4V to 36V, VOUT(MAX) = 13.5V, True Color PWM Dimming = 400:1, ISD < 1µA, TSSOP16E Package VIN: 4V to 36V, VOUT(MAX) = 13.5V, True Color PWM Dimming = 3000:1, ISD < 1µA, TSSOP20E Package Quad Output 1.5A, 36V, 2MHz High Current LED Driver VIN: 2.8V to 16V, VOUT(MAX) = 36V, True Color PWM Dimming = 1000:1, with 1000:1 Dimming ISD < 10µA, 5mm × 7mm QFN Package 3A, 42V, 3MHz Boost, Buck-Boost, Buck LED Driver 4.5A, 42V, 2.5MHz High Current LED Driver with 3000:1 Dimming VIN: 2.5V to 25V, VOUT(MAX) = 40V, Dimming = Analog/PWM, ISD < 1µA, QFN and TSSOP20E Packages VIN: 2.8V to 36V, VOUT(MAX) = 42V, True Color PWM Dimming = 3000:1, ISD < 3µA, TSSOP16E Package 3956f 0 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● LT 0510 • PRINTED IN USA www.linear.com  LINEAR TECHNOLOGY CORPORATION 2010