LTC1142HV

LTC1142/LTC1142L/LTC1142HV Dual High Efficiency Synchronous Step-Down Switching Regulators FEATURES s s s DESCRIPTIO s s s s s s s Dual Outputs: 3.3V and 5V or User Programmable Ultra-High Efficiency: Over 95% Possible Current Mode Operation for Excellent Line and Load Transient Response High Efficiency Maintained over 3 Decades of Output Current Low Standby Current at Light Loads: 160µA/Output Independent Micropower Shutdown: IQ < 40µA Wide VIN Range: 3.5V to 20V Very Low Dropout Operation: 100% Duty Cycle Synchronous FET Switching for High Efficiency Available in Standard 28-Pin SSOP The LTC®1142/LTC1142L/LTC1142HV are dual synchronous step-down switching regulator controllers featuring automatic Burst ModeTM operation to maintain high efficiencies at low output currents. The devices are composed of two separate regulator blocks, each driving a pair of external complementary power MOSFETs, at switching frequencies up to 250kHz, using a constant off-time current mode architecture providing constant ripple current in the inductor. The operating current level for both regulators is user programmable via an external current sense resistor. Wide input supply range allows operation from 3.5V* to 18V (20V maximum). Constant off-time architecture provides low dropout regulation limited only by the RDS(ON) of the external MOSFET and resistance of the inductor and current sense resistor. The LTC1142 series is ideal for applications requiring dual output voltages with high conversion efficiencies over a wide load current range in a small amount of board space. , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. *For LTC1142L-ADJ only. APPLICATI s s s s S Notebook and Palmtop Computers Battery-Operated Digital Devices Portable Instruments DC Power Distribution Systems TYPICAL APPLICATI + CIN3 22µF 25V ×2 VIN 5.2V TO 18V 0.22µF P-CH Si9430DY 23 1 1000pF 28 SENSE – 3 NDRIVE 3 PGND3 SGND3 CT3 4 3 24 VIN3 PDRIVE 3 SENSE + 3 2 0V = NORMAL >1.5V = SHUTDOWN 16 SHUTDOWN 5 10 VIN5 PDRIVE 5 SENSE + 5 LTC1142HV SENSE – 5 NDRIVE 5 ITH3 27 RC3 1k ITH5 13 RC5 1k CT5 11 SGND5 PGND5 17 18 14 20 9 15 SHUTDOWN 3 0.22µF P-CH Si9430DY VOUT3 3.3V/2A RSENSE3 0.05Ω L1 50µH D1 1N5818 6 N-CH Si9410DY + COUT3 220µF 10V ×2 25 RSENSE3, RSENSE5 : SL-C1-1/2-1R050J L1, L2: COILTRONICS CTX50-2-MP PINS 5, 7, 8, 19, 21, 22: NC CC3 CT3 CT5 CC5 560pF 3300pF 3300pF 390pF 1142 F01 NOTE: COMPONENTS OPTIMIZED FOR HIGHEST EFFICIENCY, NOT MINIMUM BOARD SPACE. Figure 1. High Efficiency Dual 3.3V, 5V U + CIN5 22µF 25V ×2 L2 50µH RSENSE5 0.05Ω VOUT5 5V/2A 1000pF D2 1N5818 N-CH Si9410DY UO UO + COUT5 220µF 10V ×2 1 LTC1142/LTC1142L/LTC1142HV ABSOLUTE AXI U RATI GS Extended Commercial Temperature Range ........................... – 40°C to 85°C Junction Temperature (Note 1) ............................ 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C Input Supply Voltage (Pins 10, 24) LTC1142, LTC1142L-ADJ ..................... 16V to – 0.3V LTC1142HV, LTC1142HV-ADJ ............. 20V to – 0.3V Continuous Output Current (Pins 6, 9, 20, 23) .... 50mA Sense Voltages (Pins 1, 14, 15, 28).......... 13V to – 0.3V Operating Ambient Temperature Range ...... 0°C to 70°C PACKAGE/ORDER I FOR ATIO TOP VIEW SENSE +3 1 2 3 4 5 6 7 8 9 LTC1142 28 SENSE 27 ITH3 –3 ORDER PART NUMBER LTC1142CG LTC1142HVCG SHUTDOWN 3 SGND 3 PGND 3 NC NDRIVE 3 NC NC PDRIVE 5 26 INT VCC3 25 CT3 24 VIN3 23 PDRIVE 3 22 NC 21 NC 20 NDRIVE 5 19 NC 18 PGND5 17 SGND5 16 SHUTDOWN 5 15 SENSE + 5 VIN5 10 CT5 11 INT VCC5 12 ITH5 13 SENSE – 5 14 G PACKAGE, 28-LEAD SSOP TJMAX = 125°C, θJA = 95°C/W Consult factory for Industrial and Military grade parts. ELECTRICAL CHARACTERISTICS SYMBOL V2, V16 I2, I16 VOUT PARAMETER Feedback Voltage Feedback Current Regulated Output Voltage 3.3V Output 5V Output Output Voltage Line Regulation Output Voltage Load Regulation 3.3V Output 5V Output Output Ripple (Burst Mode) I10, I24 Input DC Supply Current (Note 2) Normal Mode Sleep Mode Shutdown Input DC Supply Current (Note 2) Normal Mode Sleep Mode Shutdown TA = 25°C, V10 = V24 = 10V, VSHUTDOWN = 0V, unless otherwise noted. MIN 1.21 q q q CONDITIONS LTC1142HV-ADJ, LTC1142L-ADJ : V10, V24 = 9V q LTC1142HV-ADJ, LTC1142L-ADJ LTC1142, LTC1142HV ILOAD = 700mA, V24 = 9V ILOAD = 700mA, V10 = 9V V10, V24 = 7V to 12V, ILOAD = 50mA Figure 1 Circuit 5mA < ILOAD < 2A 5mA < ILOAD < 2A ILOAD = 0A LTC1142 4V < V10, V24 < 12V 4V < V24 < 12V, 6V < V10 < 12V VSD1 = VSD2 = 2.1V, 4V < V10, V24 < 12V LTC1142HV, LTC1142HV-ADJ 4V < V10, V24 < 18V 4V < V24 < 18V, 6V < V10 < 18V VSD1 = VSD2 = 2.1V, 4V < V10, V24 < 18V q q ∆VOUT 2 U U W WW U W TOP VIEW SENSE + 1 1 28 SENSE – 1 27 ITH1 26 INT VCC1 25 CT1 24 VIN1 23 PDRIVE 1 LTC1142-ADJ 22 NC 21 NC 20 NDRIVE 2 19 PGND2 18 SGND2 17 SHUTDOWN 2 16 VFB2 15 SENSE + 2 ORDER PART NUMBER LTC1142HVCG-ADJ LTC1142LCG-ADJ VFB1 2 SHUTDOWN 1 3 SGND1 4 PGND1 5 NDRIVE 1 6 NC 7 NC 8 PDRIVE 2 9 VIN2 10 CT2 11 INT VCC2 12 ITH2 13 SENSE – 2 14 G PACKAGE, 28-LEAD SSOP TJMAX = 125°C, θJA = 95°C/W TYP 1.25 0.2 MAX 1.29 1 3.43 5.20 40 65 100 UNITS V µA V V mV mV mV mVP-P 3.23 4.90 – 40 3.33 5.05 0 40 60 50 1.6 160 10 1.6 160 10 2.1 230 20 2.3 250 22 mA µA µA mA µA µA LTC1142/LTC1142L/LTC1142HV ELECTRICAL CHARACTERISTICS SYMBOL PARAMETER Input DC Supply Current (Note 2) Normal Mode Sleep Mode Shutdown V1 – V28 V15 – V14 Current Sense Threshold Voltage CONDITIONS TA = 25°C, V10 = V24 = 10V, VSHUTDOWN = 0V, unless otherwise noted. MIN TYP 1.6 160 10 25 150 25 150 25 150 0.8 1.2 50 4 70 2 5 100 MAX 2.1 230 20 UNITS mA µA µA mV mV mV mV mV mV V µA µA µA µs ns LTC1142L-ADJ (Note 5) 3.5V < V10, V24 < 12V 3.5V < V10, V24 < 12V VSD1 = VSD2 = 2.1V, 3.5V < V10, V24 < 12V LTC1142HV-ADJ, LTC1142L-ADJ V14 = V28 = VOUT + 100mV, V2 = V16 = VREF + 25mV V14 = V28 = VOUT – 100mV, V2 = V16 = VREF – 25mV LTC1142, LTC1142HV V28 = VOUT + 100mV (Forced) V28 = VOUT – 100mV (Forced) LTC1142, LTC1142HV V14 = VOUT + 100mV (Forced) V14 = VOUT – 100mV (Forced) q 130 170 q 130 170 q 130 0.5 170 2 5 90 10 6 200 VSHUTDOWN ISHUTDOWN I11, I24 tOFF tr, t f Shutdown Pin Threshold Shutdown Pin Input Current CT Pin Discharge Current Off-Time (Note 3) Driver Output Transition Times 0V < VSHUTDOWN < 8V, V10, V24 = 16V VOUT in Regulation, VSENSE – = VOUT VOUT = 0V CT = 390pF, ILOAD = 700mA CL = 3000pF (Pins 6, 9, 20, 23), V10, V24 = 6V – 40°C ≤ TA ≤ 85°C (Note 4), V10 = V24 = 10V, unless otherwise noted. V2, V16 I2, I16 VOUT Feedback Voltage Feedback Current Regulated Output Voltage 3.3V Output 5V Output Input DC Supply Current (Note 2) Normal Mode Sleep Mode Shutdown Input DC Supply Current (Note 2) Normal Mode Sleep Mode Shutdown Input DC Supply Current (Note 2) Normal Mode Sleep Mode Shutdown V1 – V28 V15 – V14 Current Sense Threshold Voltage LTC1142HV-ADJ Only: V10, V24 = 9V LTC1142HV-ADJ Only LTC1142, LTC1142HV ILOAD = 700mA, V24 = 9V ILOAD = 700mA, V10 = 9V LTC1142 4V < V10, V24 < 12V 4V < V24 < 12V, 6V < V10 < 12V VSHUTDOWN = 2.1V, 4V < V10, V24 < 12V LTC1142HV-ADJ, LTC1142HV 4V < V10, V24 < 18V 4V < V24 < 18V, 6V < V10 < 18V VSHUTDOWN = 2.1V, 4V < V10, V24 < 12V LTC1142L-ADJ (Note 5) 3.5V < V10, V24 < 12V 3.5V < V10, V24 < 12V VSD1 = VSD2 = 2.1V, 3.5V < V10, V24 < 12V LTC1142HV-ADJ, LTC1142L-ADJ V14 = V28 = VOUT + 100mV, V2 = V16 = VREF + 25mV V14 = V28 = VOUT – 100mV, V2 = V16 = VREF – 25mV LTC1142, LTC1142HV V28 = VOUT + 100mV (Forced) V28 = VOUT – 100mV (Forced) LTC1142, LTC1142HV V14 = VOUT + 100mV (Forced) V14 = VOUT – 100mV (Forced) VSHUTDOWN tOFF Shutdown Pin Threshold Off-Time (Note 3) CT = 390pF, ILOAD = 700mA 3.17 4.85 1.21 1.25 0.2 3.33 5.05 1.6 160 10 1.6 160 10 1.6 160 10 25 150 25 150 25 150 0.8 5 1.29 1 3.40 5.20 2.4 260 22 2.6 280 24 2.4 260 22 V µA V V mA µA µA mA µA µA mA µA µA mV mV mV mV mV mV V µs I10, I24 130 170 125 175 125 0.55 3.8 175 2 6 3 LTC1142/LTC1142L/LTC1142HV ELECTRICAL CHARACTERISTICS The q denotes specifications which apply over the full operating temperature range. Note 1: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC1142CG: TJ = TA + (PD × 95°C/ W) Note 2: This current is for one regulator block. Total supply current is the sum of Pins 10 and 24 currents. Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See the Applications Information section. Note 3: In applications where RSENSE is placed at ground potential, the offtime increases approximately 40%. Note 4: The LTC1142/LTC1142HV-ADJ/LTC1142HV/LTC1142L-ADJ are not tested and quality-assurance sampled at – 40°C to 85°C. These specifications are guaranteed by design and/or correlation. Note 5: The LTC1142L-ADJ allows operation down to VIN = 3.5V. TYPICAL PERFOR A CE CHARACTERISTICS 5V Output Efficiency 100 VIN = 6V VIN = 5V 100 EFFICIENCY (%) VIN = 10V EFFICIENCY (%) EFFICIENCY (%) 95 90 85 0.01 0.1 LOAD CURRENT (A) 3.3V Efficiency vs Input Voltage 100 98 96 EFFICIENCY (%) FIGURE 1 CIRCUIT VOUT = 3.3V 94 ∆VOUT (mV) 92 90 88 86 84 82 80 0 4 ILOAD = 100mA 0 – 10 – 20 – 30 ∆VOUT (mV) ILOAD = 1A 12 8 INPUT VOLTAGE (V) 4 UW 1 2 1142 G01 3.3V Output Efficiency 100 98 96 94 92 90 88 86 84 82 85 0.01 5V Efficiency vs Input Voltage FIGURE 1 CIRCUIT VOUT = 5V ILOAD = 1A 95 VIN = 10V 90 ILOAD = 100mA 80 0.1 LOAD CURRENT (A) 1 2 0 4 12 8 INPUT VOLTAGE (V) 16 20 1142 G03 1142 G02 Line Regulation 40 30 20 10 –20 – 40 – 60 – 80 – 100 0 4 12 8 INPUT VOLTAGE (V) 16 20 1142 G05 Load Regulation 20 FIGURE 1 CIRCUIT RSENSE = 0.05Ω VIN = 6V VIN = 12V FIGURE 1 CIRCUIT ILOAD = 1A 0 VIN = 6V VIN = 12V VOUT = 5V VOUT = 3.3V 0 0.5 1.5 2.0 1.0 LOAD CURRENT (A) 2.5 1142 G06 16 20 1142 G04 – 40 LTC1142/LTC1142L/LTC1142HV TYPICAL PERFOR A CE CHARACTERISTICS DC Supply Current 2.1 1.8 SUPPLY CURRENT (mA) 20 18 16 SUPPLY CURRENT (µA) PER REGULATOR BLOCK PINS 10, 24 VSHUTDOWN = 2V NORMALIZED FREQUENCY 1.5 1.2 0.9 0.6 0.3 0 0 2 4 ACTIVE MODE PER REGULATOR BLOCK NOT INCLUDING GATE CHARGE CURRENT PINS 10, 24 SLEEP MODE 6 8 10 12 14 INPUT VOLTAGE (V) Gate Charge Supply Current 28 24 80 70 GATE CHARGE CURRENT (mA) 20 16 12 8 QN + QP = 100nC OFF-TIME (µs) SENSE VOLTAGE (mV) QN + QP = 50nC 4 0 20 80 200 260 140 OPERATING FREQUENCY (kHz) 1142 G10 PI FU CTIO S LTC1142/LTC1142HV SENSE + 3 (Pin 1): The (+) Input to the 3.3V Section Current Comparator. A built-in offset between Pins 1 and 28 in conjunction with RSENSE3 sets the current trip threshold for the 3.3V section. SHUTDOWN 3 (Pin 2): When grounded, the 3.3V section operates normally. Pulling Pin 2 high holds both MOSFETs off and puts the 3.3V section in micropower shutdown mode. Requires CMOS logic-level signal with tr, t f < 1µs. Do not “float” Pin 2. SGND3 (Pin 3): The 3.3V section small-signal ground must be routed separately from other grounds to the (–) terminal of the 3.3V section output capacitor. PGND3 (Pin 4): The 3.3V section driver power ground connects to source of N-channel MOSFET and the (–) terminal of the 3.3V section input capacitor. NC (Pin 5): No Connection. NDRIVE 3 (Pin 6): High Current Drive for Bottom N-Channel MOSFET, 3.3V Section. Voltage swing at Pin 6 is from ground to VIN3. UW 16 1142 G07 Supply Current in Shutdown 1.6 1.4 Operating Frequency vs VIN – VOUT VOUT = 5V 0°C 1.2 1.0 0.8 0.6 0.4 0.2 0 25°C 70°C 14 12 10 8 6 4 2 0 18 0 2 4 6 8 10 12 14 INPUT VOLTAGE (V) 16 18 0 2 4 6 8 10 12 1142 G09 VIN – VOUT VOLTAGE (V) 1142 G08 Off-Time vs Output Voltage 175 Current Sense Threshold Voltage 150 125 100 75 50 25 0 MINIMUM THRESHOLD MAXIMUM THRESHOLD VSENSE = VOUT 60 50 40 30 20 10 0 0 VOUT = 3.3V 1 3 4 2 OUTPUT VOLTAGE (V) 5 1142 G11 VOUT = 5V 0 20 60 40 TEMPERATURE (°C) 80 100 1142 G12 U U U 5 LTC1142/LTC1142L/LTC1142HV PI FU CTIO S NC (Pins 7, 8): No Connection. PDRIVE 5 (Pin 9): High Current Drive for Top P-Channel MOSFET, 5V Section. Voltage swing at this pin is from VIN5 to ground. VIN5 (Pin 10): Supply pin, 5V section, must be closely decoupled to 5V power ground Pin 18. CT5 (Pin 11): External capacitor CT5 from Pin 11 to ground sets the operating frequency for the 5V section. (The actual frequency is also dependent upon the input voltage.) INT VCC5 (Pin 12) : Internal supply voltage for the 5V section, nominally 3.3V, can be decoupled to signal ground, Pin 17. Do not externally load this pin. ITH5 (Pin 13): Gain Amplifier Decoupling Point, 5V Section. The 5V section current comparator threshold increases with the Pin 13 voltage. SENSE – 5 (Pin 14): Connects to internal resistive divider which sets the output voltage for the 5V section. Pin 14 is also the (–) input for the current comparator on the 5V section. SENSE + 5 (Pin 15): The (+) Input to the 5V Section Current Comparator. A built-in offset between Pins 15 and 14 in conjunction with RSENSE5 sets the current trip threshold for the 5V section. SHUTDOWN 5 (Pin 16): When grounded, the 5V section operates normally. Pulling Pin 16 high holds both MOSFETs off and puts the 5V section in micropower shutdown mode. Requires CMOS logic signal with tr, t f < 1µs. Do not “float” Pin 16. SGND5 (Pin 17): The 5V section small-signal ground must be routed separately from other grounds to the (–) terminal of the 5V section output capacitor. PGND5 (Pin 18): The 5V section driver power ground connects to source of N-channel MOSFET and the (–) terminal of the 5V section input capacitor. NC (Pin 19): No Connection. NDRIVE 5 (Pin 20): High Current Drive for Bottom N-Channel MOSFET, 5V Section. Voltage swing at Pin 20 is from ground to VIN5. NC (Pins 21, 22): No Connection. PDRIVE 3 (Pin 23): High Current Drive for Top P-Channel MOSFET, 3.3V Section. Voltage swing at this pin is from VIN3 to ground. VIN3 (Pin 24): Supply pin, 3.3V section, must be closely decoupled to 3.3V power ground, Pin 4. CT3 (Pin 25): External capacitor CT3 from Pin 25 to ground sets the operating frequency for the 3.3V section. (The actual frequency is also dependent upon the input voltage.) INT VCC3 (Pin 26): Internal supply voltage for the 3.3V section, nominally 3.3V, can be decoupled to signal ground, Pin 3. Do not externally load this pin. ITH3 (Pin 27): Gain Amplifier Decoupling Point, 3.3V Section. The 3.3V section current comparator threshold increases with the Pin 27 voltage. SENSE – 3 (Pin 28): Connects to internal resistive divider which sets the output voltage for the 3.3V section. Pin 28 is also the (–) input for the current comparator on the 3.3V section. 6 U U U LTC1142HV-ADJ/LTC1142L-ADJ SENSE + 1 (Pin 1): The (+) Input to the Section 1 Current Comparator. A built-in offset between Pins 1 and 28 in conjunction with RSENSE1 sets the current trip threshold for this section. VFB1 (Pin 2): This pin serves as the feedback pin from an external resistive divider used to set the output voltage for section 1. SHUTDOWN 1 (Pin 3): When grounded, the section 1 regulator operates normally. Pulling Pin 3 high holds both MOSFETs off and puts this section in micropower shutdown mode. Requires CMOS logic signal with tr, t f < 1µs. Do not “float” Pin 3. SGND1 (Pin 4): The section 1 small-signal ground must be routed separately from other grounds to the (–) terminal of the section 1 output capacitor. PGND1 (Pin 5): The section 1 driver power ground connects to source of N-channel MOSFET and the (–) terminal of the section 1 input capacitor. LTC1142/LTC1142L/LTC1142HV PI FU CTIO S NDRIVE 1 (Pin 6): High Current Drive for Bottom N-Channel MOSFET, Section 1. Voltage swing at Pin 6 is from ground to VIN1. NC (Pins 7, 8): No Connection. PDRIVE 2 (Pin 9): High Current Drive for Top P-Channel MOSFET, Section 2. Voltage swing at this pin is from VIN2 to ground. VIN2 (Pin 10): Supply pin, section 2, must be closely decoupled to section 2 power ground, Pin 19. CT2 (Pin 11): External capacitor CT2 from Pin 11 to ground sets the operating frequency for the section 2. (The actual frequency is also dependent upon the input voltage.) INT VCC2 (Pin 12) : Internal supply voltage for section 2, nominally 3.3V, can be decoupled to signal ground, Pin 18. Do not externally load this pin. ITH2 (Pin 13): Gain Amplifier Decoupling Point, Section 2. The section 2 current comparator threshold increases with the Pin 13 voltage. SENSE – 2 (Pin 14): Connects (–) input for the current comparator on section 2. SENSE + 2 (Pin 15): The (+) Input to the Section 2 Current Comparator. A built-in offset between Pins 15 and 14 in conjunction with RSENSE2 sets the current trip threshold for this section. VFB2 (Pin 16): This pin serves as the feedback pin from an external resistive divider used to set the output voltage for section 2. SHUTDOWN 2 (Pin 17): When grounded, the section 2 regulator operates normally. Pulling Pin 17 high holds both MOSFETs off and puts section 2 in micropower shutdown mode. Requires CMOS logic signal with tr, tf < 1µs. Do not “float” Pin 17. SGND2 (Pin 18): The section 2 small-signal ground must be routed separately from other grounds to the (–) terminal of the section 2 output capacitor. PGND2 (Pin 19): The section 2 driver power ground connects to source of the N-channel MOSFET and the (–) terminal of the section 2 input capacitor. NDRIVE 2 (Pin 20): High Current Drive for Bottom NChannel MOSFET, Section 2. Voltage swing at Pin 20 is from ground to VIN2. NC (Pins 21, 22): No Connection. PDRIVE 1 (Pin 23): High Current Drive for Top P-Channel MOSFET, Section 1. Voltage swing at this pin is from VIN1 to ground. VIN1 (Pin 24): Supply Pin, Section 1. Must be closely decoupled to section 1 power ground Pin 5. CT1 (Pin 25): External capacitor CT1 from Pin 25 to ground sets the operating frequency for section 1. (The actual frequency is also dependent upon the input voltage.) INT VCC1 (Pin 26): Internal supply voltage for section 1, nominally 3.3V, can be decoupled to signal ground, Pin 4. Do not externally load this pin. ITH1 (Pin 27): Gain Amplifier Decoupling Point, Section 1. The section 1 current comparator threshold increases with the Pin 27 voltage. SENSE – 1 (Pin 28): Connects to the (–) input for the current comparator on section 1. U U U 7 LTC1142/LTC1142L/LTC1142HV FU CTIO AL DIAGRA 2(16) LTC1142-ADJ 3(17) PIN NUMBERS FOR LTC1142, LTC1142HV PIN NUMBERS FOR LTC1142L-ADJ LTC1142HV-ADJ Only one regulator block shown. Pin numbers are for 3.3V (5V) sections for LTC1142/LTC1142HV, and VOUT1 (VOUT2) for LTC1142L-ADJ/LTC1142HV-ADJ. SGND 24(10) VIN 23(9) PDRIVE 3(17) LTC1142L-ADJ LTC1142HV-ADJ 4(18) SENSE + 1(15) SENSE – 28(14) SLEEP + S Q S VTH2 VTH1 – – + T 25(11) CT OFF-TIME CONTROL OPERATIO Refer to Functional Diagram The LTC1142 series consists of two individual regulator blocks, each using current mode, constant off-time architectures to synchronously switch an external pair of complementary power MOSFETs. The two regulators are internally set to provide output voltages of 3.3V and 5V for the LTC1142. The LTC1142HV-ADJ/LTC1142L-ADJ are configured to provide two user selectable output voltages, each set by external resistor dividers. Operating frequency is individually set on each section by the external capacitors at CT, Pins 11 and 25. The output voltage is sensed by an internal voltage divider connected to Sense –, Pin 28 (14) (LTC1142) or external divider returned to VFB, Pin 2 (16) (LTC1142-ADJ). A voltage comparator V and a gain block G compare the divided output voltage with a reference voltage of 1.25V. To optimize efficiency, the LTC1142 series automatically switches between two modes of operation, Burst Mode and continuous mode. The voltage comparator is the primary control element when the device is in Burst Mode operation, while the gain block controls the output voltage in continuous mode. 8 W 6(20) NDRIVE LTC1142L-ADJ LTC1142HV-ADJ 2(16) NC/ADJ 4(18) PGND LTC1142L-ADJ, LTC1142HV-ADJ: 5(19) V U U U – + R – C 25mV TO 150mV + ITH – + VOS 13k G 5pF – + 1.25V 100k INT VCC REFERENCE 26(12) 27(13) VIN SENSE – SHUTDOWN 2(16) LTC1142L-ADJ LTC1142HV-ADJ 3(17) 1142 BD During the switch “ON” cycle in continuous mode, current comparator C monitors the voltage between Pins 1 (15) and 28 (14) connected across an external shunt in series with the inductor. When the voltage across the shunt reaches its threshold value, the PDrive output is switched to VIN, turning off the P-channel MOSFET. The timing capacitor connected to Pin 25 (11) is now allowed to discharge at a rate determined by the off-time controller. The discharge current is made proportional to the output voltage [measured by Pin 28 (14)] to model the inductor current, which decays at a rate that is also proportional to the output voltage. While the timing capacitor is discharging, the NDrive output goes to VIN, turning on the N-channel MOSFET. When the voltage on the timing capacitor has discharged past VTH1, comparator T trips, setting the flip-flop. This causes the NDrive output to go low (turning off the N-channel MOSFET) and the PDrive output to also go low (turning the P-channel MOSFET back on). The cycle then repeats. LTC1142/LTC1142L/LTC1142HV OPERATIO As the load current increases, the output voltage decreases slightly. This causes the output of the gain stage [Pin 27(13)] to increase the current comparator threshold, thus tracking the load current. The sequence of events for Burst Mode operation is very similar to continuous operation with the cycle interrupted by the voltage comparator. When the output voltage is at or above the desired regulated value, the P-channel MOSFET is held off by comparator V and the timing capacitor continues to discharge below VTH1. When the timing capacitor discharges past VTH2, voltage comparator S trips, causing the internal sleep line to go low and the N-channel MOSFET to turn off. The circuit now enters sleep mode with both power MOSFETs turned off. In sleep mode a majority of the circuitry is turned off, dropping the quiescent current from 1.6mA to 160µA (for one regulator block). The load current is now being supplied from the output capacitor. When the output voltage has dropped by the amount of APPLICATIO S I FOR ATIO The basic LTC1142 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of RSENSE. Once RSENSE is known, CT and L can be chosen. Next, the power MOSFETs and D1 are selected. Finally, CIN and COUT are selected and the loop is compensated. Since the 3.3V and 5V sections in the LTC1142 are identical and similarly section 1 and section 2 in the LTC1142HV-ADJ/ LTC1142L-ADJ are identical, the process of component selection is the same for both sections. The circuit shown in Figure 1 can be configured for operation up to an input voltage of 20V. RSENSE Selection for Output Current RSENSE is chosen based on the required output current. The LTC1142 current comparators have a threshold range which extends from a minimum of 25mV/RSENSE to a maximum of 150mV/RSENSE. The current comparator threshold sets the peak of the inductor ripple current, U W UU U Refer to Functional Diagram hysteresis in comparator V, the P-channel MOSFET is again turned on and this process repeats. To avoid the operation of the current loop interfering with Burst Mode operation, a built-in offset VOS is incorporated in the gain stage. This prevents the current comparator threshold from increasing until the output voltage has dropped below a minimum threshold. To prevent both the external MOSFETs from ever being turned on at the same time, feedback is incorporated to sense the state of the driver output pins. Before the NDrive output can go high, the PDrive output must also be high. Likewise, the PDrive output is prevented from going low while the NDrive output is high. Using constant off-time architecture, the operating frequency is a function of the input voltage. To minimize the frequency variation as dropout is approached, the off-time controller increases the discharge current as VIN drops below VOUT + 1.5V. In dropout the P-channel MOSFET is turned on continuously (100% duty cycle) providing low dropout operation with VOUT ~ VIN. yielding a maximum output current IMAX equal to the peak value less half the peak-to-peak ripple current. For proper Burst Mode operation, IRIPPLE(P-P) must be less than or equal to the minimum current comparator threshold. Since efficiency generally increases with ripple current, the maximum allowable ripple current is assumed, i.e., IRIPPLE(P-P) = 25mV/RSENSE (see CT and L Selection for Operating Frequency section). Solving for RSENSE and allowing a margin for variations in the LTC1142 and external component values yields: RSENSE = 100mV IMAX A graph for Selecting RSENSE vs Maximum Output Current is given in Figure 2. The load current below which Burst Mode operation commences, IBURST, and the peak short-circuit current ISC(PK), 9 LTC1142/LTC1142L/LTC1142HV APPLICATIO S I FOR ATIO both track IMAX. Once RSENSE has been chosen, IBURST and ISC(PK) can be predicted from the following: 15mV IBURST ≈ RSENSE ISC(PK) = 150mV RSENSE The LTC1142 automatically extends tOFF during a short circuit to allow sufficient time for the inductor current to decay between switch cycles. The resulting ripple current causes the average short-circuit current ISC(AVG) to be reduced to approximately IMAX. 0.20 0.15 RSENSE (Ω) 0.10 0.05 0 0 1 3 4 2 MAXIMUM OUTPUT CURRENT (A) 5 1142 F02 Figure 2. Selecting RSENSE L and CT Selection for Operating Frequency Each regulator section of the LTC1142 uses a constant offtime architecture with tOFF determined by an external timing capacitor CT. Each time the P-channel MOSFET switch turns on, the voltage on CT is reset to approximately 3.3V. During the off-time, CT is discharged by a current which is proportional to VOUT. The voltage on CT is analogous to the current in inductor L, which likewise decays at a rate proportional to VOUT. Thus the inductor value must track the timing capacitor value. The value of CT is calculated from the desired continuous mode operating frequency: CAPACITANCE (pF) CT = 1 2.6 × 104 × f Assumes VIN = 2VOUT, Figure 1 circuit. 10 U A graph for selecting CT versus frequency including the effects of input voltage is given in Figure 3. As the operating frequency is increased the gate charge losses will be higher, reducing efficiency (see Efficiency Considerations section). The complete expression for operating frequency of the circuit in Figure 1 is given by: f= 1  VOUT  1− tOFF  VIN    W UU where: V  tOFF = 1.3 × 104 × CT ×  REG   VOUT  VREG is the desired output voltage (i.e., 5V, 3.3V). VOUT is the measured output voltage. Thus VREG / VOUT = 1 in regulation. Note that as VIN decreases, the frequency decreases. When the input-to-output voltage differential drops below 1.5V for a particular section, the LTC1142 reduces tOFF in that section by increasing the discharge current in CT. This prevents audible operation prior to dropout. 1000 VSENSE = VOUT = 5V 800 600 VIN = 12V 400 VIN = 7V 200 VIN = 10V 0 0 50 150 200 100 FREQUENCY (kHz) 250 300 1142 F03 Figure 3. Timing Capacitor Value Once the frequency has been set by CT, the inductor L must be chosen to provide no more than 25mV/RSENSE of peakto-peak inductor ripple current. This results in a minimum required inductor value of: LMIN = 5.1 × 105 × RSENSE × CT × VREG As the inductor value is increased from the minimum value, the ESR requirements for the output capacitor are LTC1142/LTC1142L/LTC1142HV APPLICATIO S I FOR ATIO eased at the expense of efficiency. If too small an inductor is used, the inductor current will decrease past zero and change polarity. A consequence of this is that the LTC1142 may not enter Burst Mode operation and efficiency will be severely degraded at low currents. Inductor Core Selection Once the minimum value for L is known, the type of inductor must be selected. The highest efficiency will be obtained using ferrite, molypermalloy (MPP), or Kool Mµ® cores. Lower cost powdered iron cores provide suitable performance, but cut efficiency by 3% to 7%. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple which can cause Burst Mode operation to be falsely triggered. Do not allow the core to saturate! Kool Mµ (from Magnetics, Inc.) is a very good, low loss core material for toroids with a “soft” saturation characteristic. Molypermalloy is slightly more efficient at high (>200kHz) switching frequencies, but it is quite a bit more expensive. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, new designs for surface mount are available from Coiltronics and Beckman Industrial Corporation which do not increase the height significantly. Power MOSFET and D1, D2 Selection Two external power MOSFETs must be selected for use with each section of the LTC1142: a P-channel MOSFET for the main switch, and an N-channel MOSFET for the synchronous switch. The main selection criteria for the power MOSFETs are the threshold voltage VGS(TH) and on- resistance RDS(ON). Kool Mµ is a registered trademark of Magnetics, Inc. U The minimum input voltage determines whether standard threshold or logic-level threshold MOSFETs must be used. For V IN > 8V, standard threshold MOSFETs (VGS(TH) < 4V) may be used. If VIN is expected to drop below 8V, logic-level threshold MOSFETs (VGS(TH) < 2.5V) are strongly recommended. When logic-level MOSFETs are used, the LTC1142 supply voltage must be less than the absolute maximum VGS ratings for the MOSFETs. The maximum output current IMAX determines the RDS(ON) requirement for the two MOSFETs. When the LTC1142 is operating in continuous mode, the simplifying assumption can be made that one of the two MOSFETs is always conducting the average load current. The duty cycles for the two MOSFETs are given by: P-Ch Duty Cycle = N-Ch Duty Cycle = VOUT VIN VIN − VOUT VIN From the duty cycles the required R DS(ON) for each MOSFET can be derived: W UU P-Ch RDS(ON) = VIN × PP 2 VOUT × IMAX × 1 + δP VIN × PN ( ) N-Ch RDS(ON) = (VIN − VOUT) × IMAX2 × (1 + δN) where PP and PN are the allowable power dissipations and δP and δN are the temperature dependencies of RDS(ON). PP and PN will be determined by efficiency and/or thermal requirements (see Efficiency Considerations). (1 + δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs Temperature curve, but δ = 0.007/°C can be used as an approximation for low voltage MOSFETs. The Schottky diodes D1 and D2 shown in Figure 1 only conduct during the dead-time between the conduction of the respective power MOSFETs. The sole purpose of D1 and D2 is to prevent the body diode of the N-channel MOSFET from turning on and storing charge during the 11 LTC1142/LTC1142L/LTC1142HV APPLICATIO S I FOR ATIO dead-time, which could cost as much as 1% in efficiency (although there are no other harmful effects if D1 and D2 are omitted). Therefore, D1 and D2 should be selected for a forward voltage of less than 0.6V when conducting IMAX. CIN and COUT Selection In continuous mode, the source current of the P-channel MOSFET is a square wave of duty cycle VOUT/ VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN Required IRMS ≈ IMAX [( VOUT VIN − VOUT VIN )] 1/ 2 This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst case conditon is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. An additional 0.1µF to 1µF ceramic capacitor is also required on each VIN line (Pins 10 and 24) for high frequency decoupling. The selection of COUT is driven by the required Effective Series Resistance (ESR). The ESR of COUT must be less than twice the value of RSENSE for proper operation of the LTC1142: COUT Required ESR < 2RSENSE Optimum efficiency is obtained by making the ESR equal to RSENSE. As the ESR is increased up to 2RSENSE, the efficiency degrades by less than 1%. If the ESR is greater than 2RSENSE, the voltage ripple on the output capacitor will prematurely trigger Burst Mode operation, resulting in disruption of continuous mode and an efficiency hit which can be several percent. Manufacturers such as Nichicon and United Chemicon should be considered for high performance capacitors. The OS-CON semiconductor dielectric capacitor available OUTPUT CAPACITANCE (µF) 12 U from Sanyo has the lowest ESR/size ratio of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications multiple capacitors may have to be parallel to meet the capacitance, ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. For example, if 200µF/10V is called for in an application requiring 3mm height, two AVX 100µF/10V (P/N TPSD 107K010) could be used. Consult the manufacturer for other specific recommendations. At low supply voltages, a minimum capacitance at COUT is needed to prevent an abnormal low frequency operating mode (see Figure 4). When COUT is made too small, the output ripple at low frequencies will be large enough to trip the voltage comparator. This causes Burst Mode operation to be activated when the LTC1142 would normally be in continuous operation. The output remains in regulation at all times. 1000 L = 50µH RSENSE = 0.02Ω L = 25µH RSENSE = 0.02Ω 800 600 400 L = 50µH RSENSE = 0.05Ω 200 0 0 1 3 4 2 VIN – VOUT VOLTAGE (V) 5 1142 F04 W UU Figure 4. Minimum Value of COUT Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in DC (resistive) load LTC1142/LTC1142L/LTC1142HV APPLICATIO S I FOR ATIO current. When a load step occurs, VOUT shifts by an amount equal to ∆ILOAD × ESR, where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT until the regulator loop adapts to the current change and returns VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing which would indicate a stability problem. The Pin 27 (13) external components shown in the Figure 1 circuit will prove adequate compensation for most applications. A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately 25 × CLOAD. Thus a 10µF capacitor would require a 250µs rise time, limiting the charging current to about 200mA. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc., are the individual losses as a percentage of input power. (For high efficiency circuits only small errors are incurred by expressing losses as a percentage of output power.) Although all dissipative elements in the circuit produce losses, three main sources usually account for most of the losses in LTC1142 circuits: 1. LTC1142 DC bias current 2. MOSFET gate charge current 3. I2R losses 1. The DC supply current is the current which flows into VIN (pin 24 for the 3.3V section, Pin 10 for the 5V U section) less the gate charge current. For VIN = 10V the LTC1142 DC supply current for each section is 160µA with no load, and increases proportionally with load up to a constant 1.6mA after the LTC1142 has entered continuous mode. Because the DC bias current is drawn from VIN, the resulting loss increases with input voltage. For VIN = 10V the DC bias losses are generally less than 1% for load currents over 30mA. However, at very low load currents the DC bias current accounts for nearly all of the loss. 2. MOSFET gate charge current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN which is typically much larger than the DC supply current. In continuous mode, IGATE(CHG) = f (QN + QP). The typical gate charge for a 0.1Ω N-channel power MOSFET is 25nC, and for a P-channel about twice that value. This results in IGATE(CHG) = 7.5mA in 100kHz continuous operation, for a 2% to 3% typical mid-current loss with VIN = 10V. Note that the gate charge loss increases directly with both input voltage and operating frequency. This is the principal reason why the highest efficiency circuits operate at moderate frequencies. Furthermore, it argues against using larger MOSFETs than necessary to control I2R losses, since overkill can cost efficiency as well as money! 3. I2R losses are easily predicted from the DC resistances of the MOSFET, inductor, and current shunt. In continuous mode the average output current flows through L and RSENSE, but is “chopped” between the P-channel and N-channel MOSFETs. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L and RSENSE to obtain I2R losses. For example, if each RDS(ON) = 0.1Ω, RL = 0.15Ω, and RSENSE = 0.05Ω, then the total resistance is 0.3Ω. This results in losses ranging from 3% to 12% as the output current increases from 0.5A to 2A. I2R losses cause the efficiency to roll off at high output currents. W UU 13 LTC1142/LTC1142L/LTC1142HV APPLICATIO S I FOR ATIO Figure 5 shows how the efficiency losses in one section of a typical LTC1142 regulator end up being apportioned. The gate charge loss is responsible for the majority of the efficiency lost in the mid-current region. If Burst Mode operation was not employed at low currents, the gate charge loss alone would cause efficiency to drop to unacceptable levels. With Burst Mode operation, the DC supply current represents the lone (and unavoidable) loss component which continues to become a higher percentage as output current is reduced. As expected, the I2R losses dominate at high load currents. Other losses including CIN and COUT ESR dissipative losses, MOSFET switching losses, Schottky conduction losses during dead-time and inductor core losses, generally account for less than 2% total additional loss. 100 I2R GATE CHARGE 95 1/2 LTC1142 IQ 90 EFFICIENCY/LOSS (%) 85 80 0.01 0.03 0.3 1 0.1 OUTPUT CURRENT (A) 3 1142 F05 Figure 5. Efficiency Loss Design Example As a design example, assume VIN = 12V (nominal), 5V section, IMAX = 2A and f = 200kHz; RSENSE, CT and L can immediately be calculated: RSENSE = 100mV/2 = 0.05Ω tOFF = (1/200kHz) × [1 – (5/12)] = 2.92µs CT5 = 2.92µs/(1.3 × 104) = 220pF L2MIN = 5.1 × 105 × 0.05Ω × 220pF × 5V = 28µH Assume that the MOSFET dissipations are to be limited to PN = PP = 250mW. If TA = 50°C and the thermal resistance of each MOSFET is 50°C/ W, then the junction temperatures will be 63°C 14 U and δP = δN = 0.007(63 – 25) = 0.27. The required RDS(ON) for each MOSFET can now be calculated: W UU P - Ch RDS(ON) = N - Ch RDS(ON) = 12(0.25) 5(2)2 (1.27) 12(0.25) 5(2) (1.27) 2 = 0.12Ω = 0.085Ω The P-channel requirement can be met by a Si9430DY, while the N-channel requirement is exceeded by a Si9410DY. Note that the most stringent requirement for the N-channel MOSFET is with VOUT = 0 (i.e., short circuit). During a continuous short circuit, the worst case N-channel dissipation rises to: PN = ISC(AVG)2 × RDS(ON) × (1 + δN) With the 0.05Ω sense resistor, ISC(AVG) = 2A will result, increasing the 0.085Ω N-channel dissipation to 450mW at a die temperature of 73°C. CIN will require an RMS current rating of at least 1A at temperature, and COUT will require an ESR of 0.05Ω for optimum efficiency. Now allow VIN to drop to its minimum value. At lower input voltages the operating frequency will decrease and the P-channel will be conducting most of the time, causing its power dissipation to increase. At VIN(MIN) = 7V: fMIN = (1/2.92µs)[1 – (5V/ 7V)] = 98kHz 5V(0.12Ω)(2A)2 (1.27) PP = = 435mV 7V A similar calculation for the 3.3V section results in the component values shown in Figure 14. LTC1142HV-ADJ/LTC1142L-ADJ Adjustable Applications When an output voltage other than 3.3V or 5V is required, the LTC1142 adjustable version is used with an external resistive divider from VOUT to VFB, Pin 2 (16). The regulated output voltage is determined by:  R2  VOUT = 1.25 1 +   R1 LTC1142/LTC1142L/LTC1142HV APPLICATIO S I FOR ATIO To prevent stray pickup a 100pF capacitor is suggested across R1 located close to the LTC1142HV-ADJ/LTC1142LADJ as in Figure 6. The external divider network must be placed across COUT with the negative plate of COUT returned to signal ground. Refer to the Board Layout Checklist. RSENSE VFB [PIN 2(16)] 100pF SGND [PIN 4(18)] 1142 F06 R2 + R1 VOUT COUT Figure 6. LTC1142-ADJ External Feedback Network Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the 1000pF + RSENSE3 VOUT3 + COUT3 SHUTDOWN (3.3V OUTPUT) 1 2 SENSE +3 SHUTDOWN 3 SENSE –3 ITH3 INT VCC3 CT3 VIN3 28 3300pF 27 26 25 VIN3 CT3 1k – 3 SGND3 L1 4 PGND3 + 5 NC 6 N-CH NDRIVE 3 7 NC 8 NC P-CH 1µF VIN5 LTC1142 – D1 9 PDRIVE 5 VIN3 + CT5 1k 3300pF 13 14 BOLD LINES INDICATE HIGH CURRENT PATHS 1000pF 1142 F07 Figure 7. LTC1142 Layout Diagram (see Board Layout Checklist) + + CIN3 10 11 12 U LTC1142. These items are also illustrated graphically in the layout diagram of Figure 7. In general each block should be self-contained with little cross coupling for best performance. Check the following in your layout: 1. Are the signal and power grounds segregated? The LTC1142 signal ground [Pin 3 (17) for the LTC1142, Pin 4 (18) for LTC1142-ADJ] must return to the (–) plate of COUT. The power ground returns to the source of the N-channel MOSFET, anode of the Schottky diode, and (–) plate of CIN, which should have as short lead lengths as possible. 2. Does the LTC1142 Sense – , Pin 28 (14) connect to a point close to RSENSE and the (+) plate of COUT? 3. Are the Sense – and Sense + leads routed together with minimum PC trace spacing? The 1000pF capacitor + 24 CIN5 PDRIVE 3 23 NC 22 NC 21 20 N-CH W UU + P-CH VIN5 1µF D2 – NDRIVE 5 VIN5 CT5 INT VCC5 ITH5 SENSE – 5 NC 19 PGND5 SGND5 SHUTDOWN 5 18 17 16 SHUTDOWN (5V OUTPUT) VOUT5 COUT5 RSENSE5 L2 + – 15 SENSE +5 + 15 LTC1142/LTC1142L/LTC1142HV APPLICATIO S I FOR ATIO between Pins 1 (15) and 28 (14) should be as close as possible to the LTC1142. 4. Does the (+) plate of CIN connect to the source of the P-channel MOSFET as closely as possible? This capacitor provides the AC current to the P-channel MOSFET. 5. Is the input decoupling capacitor (1µF/0.22µF) connected closely between Pin 24 (10) and power ground [Pin 4 (18) for the LTC1142, Pin 5 (19) for the LTC1142ADJ]? This capacitor carries the MOSFET driver peak currents. 6. Are the shutdown Pins 2 and 16 for the LTC1142 (Pins 3 and 17 for the LTC1142-ADJ) actively pulled to ground during normal operation? Both Shutdown pins are high impedance and must not be allowed to float. Both pins can be driven by the same external signal if needed. 7. For the LTC1142-ADJ adjustable applications, the resistive divider R1, R2 must be connected between the (+) plate of COUT and signal ground. Output Crowbar An added feature to using an N-channel MOSFET as the synchronous switch is the ability to crowbar the output with the same MOSFET. Pulling the CT , Pin 25 (11) above 1.5V when the output voltage is greater than the desired regulated value will turn “on” the N-channel MOSFET for that regulator section. A fault condition which causes the output voltage to go above a maximum allowable value can be detected by external circuitry. Turning on the N-channel MOSFET when this fault is detected will cause large currents to flow and blow the system fuse. The N-channel MOSFET needs to be sized so it will safely handle this overcurrent condition. The typical delay from pulling the CT pin high and the NDrive Pin 6 (20) going high is 250ns. Note: Under shutdown conditions, the N-channel is held OFF and pulling the CT pin high will not cause the N-channel MOSFET to crowbar the output. A simple N-channel FET can be used as an interface between the overvoltage detect circuitry and the LTC1142 as shown in Figure 8. 16 U PIN 26(12) FROM CROWBAR DETECT CIRCUIT (ACTIVE WHEN VGATE = VIN OFF WHEN VGATE = GND) VN2222LL PIN 25(11) INT VCC LTC1142 CT 1142 F08 W UU Figure 8. Output Crowbar Interface Troubleshooting Hints Since efficiency is critical to LTC1142 applications, it is very important to verify that the circuit is functioning correctly in both continuous and Burst Mode operation. The waveform to monitor is the voltage on the CT, Pins 25 and 11. In continuous mode (ILOAD > IBURST) the voltage on the CT pin should be a sawtooth with a 0.9VP-P swing. This voltage should never dip below 2V as shown in Figure 9a. When load currents are low (ILOAD < IBURST) Burst Mode operation occurs. The voltage on the CT pin now falls to ground for periods of time as shown in Figure 9b. 3.3V 0V (a) CONTINUOUS MODE OPERATION 3.3V 0V (b) Burst Mode OPERATION 1142 F09 Figure 9. CT Waveforms Inductor current should also be monitored. Look to verify that the peak-to-peak ripple current in continuous mode operation is approximately the same as in Burst Mode operation. If Pin 25 or Pin 11 is observed falling to ground at high output currents, it indicates poor decoupling or improper grounding. Refer to the Board Layout Checklist. Auxiliary Windings––Suppressing Burst Mode Operation The LTC1142 synchronous switch removes the normal limitation that power must be drawn from the inductor primary winding in order to extract power from auxiliary windings. With synchronous switching, auxiliary outputs may be loaded without regard to the primary output load, LTC1142/LTC1142L/LTC1142HV APPLICATIO S I FOR ATIO providing that the loop remains in continuous mode operation. Burst Mode operation can be suppressed at low output currents with a simple external network which cancels the 25mV minimum current comparator threshold. This technique is also useful for eliminating audible noise from certain types of inductors in high current (IOUT > 5A) applications when they are lightly loaded. An external offset is put in series with the Sense – pin to subtract from the built-in 25mV offset. An example of this technique is shown in Figure 10. Two 100Ω resistors are inserted in series with the sense leads from the sense resistor. SENSE + [PIN 1(15)] 1000pF SENSE – [PIN 28(14)] R3 R2 100Ω R1 100Ω RSENSE VOUT + COUT 1142 F10 Figure 10. Suppression of Burst Mode Operation TYPICAL APPLICATIONS N VIN 5.2V TO 18V + CIN1 22µF 35V ×2 L1 25µH 0.22µF P-CH Si9430DY 23 1 1000pF 24 VIN1 PDRIVE 1 SENSE + 1 SENSE – 1 VFB1 NDRIVE 1 3 VOUT1 3.6V/2A RSENSE1 0.05Ω SHUTDOWN 1 COUT1 220µF 10V ×2 + D1 MBRS130T3 N-CH Si9410DY 28 2 6 R2 100k 1% R1 52.3k 1% PGND1 SGND1 CT1 5 4 25 100pF RSENSE1, RSENSE2 : KRL SL-1/2-R050J L1: COILTRONICS CTX25-4 L2: COILTRONICS CTX33-5 Figure 11. LTC1142HV-ADJ Dual Regulator with 3.6V/2A and 5V/2A Outputs U With the addition of R3 a current is generated through R1 causing an offset of:  R1  VOFFSET = VOUT ×    R1 + R3  W U UU If VOFFSET > 25mV, the built-in offset will be cancelled and Burst Mode operation is prevented from occurring. Since VOFFSET is constant, the maximum load current is also decreased by the same offset. Thus, to get back to the same IMAX, the value of the sense resistor must be lower: R SENSE ≈ 75mV I MAX To prevent noise spikes from erroneously tripping the current comparator, a 1000pF capacitor is needed across Pins 1 (15) and Pins 28 (14). 0V = NORMAL >1.5V = SHUTDOWN 17 SHUTDOWN 2 10 VIN2 9 PDRIVE 2 SENSE + 2 LTC1142HV-ADJ SENSE – 2 VFB2 NDRIVE 2 ITH1 27 RC1 1k ITH2 13 RC2 1k CT2 11 SGND2 PGND2 18 19 15 0.22µF P-CH Si9430DY L2 33µH + CIN2 22µF 35V ×2 RSENSE2 0.05Ω VOUT2 5V/2A 1000pF 14 16 20 N-CH Si9410DY D2 MBRS130T3 + R4 150k 1% R3 49.9k 1% COUT2 220µF 10V ×2 CC1 CT1 CT2 CC2 270pF 3300pF 3300pF 270pF 100pF 1142 F11 17 LTC1142/LTC1142L/LTC1142HV TYPICAL APPLICATIONS N VIN 4.5V TO 18V + CIN1 22µF 35V ×2 L1 33µH 0.22µF P-CH Si9430DY 23 1 1000pF 24 VIN1 PDRIVE 1 SENSE + 1 SENSE – 1 VFB1 NDRIVE 1 PGND1 SGND1 CT1 5 4 3 VOUT1 2.5V/1.5A RSENSE1 0.075Ω COUT1 220µF 10V ×2 + D1 MBRS130T3 N-CH Si9410DY R2 49.9k 1% R1 49.9k 1% 100pF RSENSE1: KRL SL-C1-1/2-1R075J RSENSE2: KRL SL-C1-1/2-1R050J Figure 12. LTC1142HV-ADJ High Efficiency Regulator with 3.3V/2A and 2.5V/1.5A Outputs CIN3 22µF 25V ×2 + 0.22µF P-CH Si9433DY 24 VIN3 23 PDRIVE 3 1 1000pF 28 SENSE – 3 NDRIVE 3 SENSE + 3 VOUT3 3.3V/3A RSENSE3 0.033Ω L1 10µH D1 MBRS130T3 + COUT3 100µF 10V ×3 N-CH Si9410DY RSENSE3: KRL SL-C1-1/2-R033J RSENSE5: KRL SL-C1-1/2-R050J L1: COILTRONICS CTX10-4 L2: COILTRONICS CTX20-4 Figure 13. LTC1142HV High Efficiency Regulator with 3.3V/3A and 5V/2A Outputs 18 U 6 0V = NORMAL >1.5V = SHUTDOWN 17 SHUTDOWN 2 10 VIN2 9 PDRIVE 2 SENSE + 2 LTC1142HV-ADJ SENSE – 2 VFB2 NDRIVE 2 ITH1 27 RC1 1k ITH2 13 RC2 1k CT2 11 SGND2 PGND2 18 19 15 SHUTDOWN 1 0.22µF P-CH Si9430DY L2 25µH + CIN2 22µF 35V ×2 RSENSE2 0.05Ω VOUT2 3.3V/2A 1000pF 28 2 6 14 16 20 N-CH Si9410DY D2 MBRS130T3 + R4 84.5k 1% R3 51k 1% COUT2 220µF 10V ×2 25 CC1 CT1 CT2 CC2 330pF 3300pF 3300pF 330pF L1: COILTRONICS CTX33-4 L2: COILTRONICS CTX25-4 100pF 1142 F12 VIN 5.2V TO 8V 0V = NORMAL >1.5V = SHUTDOWN 2 SHUTDOWN 3 0.22µF 16 10 VIN5 PDRIVE 5 SENSE + 5 9 15 1000pF SENSE – 5 NDRIVE 5 14 20 N-CH Si9410DY P-CH Si9430DY + CIN5 22µF 25V ×2 SHUTDOWN 5 L2 20µH RSENSE5 0.05Ω VOUT5 5V/2A LTC1142HV D2 MBRS130T3 PGND3 SGND3 CT3 4 3 25 ITH3 ITH5 13 RC5 1k CT5 11 SGND5 PGND5 17 18 27 RC3 510Ω + COUT5 220µF 10V ×2 CT3 CC3 CT5 CC5 200pF 3300pF 3300pF 150pF 1142 F13 LTC1142/LTC1142L/LTC1142HV TYPICAL APPLICATIONS N VIN 6.5V TO 14V 22µF 25V ×2 + 1µF P-CH Si9430DY VOUT3 3.3V/2A RSENSE3 0.05Ω L1 33µH 23 1 0.01µF 28 D1 MBRS140T3 + 100µF 10V ×2 N-CH Si9410DY 12V ENABLE 0V = 12V OFF >3V = 12V ON (6V MAX) 12V/150mA 22µF 25V SHUTDOWN ADJ LT1121 VIN GND 3 1000pF 8 1142 F14 RSENSE3: KRL SL-C1-1/2-0R050J RSENSE5: KRL SL-C1-1/2-0R040J L1: COILTRONICS CTX33-4 T1: DALE LPE-6562-A026 R4 294k 1% PRIMARY: SECONDARY = 1:1.8 Figure 14. LTC1142 Triple Output Regulator with Switched 12V Output VIN 8V TO 18V FROM WALL ADAPTER 0V = CHARGE ON >1.5V = CHARGE OFF 0V = OUTPUT ON >1.5V = 3.3V OUTPUT OFF + D3 MBRS340T3 L1 50µH + CIN1 22µF 35V ×2 0.22µF P-CH Si9430DY 23 1 1000pF 24 VIN1 PDRIVE 1 SENSE + 1 SENSE – 1 VFB1 NDRIVE 1 PGND1 SGND1 CT1 5 4 25 ITH1 27 RC1 1k ITH2 13 RC2 1k CT2 CC2 3300pF 330pF 100pF CT2 11 LTC1142HV-ADJ 3 SHUTDOWN 1 17 SHUTDOWN 2 10 VIN2 9 PDRIVE 2 SENSE + 2 SENSE – 2 VFB2 NDRIVE 2 SGND2 PGND2 18 19 15 1000pF 28 2 14 16 20 N-CH Si9410DY D2 MBRS140T3 0.22µF P-CH Si9433DY L2 25µH CIN2 22µF 25V ×2 RSENSE1 0.1Ω RSENSE2 0.05Ω COUT1 220µF 10V + D1 MBRS140T3 N-CH Si9410DY 6 R2 274k 1% R1 49.9k 1% R4 84.5k 1% R3 51k 1% 100pF CT1 200pF VN2222LL CC1 3300pF RSENSE1: KRL SL-C1-1/2-1R100J RSENSE2: KRL SL-C1-1/2-1R050J L1: COILTRONICS CTX50-4 L2: COILTRONICS CTX25-4 “1” FOR TRICKLE CHARGE RX 51Ω FAST CHARGE = 130mV/RSENSE1 = 1.3A TRICKLE CHARGE = 130mV/RSENSE1 = 100mA Figure 15. LTC1142HV-ADJ High Efficiency Power Supply Providing 3.3V/2A with Built-In Battery Charger Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. + U 6 + 24 VIN3 PDRIVE 3 SENSE + 3 2 0V = NORMAL >1.5V = SHUTDOWN 16 SHUTDOWN 5 SHUTDOWN 3 VIN5 + 10 PDRIVE 5 9 1µF P-CH Si9430DY + 22µF 25V ×2 T1 30µH 100Ω RSENSE5 0.04Ω 15 SENSE + 5 LTC1142 1000pF SENSE 5 NDRIVE 5 ITH3 ITH5 CT5 SGND5 PGND5 17 18 – VOUT5 5V/2A SENSE – R1 100Ω R5 18k 3 14 20 N-CH Si9410DY D2 MBRS140T3 NDRIVE 3 PGND3 SGND3 CT3 4 3 25 27 RC3 510Ω 13 11 RC5 510Ω 220µF 10V ×2 + VN2222LL CT3 CT5 CC3 CC5 390pF 3300pF 3300pF 200pF + 20pF 1 R3 649k 1% 2 VOUT 5 D3 MBRS140T3 22Ω C9 22µF 35V VBATT 4 CELLS NiCAD VOUT2 3.3V/2A + COUT2 220µF 10V ×2 1142 F15 19 LTC1142/LTC1142L/LTC1142HV TYPICAL APPLICATIONS N 1400 1200 OUTPUT CURRENT (mA) Figure 16. LTC1142HV-ADJ Output Current vs Trickle Charge Set Resistance (RX) for the Circuit in Figure 15 Using a 0.1Ω Current Sense Resistor RSENSE1 Note: For additional high efficiency circuits, see Application Note 54. PACKAGE DESCRIPTIO 0.205 – 0.212** (5.20 – 5.38) 0° – 8° 0.301 – 0.311 (7.65 – 7.90) 0.002 – 0.008 (0.05 – 0.21) 0.005 – 0.009 (0.13 – 0.22) 0.010 – 0.015 *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH (0.25 – 0.38) SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.022 – 0.037 (0.55 – 0.95) 0.0256 (0.65) BSC RELATED PARTS PART NUMBER LTC1143 LTC1147 LTC1148 LTC1149 LTC1159 LTC1174 LTC1265 LTC1266 LTC1267 LTC1574 DESCRIPTION Dual Step-Down Switching Regulator Controller Step-Down Switching Regulator Controller Step-Down Switching Regulator Controller Step-Down Switching Regulator Controller Step-Down Switching Regulator Controller Step-Down Switching Regulator with Internal 0.5A Switch Step-Down Switching Regulator with Internal 1.2A Switch Step-Up/Down Switching Regulator Controller Dual High Efficiency Synchronous Switching Regulator Step-Down Switching Regulator with Internal 0.5A Switch and Schottky Diode COMMENTS Dual Version of LTC1147 Nonsynchronous, 8-Pin, VIN ≤ 16V Synchronous, VIN ≤ 20V Synchronous, VIN ≤ 48V, for Standard Threshold FETs Synchronous, VIN ≤ 40V, for Logic Level FETs VIN ≤ 18.5V, Comparator/Low Battery Detector VIN ≤ 13V, Comparator/Low Battery Detector Synchronous N- or P-Channel FETs, Comparator/Low Battery Detector VIN to 40V VIN ≤ 18.5V, Comparator 20 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 q FAX: (408) 434-0507 q TELEX: 499-3977 U U 1000 800 600 400 200 0 0 1 2 3 SET RESISTANCE (kΩ) 4 1142 F16 Dimensions in inches (millimeters) unless otherwise noted. G Package 28-Lead Plastic SSOP (0.209) (LTC DWG # 05-08-1640) 0.068 – 0.078 (1.73 – 1.99) 0.397 – 0.407* (10.07 – 10.33) 28 27 26 25 24 23 22 21 20 19 18 17 16 15 1 2 3 4 5 6 7 8 9 10 11 12 13 14 G28 SSOP 0694 LT/GP 1196 5K REV C • PRINTED IN USA © LINEAR TECHNOLOGY CORPORATION 1995