LTC1474CMS8-5

LTC1474/LTC1475 Low Quiescent Current High Efficiency Step-Down Converters DESCRIPTION The LT C®1474/LTC1475 series are high efficiency stepdown converters with internal P-channel MOSFET power switches that draw only 10µA typical DC supply current at no load while maintaining output voltage. The LTC1474 uses logic-controlled shutdown while the LTC1475 features pushbutton on/off. The low supply current coupled with Burst ModeTM operation enables the LTC1474/LTC1475 to maintain high efficiency over a wide range of loads. These features, along with their capability of 100% duty cycle for low dropout and wide input supply range, make the LTC1474/LTC1475 ideal for moderate current (up to 300mA) battery-powered applications. The peak switch current is user-programmable with an optional sense resistor (defaults to 325mA minimum if not used) providing a simple means for optimizing the design for lower current applications. The peak current control also provides short-circuit protection and excellent startup behavior. A low-battery detector that remains functional in shutdown is provided . The LTC1474/LTC1475 series availability in 8-lead MSOP and SO packages and need for few additional components provide for a minimum area solution. , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. FEATURES s s s s s s s s s s s s s High Efficiency: Over 92% Possible Very Low Standby Current: 10µA Typ Available in Space Saving 8-Lead MSOP Package Internal 1.4Ω Power Switch (VIN = 10V) Wide VIN Range: 3V to 18V Operation Very Low Dropout Operation: 100% Duty Cycle Low-Battery Detector Functional During Shutdown Programmable Current Limit with Optional Current Sense Resistor (10mA to 400mA Typ) Short-Circuit Protection Few External Components Required Active Low Micropower Shutdown: IQ = 6µA Typ Pushbutton On/Off (LTC1475 Only) 3.3V, 5V and Adjustable Output Versions APPLICATIONS s s s s s s s Cellular Telephones and Wireless Modems 4mA to 20mA Current Loop Step-Down Converter Portable Instruments Battery-Operated Digital Devices Battery Chargers Inverting Converters Intrinsic Safety Applications TYPICAL APPLICATION 100 VIN 4V TO 18V LOW BATTERY OUT + 10µF 25V 0.1µF 6 7 VIN 1 SENSE VFB LTC1474-3.3 3 2 LBI LBO 8 RUN SW 5 D1 MBR0530 1474/75 F01 VOUT 3.3V AT 250mA L1 100µH EFFICIENCY (%) LOW BATTERY IN RUN SHDN 100k + 100µF 6.3V GND 4 L1 = SUMIDA CDRH74-101 Figure 1. High Efficiency Step-Down Converter U U U LTC1474 Efficiency 90 VIN = 5V VIN = 10V 80 VIN = 15V 70 60 L = 100µH VOUT = 3.3V RSENSE = 0Ω 0.03 0.3 30 3 LOAD CURRENT (mA) 300 1474/75 TA01 50 1 LTC1474/LTC1475 ABSOLUTE MAXIMUM RATINGS Input Supply Voltage (VIN).........................– 0.3V to 20V Switch Current (SW, SENSE) .............................. 750mA Switch Voltage (SW).............. (VIN – 20V) to (VIN + 0.3V) VFB (Adjustable Versions) ..........................– 0.3V to 12V VOUT (Fixed Versions) ................................ –0.3V to 20V LBI, LBO ....................................................– 0.3V to 20V RUN, SENSE .................................. – 0.3V to (VIN + 0.3V) Operating Ambient Temperature Range Commercial ............................................ 0°C to 70°C Industrial ............................................ – 40°C to 85°C Junction Temperature (Note 1) ............................ 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C PACKAGE/ORDER INFORMATION TOP VIEW TOP VIEW VOUT/VFB LBO LBI GND 1 2 3 4 8 7 6 5 RUN VIN SENSE SW VOUT/VFB LBO LBI/OFF GND 1 2 3 4 TOP VIEW 8 7 6 5 ON VIN SENSE SW MS8 PACKAGE 8-LEAD PLASTIC MSOP MS8 PACKAGE 8-LEAD PLASTIC MSOP TJMAX = 125°C, θJA = 150°C/ W TJMAX = 125°C, θJA = 150°C/ W ORDER PART NUMBER LTC1474CMS8 LTC1474CMS8-3.3 LTC1474CMS8-5 ORDER PART NUMBER LTC1475CMS8 LTC1475CMS8-3.3 LTC1475CMS8-5 MS8 PART MARKING LTBW LTCR LTCS MS8 PART MARKING LTBK LTCP LTCQ Consult factory for Military grade parts. 2 U U W WW U W TOP VIEW 8 RUN 7 VIN 6 SENSE 5 SW VOUT/VFB 1 LBO 2 LBI/OFF 3 GND 4 8 ON 7 VIN 6 SENSE 5 SW VOUT/VFB 1 LBO 2 LBI 3 GND 4 S8 PACKAGE 8-LEAD PLASTIC SO S8 PACKAGE 8-LEAD PLASTIC SO TJMAX = 125°C, θJA = 110°C/ W TJMAX = 125°C, θJA = 110°C/ W ORDER PART NUMBER LTC1474CS8 LTC1474IS8 LTC1474CS8-3.3 LTC1474CS8-5 LTC1474IS8-3.3 LTC1474IS8-5 S8 PART MARKING 1474 1474I 14743 14745 14743I 14745I ORDER PART NUMBER LTC1475CS8 LTC1475IS8 LTC1475CS8-3.3 LTC1475CS8-5 S8 PART MARKING 1475 1475I 14753 14755 LTC1474/LTC1475 ELECTRICAL CHARACTERISTICS SYMBOL VFB VOUT PARAMETER Feedback Voltage LTC1474/LTC1475 Regulated Output Voltage LTC1474-3.3/LTC1475-3.3 LTC1474-5/LTC1475-5 Feedback Current LTC1474/LTC1475 Only No Load Supply Current (Note 3) Output Voltage Line Regulation Output Voltage Load Regulation Output Ripple IQ Input DC Supply Current (Note 2) Active Mode (Switch On) Sleep Mode (Note 3) Shutdown Switch Resistance Current Comp Max Current Trip Threshold Current Comp Sense Voltage Trip Threshold Voltage Comparator Hysteresis Switch Off-Time Low Battery Comparator Threshold Run/ON Pin Threshold OFF Pin Threshold (LTC1475 Only) Sink Current into Pin 2 Source Current from Pin 8 Switch Leakage Current Leakage Current into Pin 3 Leakage Current into Pin 2 TA = 25°C, VIN = 10V, VRUN = open, RSENSE = 0, unless otherwise noted. CONDITIONS ILOAD = 50mA ILOAD = 50mA q q q q MIN 1.205 TYP 1.230 MAX 1.255 UNITS V 3.234 4.900 3.300 5.000 0 10 5 2 50 100 9 6 1.4 3.366 5.100 30 V V nA µA IFB ISUPPLY ∆VOUT ILOAD = 0 (Figure 1 Circuit) VIN = 7V to 12V, ILOAD = 50mA ILOAD = 0mA to 50mA ILOAD = 10mA (Exclusive of Driver Gate Charge Current) VIN = 3V to 18V VIN = 3V to 18V VIN = 3V to 18V, VRUN = 0V ISW = 100mA RSENSE = 0Ω RSENSE = 1.1Ω q 20 15 mV mV mVP-P 175 15 12 1.6 85 110 6.0 1.27 1.0 1.0 1.2 1 0.1 0.5 µA µA µA Ω mA mA mV mV µs µs V V V mA µA µA µA µA RON IPEAK VSENSE VHYST tOFF VLBI, TRIP VRUN VLBI, OFF ILBO, SINK IRUN, SOURCE ISW, LEAK ILBI, LEAK ILBO, LEAK 325 70 90 3.5 q 400 76 100 5 4.75 65 1.23 0.7 0.7 0.70 0.8 0.015 0 0 VOUT at Regulated Value VOUT = 0V 1.16 0.4 0.4 VLBI = 0V, VLBO = 0.4V VRUN = 0V VIN = 18V, VSW = 0V, VRUN = 0V VLBI = 18V, VIN = 18V VLBI = 2V, VLBO = 5V 0.45 0.4 The q denotes specifications which apply over the full operating temperature range. Note 1: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC1474CS8/LTC1475CS8: TJ = TA + (PD • 110°C/W) LTC1474CMS8/LTC1475CMS8: TJ = TA + (PD • 150°C/W) Note 2: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 3: No load supply current consists of sleep mode DC current (9µA typical) plus a small switching component (about 1µA for Figure 1 circuit) necessary to overcome Schottky diode and feedback resistor leakage. 3 LTC1474/LTC1475 TYPICAL PERFORMANCE CHARACTERISTICS Efficiency vs Input Voltage 100 95 FIGURE 1 CIRCUIT L: CDRH73-101 40 30 20 FIGURE 1 CIRCUIT ILOAD = 100mA RSENSE = 0Ω EFFICIENCY (%) 90 85 80 ∆VOUT (mV) ∆VOUT (mV) ILOAD = 25mA ILOAD = 200mA ILOAD = 1mA 75 70 –10 –20 0 4 12 8 INPUT VOLTAGE (V) Current Trip Threshold vs Temperature 500 VIN = 10V 5 CURRENT TRIP THRESHOLD (mA) 400 RSENSE = 0Ω 300 3 T = 70°C SUPPLY CURRENT (µA) RDS(ON) (Ω) 200 RSENSE = 1.1Ω 100 0 0 20 40 TEMPERATURE (°C) 60 Switch Leakage Current vs Temperature 1.0 VIN = 18V LEAKAGE CURRENT (µA) 0.8 SUPPLY CURRENT (µA) 100 120 0.6 OFF-TIME (µs) 0.4 0.2 0 0 20 40 60 TEMPERATURE (°C) 4 UW 16 1474/75 G01 Line Regulation 40 30 20 10 Load Regulation FIGURE 1 CIRCUIT VIN = 15V 10 RSENSE = 0.33Ω 0 VIN = 10V 0 –10 –20 –30 VIN = 5V 0 4 12 8 INPUT VOLTAGE (V) 16 1474/75 G02 0 50 150 200 100 LOAD CURRENT (mA) 250 300 1474/75 G03 Switch Resistance vs Input Voltage 10.0 Supply Current in Shutdown 4 7.5 5.0 2 1 T = 25°C 2.5 0 80 1474/75 G04 0 0 5 10 INPUT VOLTAGE (V) 1474/75 G05 15 20 0 5 10 INPUT VOLTAGE (V) 15 20 1474/75 G06 VIN DC Supply Current 80 ACTIVE MODE Off-Time vs Output Voltage VIN = 10V 60 80 60 40 40 20 20 0 SLEEP MODE 0 0 4 12 8 INPUT VOLTAGE (V) 16 20 80 100 1474/75 G07 0 40 100 20 60 80 % OF REGULATED OUTPUT VOLTAGE (%) 1474/75 G09 1474/75 G08 LTC1474/LTC1475 PIN FUNCTIONS VOUT/VFB (Pin 1): Feedback of Output Voltage. In the fixed versions, an internal resistive divider divides the output voltage down for comparison to the internal 1.23V reference. In the adjustable versions, this divider must be implemented externally. LBO (Pin 2): Open Drain Output of the Low Battery Comparator. This pin will sink current when Pin 3 is below 1.23V. LBI/OFF (Pin 3): Input to Low Battery Comparator. This input is compared to the internal 1.23V reference. For the LTC1475, a momentary ground on this pin puts regulator in shutdown mode. GND (Pin 4): Ground Pin. SW (Pin 5): Drain of Internal PMOS Power Switch. Cathode of Schottky diode must be closely connected to this pin. SENSE (Pin 6): Current Sense Input for Monitoring Switch Current and Source of Internal PMOS Power Switch. Maximum switch current is programmed with a resistor between SENSE and VIN pins. VIN (Pin 7): Main Supply Pin. RUN/ON (Pin 8): On LTC1474, voltage level on this pin controls shutdown/run mode (ground = shutdown, open/ high = run). On LTC1475, a momentary ground on this pin puts regulator in run mode. A 100k series resistor must be used between Pin 8 and the switch or control voltage. FUNCTIONAL DIAGRA LBI/OFF 1µA LTC1474: RUN 8 LTC1475: ON 4.75µs 1-SHOT TRIGGER OUT SW 5 VOUT 1× 20× STRETCH WAKEUP LBO 2 + LB – READY 3 LTC1474: LBI LTC1475: LBI/OFF × CONNECTION NOT PRESENT IN LTC1474 SERIES CONNECTION PRESENT IN LTC1474 SERIES ONLY W U U U U U 100mV VIN 7 RSENSE (OPTIONAL) VCC – × C ON + VIN + – V ON 6 5Ω SENSE + + 1.23V 1.23V REFERENCE 4 GND 1M 3M (5V VERSION) 1.68M (3.3V VERSION) VOUT/VFB 1 OUTPUT DIVIDER IS IMPLEMENTED EXTERNALLY IN ADJUSTABLE VERSIONS 1474/75 FD 5 LTC1474/LTC1475 OPERATIO The LTC1474/LTC1475 are step-down converters with internal power switches that use Burst Mode operation to keep the output capacitor charged to the proper output voltage while minimizing the quiescent current. Burst Mode operation functions by using short “burst” cycles to ramp the inductor current through the internal power switch and external Schottky diode, followed by a sleep cycle where the power switch is off and the load current is supplied by the output capacitor. During sleep mode, the LTC1474/LTC1475 draw only 9µA typical supply current. At light loads, the burst cycles are a small percentage of the total cycle time; thus the average supply current is very low, greatly enhancing efficiency. Burst Mode Operation At the beginning of the burst cycle, the switch is turned on and the inductor current ramps up. At this time, the internal current comparator is also turned on to monitor the switch current by measuring the voltage across the internal and optional external current sense resistors. When this voltage reaches 100mV, the current comparator trips and pulses the 1-shot timer to start a 4.75µs off-time during which the switch is turned off and the inductor current ramps down. At the end of the off-time, if the output voltage is less than the voltage comparator threshold, the switch is turned back on and another cycle commences. To minimize supply current, the current comparator is turned on only during the switch-on period when it is needed to monitor switch current. Likewise, the 1-shot timer will only be on during the 4.75µs off-time period. The average inductor current during a burst cycle will normally be greater than the load current, and thus the output voltage will slowly increase until the internal voltage comparator trips. At this time, the LTC1474/LTC1475 go into sleep mode, during which the power switch is off and only the minimum required circuitry is left on: the voltage comparator, reference and low battery comparator. During sleep mode, with the output capacitor supplying the load current, the VFB voltage will slowly decrease until it reaches the lower threshold of the voltage comparator (about 5mV below the upper threshold). The voltage comparator then trips again, signaling the LTC1474/ LTC1475 to turn on the circuitry necessary to begin a new burst cycle. 6 U (Refer to Functional Diagram) Peak Inductor Current Programming The current comparator provides a means for programming the maximum inductor/switch current for each switch cycle. The 1X sense MOSFET, a portion of the main power MOSFET, is used to divert a sample (about 5%) of the switch current through the internal 5Ω sense resistor. The current comparator monitors the voltage drop across the series combination of the internal and external sense resistors and trips when the voltage exceeds 100mV. If the external sense resistor is not used (Pins 6 and 7 shorted), the current threshold defaults to the 400mA maximum due to the internal sense resistor. Off-Time The off-time duration is 4.75µs when the feedback voltage is close to the reference; however, as the feedback voltage drops, the off-time lengthens and reaches a maximum value of about 65µs when this voltage is zero. This ensures that the inductor current has enough time to decay when the reverse voltage across the inductor is low such as during short circuit. Shutdown Mode Both LTC1474 and LTC1475 provide a shutdown mode that turns off the power switch and all circuitry except for the low battery comparator and 1.23V reference, further reducing DC supply current to 6µA typical. The LTC1474’s run/shutdown mode is controlled by a voltage level at the RUN pin (ground = shutdown, open/high = run). The LTC1475’s run/shutdown mode, on the other hand, is controlled by an internal S/R flip-flop to provide pushbutton on/off control. The flip-flop is set (run mode) by a momentary ground at the ON pin and reset (shutdown mode) by a momentary ground at the LBI/OFF pin. Low Battery Comparator The low battery comparator compares the voltage on the LBI pin to the internal reference and has an open drain N-channel MOSFET at its output. If LBI is above the reference, the output FET is off and the LBO output is high impedance. If LBI is below the reference, the output FET is on and sinks current. The comparator is still active in shutdown. LTC1474/LTC1475 APPLICATIONS INFORMATION The basic LTC1474/LTC1475 application circuit is shown in Figure 1, a high efficiency step-down converter. External component selection is driven by the load requirement and begins with the selection of RSENSE. Once RSENSE is known, L can be chosen. Finally D1, CIN and COUT are selected. RSENSE Selection The current sense resistor (RSENSE) allows the user to program the maximum inductor/switch current to optimize the inductor size for the maximum load. The LTC1474/ LTC1475 current comparator has a maximum threshold of 100mV/(RSENSE + 0.25). The maximum average output current IMAX is equal to this peak value less half the peakto-peak ripple current ∆IL. Allowing a margin for variations in the LTC1474/LTC1475 and external components, the required RSENSE can be calculated from Figure 2 and the following equation: RSENSE = (0.067/IMAX) – 0.25 for 10mA < IMAX < 200mA. 5 4 FOR LOWEST NOISE RSENSE (Ω) 3 FOR BEST EFFICIENCY 2 1 0 0 250 100 150 200 50 MAXIMUM OUTPUT CURRENT (mA) 300 1474/75 F02 Figure 2. RSENSE Selection For IMAX above 200mA, RSENSE is set to zero by shorting Pins 6 and 7 to provide the maximum peak current of 400mA (limited by the fixed internal sense resistor). This 400mA default peak current can be used for lower IMAX if desired to eliminate the need for the sense resistor and associated decoupling capacitor. However, for optimal performance, the peak inductor current should be set to no more than what is needed to meet the load current require- U W U U ments. Lower peak currents have the advantage of lower output ripple (∆VOUT = IPEAK • ESR), lower noise, and less stress on alkaline batteries and other circuit components. Also, lower peak currents allow the use of inductors with smaller physical size. Peak currents as low as 10mA can be programmed with the appropriate sense resistor. Increasing RSENSE above 10Ω, however, gives no further reduction of IPEAK. For RSENSE values less than 1Ω, it is recommended that the user parallel standard resistors (available in values ≥ 1Ω) instead of using a special low valued shunt resistor. Although a single resisor could be used with the desired value, these low valued shunt resistor types are much more expensive and are currently not available in case sizes smaller than 1206. Three or four 0603 size standard resistors require about the same area as one 1206 size current shunt resistor at a fraction of the cost. At higher supply voltages and lower inductances, the peak currents may be slightly higher due to current comparator overshoot and can be predicted from the second term in the following equation: 0.1 IPEAK = + 0.25 + RSENSE (1) (2.5)10−7 (VIN − VOUT) L (2) Note that RSENSE only sets the maximum inductor current peak. At lower dI/dt (lower input voltages and higher inductances), the observed peak current at loads less than IMAX may be less than this calculated peak value due to the voltage comparator tripping before the current ramps up high enough to trip the current comparator. This effect improves efficiency at lower loads by keeping the I2R losses down (see Efficiency Considerations section). Inductor Value Selection Once RSENSE and IPEAK are known, the inductor value can be determined. The minimum inductance recommended as a function of IPEAK and IMAX can be calculated from: 0.75 VOUT + VD tOFF   L MIN ≥   IPEAK − IMAX    where tOFF = 4.75µs. ( ) (3) 7 LTC1474/LTC1475 APPLICATIONS INFORMATION If the LMIN calculated is not practical, a larger IPEAK should be used. Although the above equation provides the minimum, better performance (efficiency, line/load regulation, noise) is usually gained with higher values. At higher inductances, peak current and frequency decrease (improving efficiency) and inductor ripple current decreases (improving noise and line/load regulation). For a given inductor type, however, as inductance is increased, DC resistance (DCR) increases, increasing copper losses, and current rating decreases, both effects placing an upper limit on the inductance. The recommended range of inductances for small surface mount inductors as a function of peak current is shown in Figure 3. The values in this range are a good compromise between the trade-offs discussed above. If space is not a premium, inductors with larger cores can be used, which extends the recommended range of Figure 3 to larger values. 1000 INDUCTOR VALUE (µH) 500 100 50 10 100 PEAK INDUCTOR CURRENT (mA) 1000 1474/75 F03 Figure 3. Recommended Inductor Values Inductor Core Selection Once the value of L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mµ® cores. Actual core loss is independent of core size for a fixed inductor value, but is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, as discussed in the previous 8 U W U U section, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite and Kool Mµ designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor current above IPEAK and consequent increase in voltage ripple. Do not allow the core to saturate! Coiltronics, Coilcraft, Dale and Sumida make high performance inductors in small surface mount packages with low loss ferrite and Kool Mµ cores and work well in LTC1474/LTC1475 regulators. Catch Diode Selection The catch diode carries load current during the off-time. The average diode current is therefore dependent on the P-channel switch duty cycle. At high input voltages the diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short-circuited. Under this condition, the diode must safely handle IPEAK at close to 100% duty cycle. To maximize both low and high current efficiency, a fast switching diode with low forward drop and low reverse leakage should be used. Low reverse leakage current is critical to maximize low current efficiency since the leakage can potentially approach the magnitude of the LTC1474/ LTC1475 supply current. Low forward drop is critical for high current efficiency since loss is proportional to forward drop. These are conflicting parameters (see Table 1), but a good compromise is the MBR0530 0.5A Schottky diode specified in the application circuits. Table 1. Effect of Catch Diode on Performance DIODE (D1) BAS85 MBR0530 MBRS130 LEAKAGE 200nA 1µA 20µA FORWARD NO LOAD DROP SUPPLY CURRENT EFFICIENCY* 0.6V 0.4V 0.3V 9.7µA 10µA 16µA 77.9% 83.3% 84.6% *Figure 1 circuit with VIN = 15V, IOUT = 0.1A Kool Mµ is a registered trademark of Magnetics, Inc. LTC1474/LTC1475 APPLICATIONS INFORMATION CIN and COUT Selection At higher load currents, when the inductor current is continuous, the source current of the P-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum capacitor current is given by: IMAX VOUT VIN − VOUT VIN negligible ESR. AVX and Marcon are good sources for these capacitors. In surface mount applications multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Other capacitor types include SANYO OS-CON, Nichicon PL series and Sprague 595D series. Consult the manufacturer for other specific recommendations. To avoid overheating, the output capacitor must be sized to handle the ripple current generated by the inductor. The worst-case ripple current in the output capacitor is given by: IRMS = IPEAK / 2 Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, three main sources usually account for most of the losses in LTC1474/LTC1475 circuits: VIN current, I2R losses and catch diode losses. 1. The VIN current is due to two components: the DC bias current and the internal P-channel switch gate charge current. The DC bias current is 9µA at no load and increases proportionally with load up to a constant 100µA during continuous mode. This bias current is so CIN Required IRMS = [ ( This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Do not underspecify this component. An additional 0.1µF ceramic capacitor is also required on VIN for high frequency decoupling. The selection of COUT is driven by the required effective series resistance (ESR) to meet the output voltage ripple and line regulation requirements. The output voltage ripple during a burst cycle is dominated by the output capacitor ESR and can be estimated from the following relation: 25mV < ∆VOUT, RIPPLE = ∆IL • ESR where ∆IL ≤ IPEAK and the lower limit of 25mV is due to the voltage comparator hysteresis. Line regulation can also vary with COUT ESR in applications with a large input voltage range and high peak currents. ESR is a direct function of the volume of the capacitor. Manufacturers such as Nichicon, AVX and Sprague should be considered for high performance capacitors. The OS-CON semiconductor dielectric capacitor available from SANYO has the lowest ESR for its size at a somewhat higher price. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. For lower current applications with peak currents less than 50mA, 10µF ceramic capacitors provide adequate filtering and are a good choice due to their small size and almost U W U U )] 1/ 2 9 LTC1474/LTC1475 APPLICATIONS INFORMATION small that this loss is negligible at loads above a milliamp but at no load accounts for nearly all of the loss. The second component, the gate charge current, results from switching the gate capacitance of the internal P-channel switch. Each time the gate is switched from high to low to high again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is the current out of VIN which is typically much larger than the DC bias current. In continuous mode, IGATECHG = fQP where QP is the gate charge of the internal switch. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are predicted from the internal switch, inductor and current sense resistor. At low supply voltages where the switch on-resistance is higher and the switch is on for longer periods due to higher duty cycle, the switch losses will dominate. Keeping the peak currents low with the appropriate RSENSE and with larger inductance helps minimize these switch losses. At higher supply voltages, these losses are proportional to load and result in the flat efficiency curves seen in Figure 1. 3. The catch diode loss is due to the VDID loss as the diode conducts current during the off-time and is more pronounced at high supply voltage where the on-time is short. This loss is proportional to the forward drop. However, as discussed in the Catch Diode section, diodes with lower forward drops often have higher leakage current, so although efficiency is improved, the no load supply current will increase. Adjustable Applications For adjustable versions, the output voltage is programmed with an external divider from VOUT to VFB (Pin 1) as shown in Figure 4. The regulated voltage is determined by: To minimize no-load supply current, resistor values in the megohm range should be used. The increase in supply current due to the feedback resistors can be calculated from:  V V ∆I VIN =  OUT   OUT   R1 + R2  VIN  A 10pF feedforward capacitor across R2 is necessary due to the high impedances to prevent stray pickup and improve stability. VOUT  R2 VOUT = 1.23 1+   R1 10 U W U U 10pF 1 LTC1474 V FB LTC1475 GND 4 R2 R1 1474/75 F04 Figure 4. LTC1474/LTC1475 Adjustable Configuration Low Battery Comparator The LTC1474/LTC1475 have an on-chip low battery comparator that can be used to sense a low battery condition when implemented as shown in Figure 5. The resistive divider R3/R4 sets the comparator trip point as follows:  R4  VTRIP = 1.23 1 +   R3  The divided down voltage at the LBI pin is compared to the internal 1.23V reference. When VLBI < 1.23V, the LBO output sinks current. The low battery comparator is active all the time, even during shutdown mode. VIN R4 LBI LTC1474/LTC1475 LBO (4) R3 – + 1.23V REFERENCE 1474/75 F05 Figure 5. Low Battery Comparator LTC1474/LTC1475 APPLICATIONS INFORMATION LTC1475 Pushbutton On/Off and Microprocessor Interface The LTC1475 provides pushbutton control of power on/off for use with handheld products. A momentary ground on the ON pin sets an internal S/R latch to run mode while a momentary ground on the LBI/OFF pin resets the latch to shutdown mode. See Figure 6 for a comparsion of on/off operation of the LTC1474 and LTC1475 and Figure 7 for a comparison of the circuit implementations. In the LTC1475, the LBI/OFF pin has a dual function as both the shutdown control pin and the low battery comparator input. Since the “OFF” pushbutton is normally open, it does not affect the normal operation of the low battery comparator. In the unpressed state, the LBI/OFF input is the divided down input voltage from the resistive divider to the internal low battery comparator and will normally be above or just below the trip threshold of 1.23V. When shutdown mode is desired, the LBI/OFF pin is pulled below the 0.7V threshold to invoke shutdown. RUN LTC1474 MODE RUN SHUTDOWN RUN 1474/75 F07 ON OVERRIDES LBI/OFF WHILE ON IS LOW ON LBI/OFF LTC1475 MODE RUN SHUTDOWN RUN Figure 6. Comparison of LTC1474 and LTC1475 Run/Shutdown Operation The ON pin has precedence over the LBI/OFF pin. As seen in Figure 6, if both pins are grounded simultaneously, run mode wins. Figure 18 in the Typical Applications section shows an example for the use of the LTC1475 to control on/off of a microcontroller with a single pushbutton. With both the microcontroller and LTC1475 off, depressing the pushbutton grounds the LTC1475 ON pin and starts up the LTC1475 regulator which then powers up the microcontroller. When the pushbutton is depressed a second time, U W U U the depressed switch state is detected by the microcontroller through its input. The microcontroller then pulls the LBI/OFF pin low with the connection to one of its ouputs. With the LBI/OFF pin low, the LTC1475 powers down turning the microcontroller off. Note that since the I/O pins of most microcontrollers have a reversed bias diode between input and supply, a blocking diode with less than 1µA leakage is necessary to prevent the powered down microcontroller from pulling down on the ON pin. Figure 19 in the Typical Applications section shows how to use the low battery comparator to provide a low battery lockout on the “ON” switch. The LBO output disconnects the pushbutton from the ON pin when the comparator has tripped, preventing the LTC1475 from attempting to start up again until VIN is increased. 100k RUN VIN RUN LTC1474 ON LTC1475 LBI/OFF OFF 100k ON Figure 7. Simplified Implementation of LTC1474 and LTC1475 On/Off Absolute Maximum Ratings and Latchup Prevention 1474/75 F06 The absolute maximum ratings specify that SW (Pin 5) can never exceed VIN (Pin 7) by more than 0.3V. Normally this situation should never occur. It could, however, if the output is held up while the supply is pulled down. A condition where this could potentially occur is when a battery is supplying power to an LTC1474 or LTC1475 regulator and also to one or more loads in parallel with the the regulator’s VIN. If the battery is disconnected while the LTC1474 or LTC1475 regulator is supplying a light load and one of the parallel circuits is a heavy load, the input capacitor of the LTC1474 or LTC1475 regulator could be pulled down faster than the output capacitor, causing the absolute maximum ratings to be exceeded. The result is often a latchup which can be destructive if VIN is reapplied. Battery disconnect is possible as a result of mechanical stress, bad battery contacts or use of a lithium-ion battery 11 LTC1474/LTC1475 APPLICATIONS INFORMATION with a built-in internal disconnect. The user needs to assess his/her application to determine whether this situation could occur. If so, additional protection is necessary. Prevention against latchup can be accomplished by simply connecting a Schottky diode across the SW and VIN pins as shown in Figure 8. The diode will normally be reverse biased unless VIN is pulled below VOUT at which time the diode will clamp the (VOUT – VIN) potential to less than the 0.6V required for latchup. Note that a low leakage Schottky should be used to minimize the effect on no-load supply current. Schottky diodes such as MBR0530, BAS85 and BAT84 work well. Another more serious effect of the protection diode leakage is that at no load with nothing to provide a sink for this leakage current, the output voltage can potentially float above the maximum allowable tolerance. To prevent this from occuring, a resistor must be connected between VOUT and ground with a value low enough to sink the maximum possible leakage current. LATCHUP PROTECTION SCHOTTKY VIN SW VOUT LTC1474 LTC1475 + 1474/75 F08 Figure 8. Preventing Absolute Maximum Ratings from Being Exceeded Thermal Considerations In the majority of the applications, the LTC1474/LTC1475 do not dissipate much heat due to their high efficiency. However, in applications where the switching regulator is running at high ambient temperature with low supply voltage and high duty cycles, such as dropout with the switch on continuously, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated by the regulator exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = P • θJA 12 U W U U where P is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature is given by: TJ = TA + TR As an example consider the LTC1474/LTC1475 in dropout at an input voltage of 3.5V, a load current of 300mA, and an ambient temperature of 70°C. From the typical performance graph of switch resistance, the on-resistance of the P-channel switch at 70°C is 3.5Ω. Therefore, power dissipated by the part is: P = I2 • RDS(ON) = 0.315W For the MSOP package, the θJA is 150°C/W. Thus the junction temperature of the regulator is: TJ = 70°C + (0.315)(150) = 117°C which is near the maximum junction temperature of 125oC. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1474/LTC1475. These items are also illustrated graphically in the layout diagram of Figure 9. Check the following in your layout: 1. Is the Schottky diode cathode closely connected to SW (Pin 5)? 2. Is the 0.1µF input decoupling capacitor closely connected between VIN (Pin 7) and ground (Pin 4)? This capacitor carries the high frequency peak currents. 3. When using adjustable version, is the resistive divider closely connected to the (+) and (–) plates of COUT with a 10pF capacitor connected across R2? 4. Is the 1000pF decoupling capacitor for the current sense resistor connected as close as possible to Pins 6 and 7? If no current sense resistor is used, Pins 6 and 7 should be shorted. LTC1474/LTC1475 APPLICATIONS INFORMATION OUTPUT DIVIDER REQUIRED WITH ADJUSTABLE VERSION ONLY 10pF VOUT 1 R2 R1 2 3 COUT 4 VFB LBO LBI GND SENSE SW 6 5 1474/75 F09 + BOLD LINES INDICATE HIGH PATH CURRENTS Figure 9. LTC1474/LTC1475 Layout Diagram (See Board Layout Checklist) 5. Are the signal and power grounds segregated? The signal ground consists of the (–) plate of COUT, Pin 4 of the LTC1474/LTC1475 and the resistive divider. The power ground consists of the Schottky diode anode, the (–) plate of CIN and the 0.1µF decoupling capacitor. 6. Is a 100k resistor connected in series between RUN (Pin 8) and the RUN control voltage? The resistor should be as close as possible to Pin 8. Design Example (Refer to RSENSE and Inductor Selection) As a design example, assume VIN = 10V, VOUT = 3V, and a maximum average output current IMAX = 100mA. With this information, we can easily calculate all the important components: From the equation (1), RSENSE = (0.067/0.1) – 0.25 = 0.42Ω Using the standard resistors (1Ω, 1Ω and 2Ω) in parallel provides 0.4Ω without having to use a more expensive low value current shunt type resistor (see RSENSE Selection section). With RSENSE = 0.4Ω, the peak inductor current IPEAK is calculated from (2), neglecting the second term, to be U W U U LTC1474 8 RUN VIN 7 100k L 1000pF RSENSE 0.1µF CIN D1 + VIN 150mA. The minimum inductance is, therefore, from the equation (3) and assuming VD = 0.4V, L MIN = 0.75 3.3 + 0.4 4.75µs 0.15 − 0.1 ( )( ) = 264µH From Figure 3, an inductance of 270µH is chosen from the recommended region. The CDRH73-271 or CD54-271 is a good choice for space limited applications. For the feedback resistors, choose R1 = 1M to minimize supply current. R2 can then be calculated from the equation (4) to be: V  R2 =  OUT − 1 • R1 = 1.43 M  1.23  For the catch diode, the MBR0530 will work well in this application. For the input and output capacitors, AVX 4.7µF and 100µF, respectively, low ESR TPS series work well and meet the RMS current requirement of 100mA/2 = 50mA. They are available in small “C” case sizes with 0.15Ω ESR. The 0.15Ω output capacitor ESR will result in 25mV of output voltage ripple. Figure 10 shows the complete circuit for this example. 13 LTC1474/LTC1475 TYPICAL APPLICATIONS VIN 3.5V TO 18V + 4.7µF† 35V 1Ω** * SUMIDA CDRH73-271 ** 3 PARALLEL STANDARD RESISTORS PROVIDE LEAST EXPENSIVE SOLUTION (SEE R SENSE SELECTION SECTION) † AVX TPSC475M035 †† AVX TPSC107M006 Figure 10. High Efficiency 3V/100mA Regulator (Design Example) IN + 4mA TO 20mA D2†† 12V 1µF ×3 2Ω 7.5M 1000pF 6 7 VIN 1 2 5 D1 MBR0530 1474/75 F11 † TO A/D MBR0530 IN 4mA TO 20mA – * COILCRAFT DO1608-334 ** MARCON THCS50E1E106Z, AVX 1206ZG106Z † OPTIONAL RESISTOR FOR SENSING LOOP CURRENT BY A/D CONVERTER † † MOTOROLA MMBZ5242BL Figure 11. High Efficiency 3.3V/10mA Output from 4mA to 20mA Loop 14 U 10pF 1000pF 6 3 100k 8 SENSE LBI RUN 0.1µF VFB 1 2 5 D1 MBR0530 1474/75 F10 1Ω** 2Ω** 7 VIN LTC1474 LBO SW 1.43M 1% 1M 1% L* 270µH + VOUT 3V 100mA 100µF†† 6.3V RUN GND 4 SENSE VOUT LTC1474-3.3 3 LBI LBO 8 RUN SW VOUT 3.3V 10mA L* 330µH 10µF** 1M 100k RUN GND 4 LTC1474/LTC1475 TYPICAL APPLICATIONS + 22µF** 16V 6 3 100k 8 SENSE LBI RUN 0.1µF 7 VIN VFB 1 2 5 4.7M 1% 536k 1% VOUT 12V 70mA 22µF†† 25V VIN (V) D1 MBR0530 3.5 4 5 6 I LOAD(MAX) 30mA 50mA 70mA 90mA 1474/75 F12 LTC1474 LBO SW L* 200µH RUN + 10µF† L* 200µH 25V GND 4 * COILTRONICS CTX200-4 ** AVX TPSC226M016 † AVX TPSC106M025 †† AVX TPSD226M025 Figure 12. 5V to ± 12V Regulator VIN 3.5V TO 12V + 10µF** 25V 6 3 100k 8 SENSE LBI RUN 0.1µF 7 VIN VOUT LBO SW 1 2 5 L* 100µH L* 100µH VOUT 5V 200mA AT VIN = 10V LTC1474-5 + 10µF** 25V RUN GND 4 * COILTRONICS CTX100-4 ** AVX TPSC106MO25 † AVX TPSC336M010 D1 MBR0530 Figure 13. 5V Buck-Boost Converter + VIN 3.5V TO 6V + U MBR0530 10pF VOUT –12V 70mA 22µF†† 25V + 33µF† 10V VIN (V) 3.5 4 5 8 10 12 I LOAD(MAX) 70mA 95mA 125mA 180mA 200mA 225mA 1474/75 F13 15 LTC1474/LTC1475 TYPICAL APPLICATIONS VIN 3.5V TO 12V ON/OFF†† + TP0610 10µF** 25V 10M * SUMIDA CDRH74-101 ** AVX TPSC106M025 † AVX TPSC336M010 †† RUN: ON/OFF = 0, SHUTDOWN: 0N/OFF = V IN VIN 8V TO 18V + 4.7µF** 35V CHARGER ON/OFF * SUMIDA CDRH73-101 ** AVX TPSC475M035 † AVX TPSD476M016 16 U 0.1µF 7 6 3 8 SENSE LBI RUN VIN VOUT LBO SW 1 2 5 D1 MBR0530 L* 100µH VIN (V) I LOAD(MAX) 100mA 140mA 190mA 240mA 1474/75 F14 LTC1474-5 + 33µF† 10V 3.5 5 8 12 VOUT –5V 140mA AT VIN = 5V GND 4 Figure 14. Positive-to-Negative (– 5V) Converter 10pF 0.1µF 7 6 3 100k 8 SENSE LBI RUN VIN VFB 1 2 5 D1 MBR0530 1474/75 F15 4.69M 1M L* 100µH MBR0530 + LTC1474 LBO SW 47µF† 16V VOUT 4-NiCd 200mA GND 4 Figure 15. 4-NiCd Battery Charger LTC1474/LTC1475 TYPICAL APPLICATIONS VIN 4V TO 18V + 4.7µF† 35V * SUMIDA CDRH73-101 † AVX TPSC475M035 †† AVX TPSC107M006 Figure 16. High Efficiency 3.3V Regulator with Low Battery Lockout VIN 5.7V TO 18V + 4.7µF** 35V OFF VCC VIN 4V TO 18V µC * SUMIDA CDRH73-101 ** AVX TPSC475M035 † AVX TPSC107M006 Figure 18. LTC1475 Regulator with 1-Button Toggle On/Off U 0.1µF 2.2M 6 3 1M 100k RUN 8 SENSE LBI RUN 7 VIN VOUT LBO SW 1 2 5 D1 MBR0530 1474/75 F16 LTC1474-3.3 L* 100µH + 100µF†† 6.3V VOUT 3.3V 250mA GND 4 0.1µF 7 3.65M 6 3 100k 8 1M ON SENSE LBI/OFF ON VIN LTC1475-5 LBO SW VOUT 1 2 5 D1 MBR0530 * SUMIDA CDRH73-101 ** AVX TPSC475M035 † AVX TPSC336M010 1474/75 F17 + L* 100µH 33µF† 10V VOUT 5V 250mA GND 4 Figure 17. Pushbutton On/Off 5V/250mA Regulator + 4.7µF** 35V 100k 6 SENSE 7 VIN 0.1µF MMBD914LT1 8 2 ON VOUT 1 0.1µF ON/OFF LTC1475-3.3 LBO SW 5 L* 100µH GND 4 + 2.2M VOUT 3.3V 250mA 100µF† 6.3V 3 1M LBI/OFF D1 MBR0530 1474/75 F18 17 LTC1474/LTC1475 PACKAGE DESCRIPTION 0.007 (0.18) 0.021 ± 0.006 (0.53 ± 0.015) * DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE 18 U Dimensions in inches (millimeters) unless otherwise noted. MS8 Package 8-Lead Plastic MSOP (LTC DWG # 05-08-1660) 0.118 ± 0.004* (3.00 ± 0.102) 8 76 5 0.192 ± 0.004 (4.88 ± 0.10) 0.118 ± 0.004** (3.00 ± 0.102) 1 0.040 ± 0.006 (1.02 ± 0.15) 0° – 6° TYP SEATING PLANE 0.012 (0.30) 0.0256 REF (0.65) TYP 23 4 0.034 ± 0.004 (0.86 ± 0.102) 0.006 ± 0.004 (0.15 ± 0.102) MSOP (MS8) 1197 LTC1474/LTC1475 PACKAGE DESCRIPTION 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0°– 8° TYP 0.016 – 0.050 0.406 – 1.270 *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U Dimensions in inches (millimeters) unless otherwise noted. S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 8 7 6 5 0.228 – 0.244 (5.791 – 6.197) 0.150 – 0.157** (3.810 – 3.988) 1 0.053 – 0.069 (1.346 – 1.752) 2 3 4 0.004 – 0.010 (0.101 – 0.254) 0.014 – 0.019 (0.355 – 0.483) 0.050 (1.270) TYP SO8 0996 19 LTC1474/LTC1475 TYPICAL APPLICATION VIN 3.5V to 18V + 1M 4.7µF** 35V MMBT2N2222LT1 ON RELATED PARTS PART NUMBER LTC1096/LTC1098 LT1121/LT1121-3.3/LT1121-5 LTC1265 LT1375/LT1376 LTC1440/LTC1441/LTC1442 LT1495/LT1496 LT1521/LT1521-3/LT1521-3.3/ LT1521-5 LT1634-1.25 DESCRIPTION Micropower Sampling 8-Bit Serial I/O A/D Converter 150mA Low Dropout Regulator 1.2A High Efficiency Step-Down DC/DC Converter 1.5A 500kHz Step-Down Switching Regulators Ultralow Power Comparator with Reference 1.5µA Precision Rail-to-Rail Op Amps 300mA Low Dropout Regulator COMMENTS IQ = 80µA Max Linear Regulator, IQ = 30µA Selectable IPEAK = 300mA or 600mA Burst Mode Operation, Internal MOSFET 500kHz, Small Inductor, High Efficiency Switchers, 1.5A Switch IQ = 2.8µA Max IQ = 1.5µA Max Linear Regulator, IQ = 12µA LTC1174/LTC1174-3.3/LTC1174-5 High Efficiency Step-Down and Inverting DC/DC Converters LTC1574/LTC1574-3.3/LTC1574-5 High Efficiency Step-Down DC/DC Converters with Internal Schottky Diode LTC1174 with Internal Schottky Diode Micropower Precision Shunt Reference IQ(MIN) = 10µA 20 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 q (408) 432-1900 FAX: (408) 434-0507q TELEX: 499-3977 q www.linear-tech.com U 10pF 0.1µF 7 1.8M 100k 6 8 3 1M OFF SENSE ON VIN VFB LBO SW 1 2 5 D1 MBR0530 1.02M 1% 1M 1% VOUT 2.5V 250mA 100µF† 6.3V + L* 100µH LTC1475 LBI/OFF GND 4 * SUMIDA CDRH73-101 ** AVX TPSC475M035 † AVX TPSC107M006 1474/75 F19 Figure 19. Pushbutton On/Off with Low Battery Lockout 14745fa LT/TP 0398 4K REV A • PRINTED IN USA © LINEAR TECHNOLOGY CORPORATION 1997