LTC1622CS8

LTC1622 Low Input Voltage Current Mode Step-Down DC/DC Controller FEATURES s s s s s s s s s DESCRIPTIO s s s s High Efficiency Constant Frequency 550kHz Operation VIN Range: 2V to 10V Multiampere Output Currents OPTI-LOOPTM Compensation Minimizes COUT Selectable, Burst Mode Operation Low Dropout Operation: 100% Duty Cycle Synchronizable up to 750kHz Current Mode Operation for Excellent Line and Load Transient Response Low Quiescent Current: 350µA Shutdown Mode Draws Only 15µA Supply Current ± 1.9% Reference Accuracy Available in 8-Lead MSOP APPLICATIO S s s s s s s s 1- or 2-Cell Li-Ion Powered Applications Cellular Telephones Wireless Modems Portable Computers Distributed 3.3V, 2.5V or 1.8V Power Systems Scanners Battery-Powered Equipment , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode and OPTI-LOOP are a trademarks of Linear Technology Corporation. The LTC®1622 is a constant frequency current mode stepdown DC/DC controller providing excellent AC and DC load and line regulation. The device incorporates an accurate undervoltage feature that shuts the LTC1622 down when the input voltage falls below 2V. The LTC1622 boasts a ± 1.9% output voltage accuracy and consumes only 350µA of quiescent current. For applications where efficiency is a prime consideration and the load current varies from light to heavy, the LTC1622 can be configured for Burst ModeTM operation. Burst Mode operation enhances low current efficiency and extends battery run time. Burst Mode operation is inhibited during synchronization or when the SYNC/MODE pin is pulled low to reduce noise and possible RF interference. High constant operating frequency of 550kHz allows the use of a small inductor. The device can also be synchronized up to 750kHz for special applications. The high frequency operation and the available 8-lead MSOP package create a high performance solution in an extremely small amount of PCB area. To further maximize the life of the battery source, the P-channel MOSFET is turned on continuously in dropout (100% duty cycle). In shutdown, the device draws a mere 15µA. TYPICAL APPLICATIO 8 2 R1 10k C3 220pF 4 470pF VIN ITH SENSE – 1 7 VIN 2.5V TO 8.5V R2 0.03Ω Si3443DV L1 4.7µH D1 IR10BQ015 R3 159k R4 75k L1: MURATA LQN6C-4R7 R2: DALE WSL-1206 0-03Ω C1 10µF 10V Efficiency vs Load Current with Burst Mode Operation Enabled 100 VIN = 4.2V 90 EFFICIENCY (%) VIN = 3.3V PDRV LTC1622 5 SYNC/MODE RUN/SS GND VFB 3 6 80 VIN = 6V 70 60 50 VOUT = 2.5V RSENSE = 0.03Ω 1 100 1000 10 LOAD CURRENT (mA) 5000 1622 F01b VOUT 2.5V 1.5A + C2 47µF 6V C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: SANYO POSCAP 6TPA47M D1: INTERNATIONAL RECTIFIER IR10BQ015 1622 F01a 40 Figure 1. High Efficiency Step-Down Converter U VIN = 8.4V U U 1 LTC1622 ABSOLUTE MAXIMUM RATINGS Input Supply Voltage (VIN).........................– 0.3V to 10V RUN/SS Voltage .......................................– 0.3V to 2.4V SYNC/MODE Voltage ................................. – 0.3V to VIN SENSE – Voltage .......................................... 2.4V to VIN PDRV Peak Output Current (< 10µs) ......................... 1A Storage Ambient Temperature Range ... – 65°C to 150°C PACKAGE/ORDER INFORMATION ORDER PART NUMBER TOP VIEW SENSE – ITH VFB RUN/SS 1 2 3 4 8 7 6 5 VIN PDRV GND SYNC/MODE TOP VIEW SENSE – 1 ITH 2 VFB 3 8 7 6 5 VIN PDRV GND SYNC/MODE LTC1622CMS8 MS8 PART MARKING LTDB MS8 PACKAGE 8-LEAD PLASTIC MSOP TJMAX = 125°C, θJA = 250°C/ W Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS SYMBOL IVFB VFB VOVL ∆VOSENSE VLOADREG IS PARAMETER Feedback Current Regulated Feedback Voltage Output Overvoltage Lockout Reference Voltage Line Regulation Output Voltage Load Regulation Input DC Supply Current Burst Mode Inhibited Sleep Mode Shutdown Shutdown RUN/SS Threshold Soft-Start Current Source Oscillator Frequency The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V CONDITIONS (Note 3) VFB = 0.8V (Note 3) Commercial Grade (Note 3) Industrial Grade Referenced to Nominal VOUT VIN = 4.2V to 8.5V (Note 3) Measured in Servo Loop; VITH = 0.2V to 0.625V Measured in Servo Loop; VITH = 0.9V to 0.625V (Note 4) VIN = 2.3V VITH = 0V, VSYNC/MODE = 2.4V VRUN/SS = 0V VRUN/SS = 0V, VIN = VUVLO – 0.1V Commercial Grade Industrial Grade VRUN/SS = 0V VFB = 0.8V VFB = 0V VSYNC/MODE Ramping Down VIN Ramping Down VIN Ramping Up q q q q q VRUN/SS IRUN/SS fOSC VSYNC/MODE SYNC/MODE Threshold VUVLO Undervoltage Lockout 2 U U W WW U W (Note 1) Operating Temperature Range Commercial ............................................ 0°C to 70°C Industrial ........................................... – 45°C to 85°C Junction Temperature (Note 2) ............................. 125°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER LTC1622CS8 LTC1622IS8 S8 PART MARKING 1622 1622I RUN/SS 4 S8 PACKAGE 8-LEAD PLASTIC SO TJMAX = 125°C, θJA = 150°C/ W MIN 0.785 0.780 4 TYP 10 0.8 0.8 7.5 0.04 0.3 – 0.3 450 350 15 4 MAX 70 0.815 0.820 10.5 0.08 0.5 – 0.5 UNITS nA V V % %/V % % µA µA µA µA V V µA kHz kHz V V V 400 30 10 0.9 1.0 5 625 140 1.5 2.3 2.36 0.4 0.3 1 475 75 1.55 0.7 0.7 2.5 550 110 1 1.92 1.97 LTC1622 The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V SYMBOL PDRV tr PDRV tf ∆VSENSE(MAX) PARAMETER Gate Drive Rise Time Gate Drive Fall Time Maximum Current Sense Voltage CONDITIONS CLOAD = 3000pF CLOAD = 3000pF q ELECTRICAL CHARACTERISTICS MIN TYP 80 100 MAX 140 140 140 UNITS ns ns mV 80 110 Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC1622CS8; TJ = TA + (PD • 150°C/W), LTC1622CMS8; TJ = TA + (PD • 250°C/W) Note 3: The LTC1622 is tested in a feedback loop that servos VFB to the feedback point for the error amplifier (VITH = 0.8V). Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. TYPICAL PERFORMANCE CHARACTERISTICS Shutdown Current vs Supply Voltage 45 40 35 30 25 20 15 10 5 0 2 3 4 6 7 5 8 SUPPLY VOLTAGE (V) 9 10 SOFT-START CURRENT (µA) SHUTDOWN CURRENT (µA) TRIP VOLTAGE (mV) Normalized Oscillator Frequency vs Temperature 10.0 VIN = 4.2V 0.810 0.805 7.5 5.0 2.5 0 –2.5 –5.0 –7.5 –10.0 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1622 G04 UNDERVOLTAGE LOCKOUT VOLTAGE (V) NORMALIZED FREQUENCY (%) REFERENCE VOLTAGE (V) UW 1622 G01 RUN/SS Current vs Supply Voltage 3.50 110 100 Maximum Current Sense Voltage vs Duty Cycle VIN = 4.2V 3.00 90 UNSYNC 80 70 60 50 40 2.50 2.00 1.50 1.00 2 3 4 5 6 7 8 9 10 SUPPLY VOLTAGE (V) 1622 G02 30 20 30 40 50 60 70 80 DUTY CYCLE (%) 90 100 1622 G03 Reference Voltage vs Temperature VIN = 4.2V 2.10 2.05 2.00 1.95 1.90 1.85 1.80 Undervoltage Lockout Voltage vs Temperature 0.800 0.795 0.790 0.785 0.780 0.775 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1622 G05 1.75 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1622 G06 3 LTC1622 TYPICAL PERFORMANCE CHARACTERISTICS Efficiency vs Load Current for Figure 1 with Burst Mode Operation Defeated 100 VIN = 3.3V 90 EFFICIENCY (%) VIN = 4.2V 100mV/DIV 100mV/DIV ILOAD = 50mA TO 1.2A VIN = 4.2V 1622 G08 80 70 60 50 40 1 10 100 LOAD CURRENT (mA) 1000 1622 G07 VIN = 6V VIN = 8.4V VOUT = 2.5V RSENSE = 0.03Ω PIN FUNCTIONS SENSE – (Pin 1): The Negative Input to the Current Comparator. ITH (Pin 2): Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0V to 1.2V. VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output capacitor. RUN/SS (Pin 4): Combination of Soft-Start and Run Control Inputs. A capacitor to ground at this pin sets the ramp time to full output current. The time is approximately 0.45s/µF. Forcing this pin below 0.4V causes all circuitry to be shut down. SYNC/MODE (Pin 5): This pin performs three functions. Greater than 2V on this pin allows Burst Mode operation at low load currents, while grounding or applying a clock signal on this pin defeats Burst Mode operation. An external clock between 625kHz and 750kHz applied to this pin forces the LTC1622 to operate at the external clock frequency. Do not attempt to synchronize below 625kHz. Pin 5 has an internal 1µA pull-up current source. GND (Pin 6): Ground Pin. PDRV (PIN 7): Gate Drive for the External P-Channel MOSFET. This pin swings from 0V to VIN. VIN (Pin 8): Main Supply Pin. Must be closely decoupled to ground Pin 6. 4 UW Load Step Transient Response Burst Enabled Load Step Transient Response Burst Inhibited ILOAD = 50mA TO 1.2A VIN = 4.2V 1622 G09 U U U LTC1622 FUNCTIONAL DIAGRA VIN BURST DEFEAT X 1µA Y SYNC/ 5 MODE OSC 0.36V 0.3V VFB 3 – + – FREQ SHIFT VIN 0.8V VREF 2.5µA EN SENSE – 1 8 VIN 0.8V REFERENCE VREF 0.8V UVLO TRIP = 1.97V 6 SHUTDOWN GND VIN RUN/SS 4 gm = 0.5m RUN/ SOFT-START + VREF + 60mV – OV OPERATIO (Refer to Functional Diagram) Main Control Loop The LTC1622 is a constant frequency current mode switching regulator. During normal operation, the external P-channel power MOSFET is turned on each cycle when the oscillator sets the RS latch (RS1) and turned off when the current comparator (ICOMP) resets the latch. The peak inductor current at which ICOMP resets the RS latch is controlled by the voltage on the ITH pin, which is the output of the error amplifier EA. An external resistive divider connected between VOUT and ground allows EA to receive an output feedback voltage VFB. When the load current increases, it causes a slight decrease in VFB relative to the 0.8V reference, which in turn causes the ITH voltage to increase until the average inductor current matches the new load current. The main control loop is shut down by pulling the RUN/SS pin low. Releasing RUN/SS allows an internal 2.5µA Ω W Y = “0” ONLY WHEN X IS A CONSTANT “1” OTHERWISE Y = “1” VCC SLOPE COMP U U U + + EA 0.12V – + BURST 2 ITH S R Q RS1 SLEEP + SWITCHING LOGIC AND BLANKING CIRCUIT – ICOMP – VIN PDRV 7 1622 BD current source to charge up the soft-start capacitor CSS. When CSS reaches 0.7V, the main control loop is enabled with the ITH voltage clamped at approximately 5% of its maximum value. As CSS continues to charge, ITH is gradually released allowing normal operation to resume. Comparator OV guards against transient overshoots > 7.5% by turning off the P-channel power MOSFET and keeping it off until the fault is removed. Burst Mode Operation The LTC1622 can be enabled to go into Burst Mode operation at low load currents simply by leaving the SYNC/ MODE pin open or connecting it to a voltage of at least 2V. In this mode, the peak current of the inductor is set as if VITH = 0.36V (at low duty cycles) even though the voltage at the ITH pin is at lower value. If the inductor’s average current is greater than the load requirement, the voltage at 5 LTC1622 OPERATIO the ITH pin will drop. When the ITH voltage goes below 0.12V, the sleep signal goes high, turning off the external MOSFET. The sleep signal goes low when the ITH voltage rises above 0.22V and the LTC1622 resumes normal operation. The next oscillator cycle will turn the external MOSFET on and the switching cycle repeats. Frequency Synchronization The LTC1622 can be externally driven by a TTL/CMOS compatible clock signal up to 750kHz. Do not synchronize the LTC1622 below its maximum default operating frequency of 625kHz as this may cause abnormal operation and an undesired frequency spectrum. The LTC1622 is synchronized to the rising edge of the clock. The external clock pulse width must be at least 100ns and not more than the period minus 200ns. Synchronization is inhibited when the feedback voltage is below 0.3V. This is to prevent inductor current buildup under short-circuit conditions. Burst Mode operation is deactivated when the LTC1622 is externally driven by a clock. Dropout Operation When the input supply voltage decreases towards the output voltage, the rate of change of inductor current during the ON cycle decreases. This reduction means that the P-channel MOSFET will remain on for more than one oscillator cycle since the inductor current has not ramped up to the threshold set by EA. Further reduction in input supply voltage will eventually cause the P-channel MOSFET to be turned on 100%, i.e., DC. The output voltage will then be determined by the input voltage minus the voltage drop across the MOSFET, the sense resistor and the inductor. Undervoltage Lockout To prevent operation of the P-channel MOSFET below safe input voltage levels, an undervoltage lockout is incorporated into the LTC1622. When the input supply voltage drops below 2V, the P-channel MOSFET and all circuitry is turned off except the undervoltage block, which draws only several microamperes. SF = IOUT/IOUT(MAX) (%) 6 U (Refer to Functional Diagram) Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator will be reduced to about 110kHz. This lower frequency allows the inductor current to safely discharge, thereby preventing current runaway. The oscillator’s frequency will gradually increase to its nominal value when the feedback voltage increases above 0.65V. Note that synchronization is inhibited until the feedback voltage goes above 0.3V. Overvoltage Protection As a further protection, the overvoltage comparator in the LTC1622 will turn the external MOSFET off when the feedback voltage has risen 7.5% above the reference voltage of 0.8V. This comparator has a typical hysteresis of 35mV. Slope Compensation and Peak Inductor Current The inductor’s peak current is determined by: IPK = 10 RSENSE ( VITH ) when the LTC1622 is operating below 40% duty cycle. However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor current. The amount of reduction is given by the curves in Figure 2. 110 100 90 80 70 60 50 40 30 20 10 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 1622 F02 IRIPPLE = 0.4IPK AT 5% DUTY CYCLE IRIPPLE = 0.2IPK AT 5% DUTY CYCLE VIN = 4.2V UNSYNC Figure 2. Maximum Output Current vs Duty Cycle LTC1622 APPLICATIONS INFORMATION The basic LTC1622 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L and RSENSE. Next, the Power MOSFET and the output diode D1 are selected followed by CIN and COUT. RSENSE Selection for Output Current RSENSE is chosen based on the required output current. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator determines the inductor’s peak current. The output current the LTC1622 can provide is given by: IOUT = 0.08 I − RIPPLE RSENSE 2 VOUT. The inductor’s peak-to-peak ripple current is given by: V V −V +V  IRIPPLE = IN OUT  OUT D   VIN + VD  fL where IRIPPLE is the inductor peak-to-peak ripple current (see Inductor Value Calculation section). A reasonable starting point for setting ripple current is IRIPPLE = (0.4)(IOUT). Rearranging the above equation, it becomes: 1 RSENSE = for Duty Cycle < 40% 15 IOUT ( )( ) However, for operation that is above 40% duty cycle, slope compensation has to be taken into consideration to select the appropriate value to provide the required amount of current. Using Figure 2, the value of RSENSE is: RSENSE = (15)(IOUT)(100) SF Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use of a smaller inductor for the same amount of inductor ripple current. However, this is at the expense of efficiency due to an increase in MOSFET gate charge losses. The inductance value also has a direct effect on ripple current. The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VIN or U W U U () where f is the operating frequency. Accepting larger values of IRIPPLE allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is IRIPPLE = 0.4(IOUT(MAX)). Remember, the maximum IRIPPLE occurs at the maximum input voltage. With Burst Mode operation selected on the LTC1622, the ripple current is normally set such that the inductor current is continuous during the burst periods. Therefore, the peak-to-peak ripple current should not exceed: IRIPPLE ≤ 0.036 RSENSE This implies a minimum inductance of: V V −V +V  LMIN = IN OUT  OUT D   0.036   VIN + VD  f   RSENSE  (Use VIN(MAX) = VIN) A smaller value than L MIN could be used in the circuit; however, the inductor current will not be continuous during burst periods. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mu® cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are Kool Mu is a registered trademark of Magnetics, Inc. 7 LTC1622 APPLICATIONS INFORMATION preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core materials saturate “hard,” which means that the inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequently, output voltage ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same manufacturer is Kool Mu. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, new surface mountable designs that do not increase the height significantly are available. Power MOSFET Selection An external P-channel power MOSFET must be selected for use with the LTC1622. The main selection criteria for the power MOSFET are the threshold voltage VGS(TH) and the “on” resistance RDS(ON),reverse transfer capacitance CRSS and total gate charge. Since the LTC1622 is designed for operation down to low input voltages, a sublogic level threshold MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required for applications that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the LTC1622 is less than the absolute maximum MOSFET VGS rating, typically 8V. The gate drive voltage levels are from ground to VIN. The required minimum RDS(ON) of the MOSFET is governed by its allowable power dissipation. For applications that may operate the LTC1622 in dropout, i.e., 100% duty cycle, at its worst case the required RDS(ON) is given by: RDS(ON)DC=100% = In applications where the maximum duty cycle is less than 100% and the LTC1622 is in continuous mode, the RDS(ON) is governed by: (IOUT(MAX)) (1+ δp) 2 PP where PP is the allowable power dissipation and δp is the temperature dependency of RDS(ON). (1 + δp) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but δp = 0.005/°C can be used as an approximation for low voltage MOSFETs. 8 U W U U RDS(ON) ≅ (DC)IOUT (1+ δp) 2 PP where DC is the maximum operating duty cycle of the LTC1622. When the LTC1622 is operating in continuous mode, the MOSFET power dissipation is: PMOSFET = ( ) (1+ δp)RDS(ON) 2 + K ( VIN) (IOUT )(CRSS )( f) VOUT + VD IOUT VIN + VD 2 where K is a constant inversely related to gate drive current. Because of the high switching frequency, the second term relating to switching loss is important not to overlook. The constant K = 3 can be used to estimate the contributions of the two terms in the MOSFET dissipation equation. Output Diode Selection The catch diode carries load current during the off-time. The average diode current is therefore dependent on the P-channel switch duty cycle. At high input voltages the diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short circuited. Under this condition the diode must safely handle IPEAK at close to 100% duty cycle. Therefore, it is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. Under normal load conditions, the average current conducted by the diode is: V −V  ID =  IN OUT  IOUT  VIN + VD  LTC1622 APPLICATIONS INFORMATION The allowable forward voltage drop in the diode is calculated from the maximum short-circuit current as: VF ≈ PD ISC(MAX )  1 ∆VOUT ≈ IRIPPLE  ESR +  8 fCOUT   where PD is the allowable power dissipation and will be determined by efficiency and/or thermal requirements. A fast switching diode must also be used to optimize efficiency. Schottky diodes are a good choice for low forward drop and fast switching times. Remember to keep lead length short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. CIN and COUT Selection In continuous mode, the source current of the P-channel MOSFET is a square wave of duty cycle (VOUT + VD)/ (VIN + VD). To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN Required IRMS ≈ IMAX [V (V OUT IN − VOUT VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT /2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet the size or height requirements in the design. Due to the high operating frequency of the LTC1622, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (∆VOUT) is approximated by: U W U U where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage since ∆IL increases with input voltage. The choice of using a smaller output capacitance increases the output ripple voltage due to the frequency dependent term, but can be compensated for by using capacitors of very low ESR to maintain low ripple voltage. The ITH pin OPTI-LOOP compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected. Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR (size) product of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications, multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS, AVX TPSV and KEMET T510 series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Sanyo POSCAP, Nichicon PL series and the Panasonic SP series. Low Supply Operation Although the LTC1622 can function down to 2V, the maximum allowable output current is reduced when VIN decreases below 3V. Figure 3 shows the amount of change as the supply is reduced down to 2V. Also shown in Figure 3 is the effect of VIN on VREF as VIN goes below 2.3V. Remember the maximum voltage on the ITH pin defines )] 1/ 2 9 LTC1622 APPLICATIONS INFORMATION 101 100 99 98 97 96 95 2.0 VREF NORMALIZED VOLTAGE (%) VITH 2.2 2.4 2.6 2.8 INPUT VOLTAGE (V) 3.0 1622 F03 Figure 3. Line Regulation of VREF and VITH the maximum current sense voltage that sets the maximum output current. Setting Output Voltage The LTC1622 develops a 0.8V reference voltage between the feedback (Pin 3) terminal and ground (see Figure 4). By selecting resistor R1, a constant current is caused to flow through R1 and R2 to set the output voltage. The regulated output voltage is determined by:  R2  VOUT = 0.8 1 +   R1  For most applications, a 30k resistor is suggested for R1. To prevent stray pickup, an optional 100pF capacitor is suggested across R1 located close to LTC1622. VOUT LTC1622 VFB R2 3 100pF R1 1622 F04 Figure 4. Setting Output Voltage Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what 10 U W U U is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (η1 + η2 + η3 + ...) where η1, η2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1622 circuits: 1) LTC1622 DC bias current, 2) MOSFET gate charge current, 3) I2R losses, 4) voltage drop of the output diode and 5) transition losses. 1. The VIN current is the DC supply current, given in the electrical characteristics, that excludes MOSFET driver and control currents. VIN current results in a small loss which increases with VIN. 2. MOSFET gate charge current results from switching the gate capacitance of the power MOSFET. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN which is typically much larger than the DC supply current. In continuous mode, IGATECHG = f(Qp). 3. I2R losses are predicted from the DC resistances of the MOSFET, inductor and current shunt. In continuous mode the average output current flows through L but is “chopped” between the P-channel MOSFET in series with RSENSE and the output diode. The MOSFET RDS(ON) plus RSENSE multiplied by duty cycle can be summed with the resistance of the inductor to obtain I2R losses. 4. The output diode is a major source of power loss at high currents and gets worse at high input voltages. The diode loss is calculated by multiplying the forward voltage drop times the diode duty cycle multiplied by the load current. For example, assuming a duty cycle of 50% with a Schottky diode forward voltage drop of 0.4V, the loss increases from 0.5% to 8% as the load current increases from 0.5A to 2A. 5. Transition losses apply to the external MOSFET and increase with higher operating frequencies and input voltages. Transition losses can be estimated from: LTC1622 APPLICATIONS INFORMATION Transition Loss = 3(VIN)2IO(MAX)CRSS(f) Other losses including CIN and COUT ESR dissipative losses, and inductor core losses, generally account for less than 2% total additional loss. Run/Soft-Start Function The RUN/SS pin is a dual purpose pin that provides the soft-start function and a means to shut down the LTC1622. Soft-start reduces input surge current from VIN by gradually increasing the internal current limit. Power supply sequencing can also be accomplished using this pin. An internal 2.5µA current source charges up an external capacitor CSS. When the voltage on the RUN/SS reaches 0.7V the LTC1622 begins operating. As the voltage on RUN/SS continues to ramp from 0.7V to 1.8V, the internal current limit is also ramped at a proportional linear rate. The current limit begins near 0A (at VRUN/SS = 0.7V) and ends at 0.1/RSENSE (VRUN/SS ≥ 1.8V). The output current thus ramps up slowly, reducing the starting surge current required from the input power supply. If the RUN/SS has been pulled all the way to ground, there will be a delay before the current limit starts increasing and is given by: tDELAY = 2.8 • 105 • CSS in seconds Pulling the RUN/SS pin below 0.4V puts the LTC1622 into a low quiescent current shutdown (IQ < 15µA). Foldback Current Limiting As described in the Output Diode Selection, the worstcase dissipation occurs with a short-circuited output when the diode conducts the current limit value almost continuously. To prevent excessive heating in the diode, foldback current limiting can be added to reduce the current in proportion to the severity of the fault. Foldback current limiting is implemented by adding diode DFB (1N4148 or equivalent) between the output and the ITH pin as shown in Figure 5. In a hard short (VOUT = 0V), the current will be reduced to approximately 50% of the maximum output current. LTC1622 R2 ITH VFB R1 VOUT U W U U + DFB 1622 F05 Figure 5. Foldback Current Limiting Design Example Assume the LTC1622 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to a minimum of 2.7V. Load current requirement is a maximum of 1.5A but most of the time it will be on standby mode, requiring only 2mA. Efficiency at both low and high load current is important. Output voltage is 2.5V. In the above application, Burst Mode operation is enabled by connecting Pin 5 to VIN. Maximum Duty Cycle = From Figure 2, SF = 57%. VOUT + VD = 93% VIN(MIN) + VD Use the curve of Figure 2 since the operating frequency is the free running frequency of the LTC1622. RSENSE = (15)(IOUT)(100) (15)(1.5A) SF = 0.57 = 0.0253Ω In the application, a 0.025Ω resistor is used. For the inductor, the required value is: LMIN =  2.5 + 0.3  4.2 − 2.5 = 1.33µH  0.036   4.2 + 0.3    550kHz   0.025  In the application, a 3.9µH inductor is used to reduce inductor ripple current and thus, output voltage ripple. For the selection of the external MOSFET, the RDS(ON) must be guaranteed at 2.5V since the LTC1622 has to work 11 LTC1622 APPLICATIONS INFORMATION down to 2.7V. Let’s assume that the MOSFET dissipation is to be limited to PP = 250mW and its thermal resistance is 50°C/W. Hence the junction temperature at TA = 25°C will be 37.5°C and δp = 0.005 (37.5 – 25) = 0.0625. The required RDS(ON) is then given by: layout diagram in Figure 6. Check the following in your layout: 1. Is the Schottky diode closely connected between ground at (–) lead of CIN and drain of the external MOSFET? 2. Does the (+) plate of CIN connect to the sense resistor as closely as possible? This capacitor provides AC current to the MOSFET. 3. Is the input decoupling capacitor (0.1µF) connected closely between VIN (Pin 8) and ground (Pin 6)? 4. Connect the end of RSENSE as close to VIN (Pin 8) as possible. The VIN pin is the SENSE + of the current comparator. 5. Is the trace from the SENSE – (Pin 1) to the Sense resistor kept short? Does the trace connect close to RSENSE? 6. Keep the switching node, SW, away from sensitive small signal nodes. 7. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1 and R2 must be connected between the (+) plate of COUT and signal ground. Optional capacitor C1 should be located as close as possible to the LTC1622. R1 and R2 should be located as close as possible to the LTC1622. R2 should connect to the output as close to the load as practicable. VIN CIN RDS(ON) ≅ DC IOUT ( ) (1+ δp) 2 PP = 0.11Ω The P-channel MOSFET requirement can be met by an Si6433DQ. The requirement for the Schottky diode is the most stringent when VOUT = 0V, i.e., short circuit. With a 0.025Ω RSENSE resistor, the short-circuit current through the Schottky is 0.1/0.025 = 4A. An MBRS340T3 Schottky diode is chosen. With 4A flowing through, the diode is rated with a forward voltage of 0.4V. Therefore, the worstcase power dissipated by the diode is 1.6W. The addition of DFB (Figure 5) will reduce the diode dissipation to approximately 0.8W. The input capacitor requires an RMS current rating of at least 0.75A at temperature, and COUT will require an ESR of 0.1Ω for optimum efficiency. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1622. These items are illustrated graphically in the 1 2 SENSE – ITH RITH CITH C1 LTC1622 6 3 GND VFB 4 RUN/ SS CSS SYNC/ 5 MODE QUIET SGND 1622 F06 R1 R2 BOLD LINES INDICATE HIGH CURRENT PATHS Figure 6. LTC1622 Layout Diagram (See PC Board Layout Checklist) 12 U W U U RSENSE + VIN PDRV 8 7 0.1µF M1 SW L1 VOUT + COUT LTC1622 TYPICAL APPLICATIONS LTC1622 1.8V/1.5A Regulator with Burst Mode Operation Disabled 1 2 R1 10K C3 220pF C4 560pF 3 SENSE – ITH VFB VIN PDRV 8 7 LTC1622 6 GND SYNC/ 5 MODE 4 RUN/ SS C1: AVX TPSD476M016R0150 C2: AVX TPSD227M006R0100 L1: MURATA LQN6C3R3 R2: DALE WSL-1206 0.025Ω U1: INTERNATIONAL RECTIFIER FETKY TM IRF7422D2 LTC1622 2.5V/2A Regulator with Burst Mode Operation Enabled VIN 3.3V TO 8.5V 1 2 3 R1 10k C3 220pF SENSE – ITH VIN PDRV 8 7 D1 LTC1622 6 VFB GND SYNC/ 5 MODE 4 RUN/ SS C4 560pF C1: AVX TPSD476M016R0150 C2: SANYO POSCAP 6TPA47M D1: MOTOROLA MBR320T3 + VIN 2.5V TO 8.5V U C1 47µF 16V R2 0.025Ω U1 1 2 3 4 8 7 6 5 L1 3.3µH R3 93.1k R4 75k VOUT 1.8V 1.5A + C2 220µF 6V 1622 TA01 + R2 0.02Ω M1 C1 47µF 16V ×2 L1 4.7µH R3 158k R4 75k VOUT 2.5V 2A + C2 150µF 6V ×2 1622 TA02 L1: COILCRAFT D03316-472 M1: SILICONIX Si3443DV R2: DALE WSL-2010 0.02Ω FETKY is a trademark of International Rectifier Corporation. 13 LTC1622 TYPICAL APPLICATIONS U 1 2 1 2 LTC1622 2.5V/3A Regulator with External Frequency Synchronization VIN 3.3V TO 8.5V SENSE – ITH VIN 8 R2 0.01Ω + R1 10k C3 220pF 7 PDRV LTC1622 6 3 VFB GND 4 RUN/ SS C4 560pF SYNC/ 5 MODE 650kHz 1.5VP-P C1 47µF 16V ×2 M1 D1 L1 4.7µH C2 100µF 6.3V ×2 R3 158k R4 75k VOUT 2.5V 3A + C1: AVX TPSD476M016R0150 C2: AVX TPSD107M010R0065 D1: MOTOROLA MBR320T3 L1: COILCRAFT D03316-472 M1: SILICONIX Si3443DV R2: DALE WSL-2512 0.01Ω 1622 TA03 Zeta Converter with Foldback Current Limit D2 1N4818 SENSE – ITH VIN 8 R2 0.04Ω Si3441DV + R1 47k C3 470pF C4 0.1µF 7 PDRV LTC1622 6 3 GND VFB 4 RUN/ SS SYNC/ 5 MODE L1A 6.2µH C1 47µF 16V ×2 L1B 6.2µH D1 VIN 2.5V TO 8.5V 47µF 16V + + C2 100µF 10V R3 232k R4 75k VOUT 3.3V 1622 TA04 C1: AVX TPSD476M016R0150 C2: AVX TPSD107M010R0080 D1: MOTOROLA MBRS320T3 L1B L1A, L1B: BH ELECTRONICS BH511-1012 R2: DALE WSL-1206 0.04Ω 3 4 2 L1A TOP VIEW • 1 VIN (V) 2.5 3.3 5.0 6.0 8.4 IOUT(MAX) (A) 0.45 0.70 0.95 1.00 1.05 14 LTC1622 PACKAGE DESCRIPTION U Dimensions in inches (millimeters) unless otherwise noted. MS8 Package 8-Lead Plastic MSOP (LTC DWG # 05-08-1660) 0.118 ± 0.004* (3.00 ± 0.102) 8 76 5 0.193 ± 0.006 (4.90 ± 0.15) 0.118 ± 0.004** (3.00 ± 0.102) 1 0.040 ± 0.006 (1.02 ± 0.15) 0.007 (0.18) 0.021 ± 0.006 (0.53 ± 0.015) 0° – 6° TYP SEATING PLANE 0.012 (0.30) 0.0256 REF (0.65) BSC 23 4 0.034 ± 0.004 (0.86 ± 0.102) 0.006 ± 0.004 (0.15 ± 0.102) MSOP (MS8) 1098 * DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 8 7 6 5 0.228 – 0.244 (5.791 – 6.197) 0.150 – 0.157** (3.810 – 3.988) 1 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0°– 8° TYP 2 3 4 0.053 – 0.069 (1.346 – 1.752) 0.004 – 0.010 (0.101 – 0.254) 0.014 – 0.019 (0.355 – 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.016 – 0.050 (0.406 – 1.270) 0.050 (1.270) BSC SO8 1298 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LTC1622 TYPICAL APPLICATION Small Footprint 3.3V/1A Regulator + VIN 3.3V TO C1 8.5V 10µF 16V CERAMIC EFFICIENCY (%) 1 2 SENSE – ITH VIN 8 R1 10k C3 220pF 7 PDRV LTC1622 3 6 VFB GND 4 RUN/ SYNC/ 5 MODE SS C4 560pF C1: MURATA CERAMIC GRM235Y5V106Z C2: SANYO POSCAP 6TPA47M D1: MOTOROLA MBRS120LT3 Boost Converter 3.3V/2.5A 1 2 C3 470pF R1 33k C5 150pF SENSE – ITH VIN 8 VIN 3.3V L1 4.6µH M1 D1 Si6801DQ 4 RUN/ SS C4 0.1µF SYNC/ 5 MODE R3 105k R4 20k VOUT 5V 2.5A EFFICIENCY (%) PDRV LTC1622 6 3 VFB GND 7 C1, C2: SANYO POSCAP TPB SERIES D1: MOTOROLA MBRD835L L1: SUMIDA CEP123-4R6 RELATED PARTS PART NUMBER LTC1147 Series LT1375/LT1376 LTC1438/LTC1439 LTC1474/LTC1475 LTC1624 LTC1626 LTC1627/LTC1707 LTC1628 LTC1772 LTC1735 DESCRIPTION High Efficiency Step-Down Switching Regulator Controllers 1.5A, 500kHz Step-Down Switching Regulators Dual, Low Noise, Synchronous Step-Down Converters Low Quiescent Current Step-Down DC/DC Converters High Efficiency SO-8 N-Channel Switching Regulator Controller Low Voltage, High Efficiency Step-Down DC/DC Converter Low Voltage, Monolithic Synchronous Step-Down Regulator Dual High Efficiency 2-Phase Step-Down Controller SOT-23 Current Mode Step-Down Controller High Efficiency, Low Noise Synchronous Switching Controller COMMENTS 100% DC, 3.5V ≤ VIN ≤ 16V, HV Version Has 20VIN High Frequency, Small Inductor, High Efficiency 24-Pin Narrow SSOP, 3.5V ≤ VIN ≤ 36V Multiple Output Capability, 3.5V ≤ VIN ≤ 36V Monolithic, MSOP, IOUT = 10µA 8-Pin N-Channel Drive, 3.5V ≤ VIN ≤ 36V Monolithic, Constant Off-Time, 2.5V ≤ VIN ≤ 6V Low Supply Voltage Range: 2.65V to 8V, 0.5A Antiphase Drive, 3.5V ≤ VIN ≤ 36V, Protection 6-Lead SOT-23, 2.5V ≤ VIN ≤ 9.8V, 550kHz Burst Mode Operation, Protection, 3.5V ≤ VIN ≤ 36V LTC1436/LTC1436-PLL High Efficiency, Low Noise, Synchronous Step-Down Converters 16 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 q FAX: (408) 434-0507 q www.linear-tech.com U D1 Efficiency vs Load Current 100 VIN = 3.5V 90 VIN = 4.2V VIN = 6V 70 R2 0.025Ω M1 L1 2.2µH R3 232k C2 47µF 6V VOUT 3.3V 1A 80 + R4 75k 60 VOUT = 3.3V RSENSE = 0.025Ω 1622 TA05 50 1 10 100 LOAD CURRENT (mA) 1000 1622 TA05b L1: COILCRAFT D01608C-222 M1: SILICONIX Si3443DY R2: DALE WSL-2010 0.025Ω Efficiency vs Load Current With LTC1622 Configured as Boost Converter 100 VOUT = 5V RSENSE = 0.015Ω 90 VIN = 3.3V 80 + C1 100µF 10V C6 0.1µF R2 0.015Ω + C2 220µF 10V ×2 70 60 1622 TA06a M1: SILICONIX Si3442DV R2: DALE WS-L2512 0.015Ω 50 0.001 0.01 0.1 LOAD CURRENT (mA) 1 1622 TA06b 1622f LT/TP 0100 4K • PRINTED IN USA © LINEAR TECHNOLOGY CORPORATION 1998