LTC1698

LTC1698 Isolated Secondary Synchronous Rectifier Controller FEATURES s s s s s s s s s s s DESCRIPTIO High Efficiency Over Wide Load Current Range ±0.8% Output Voltage Accuracy Dual N-Channel MOSFET Synchronous Drivers Pulse Transformer Synchronization Optocoupler Feedback Driver Programmable Current Limit Protection ±5% Margin Output Voltage Adjustment Adjustable Overvoltage Fault Protection Power Good Flag Auxiliary 3.3V Logic Supply Available in 16-Lead SSOP and SO Packages The LTC ®1698 is a precision secondary-side forward converter controller that synchronously drives external N-channel MOSFETs. It is designed for use with the LT®3781 primary-side synchronous forward converter controller to create a completely isolated power supply. The LT3781 synchronizes the LTC1698 through a small pulse transformer and the LTC1698 drives a feedback optocoupler to close the feedback loop. Output accuracy of ± 0.8% and high efficiency over a wide range of load currents are obtained. The LTC1698 provides accurate secondary-side current limit using an external current sense resistor. The input voltage at the MARGIN pin provides ± 5% output voltage adjustment. A power good flag and overvoltage input are provided to ensure proper power supply conditions. An auxiliary 3.3V logic supply is included that supplies up to 10mA of output current. , LTC and LT are registered trademarks of Linear Technology Corporation. APPLICATIO S s s s s s 48V Input Isolated DC/DC Converters Isolated Telecommunication Power Systems Distributed Power Step-Down Converters Industrial Control Systems Automotive and Heavy Equipment TYPICAL APPLICATIO VIN 36V to 72V L1 + Q1 D1 VDD BIAS COUT + 1 10 VDD CG FG ISNS R1 R2 T1 Q4 Q2 RPRISEN RSECSEN 11 CSG TG BG CSYNC 15 T2 D2 Q3 2 16 12 • PWRGD 8 VFB CFB VCOMP 6 LTC1698 ISNSGND SYNC ICOMP OVPIN 13 9 RCILM • SG RSYNC MARGIN 7 VMARGIN VAUX 3.3V 10mA LT3781 RK REF VFB RE CK 5 OPTODRV PGND 3 VAUX GND 4 14 O.1µF 1681 F01 VC RF + – CF ISOLATION BOUNDARY PLEASE REFER TO FIGURE 12 IN THE TYPICAL APPLICATIONS SECTION FOR THE COMPLETE 3.3V/15A APPLICATION SCHEMATIC Figure 1. Simplified 2-Transistor Isolated Forward Converter 1698f U VOUT RC CC R5 CCILM R4 U U • • 1 LTC1698 ABSOLUTE MAXIMUM RATINGS (Note 1) PACKAGE/ORDER INFORMATION TOP VIEW VDD CG PGND GND OPTODRV VCOMP MARGIN VFB 1 2 3 4 5 6 7 8 16 FG 15 SYNC 14 VAUX 13 ICOMP 12 ISNS 11 ISNSGND 10 PWRGD 9 OVPIN VDD, PWRGD ....................................................... 13.2V Input Voltage MARGIN, VFB, OVPIN, ISNSGND, ISNS ... – 0.3V to 5.3V SYNC ..................................................... – 14V to 14V Output Voltage VCOMP, ICOMP (Note 2) ......................... – 0.3V to 5.3V Power Dissipation .............................................. 500mW Operating Temperature Range LTC1698E (Note 3) ............................ – 40°C to 85°C LTC1698I ........................................... – 40°C to 85°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER LTC1698EGN LTC1698ES LTC1698IGN LTC1698IS GN PART MARKING 1698 1698I GN PACKAGE S PACKAGE 16-LEAD PLASTIC SSOP 16-LEAD PLASTIC SO TJMAX = 125°C, θJA = 130°C/W (GN) TJMAX = 125°C, θJA = 110°C/W (SO) Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS SYMBOL VDD VUVLO IVDD PARAMETER Supply Voltage Undervoltage Lockout VDD Supply Current The q indicates specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VDD = 8V, unless otherwise noted. (Note 4) CONDITIONS q MIN 6 TYP 8 4 MAX 12.6 UNITS V V VFB, OVPIN, VISNS, VISNSGND = 0V, CFG = CCG = 1000pF, CVAUX = 0.1µF, VSYNC = 0V fSYNC = 100kHz (Note 5) q 1.8 5.0 1.223 1.215 1.233 1.233 0.05 1.65 16.5 4 MARGIN and Error Amplifier VFB IVFB VMARGIN RMARGIN ∆VFB GERR BWERR VCLAMP IVCOMP OPTODRV GOPTO BWOPTO VOPTOHIGH IOPTOSC Opto Driver DC Gain Opto Driver Unity-Gain Bandwidth Opto Driver Output High Voltage Opto Driver Output Short-Circuit Current OVPIN, VISNS, VISNSGND = 0V No Load (Note 6) VFB, OVPIN, VISNSGND = 0V, VISNS = – 50mV, IOPTODRV = –10mA OVPIN, VISNSGND, VISNS = 0V, VFB = 1.233V q q q Feedback Voltage Feedback Input Current MARGIN Voltage MARGIN Input Resistance Feedback Voltage Adjustment Error Amplifier Open-Loop DC Gain Error Amplifier Unity-Gain Bandwidth Error Amplifier Output Clamp Voltage Error Amplifier Source Current Error Amplifier Sink Current MARGIN = Open, VCOMP = 1V (Note 7) q 1.243 1.251 1 VFB = 1.233V MARGIN = Open VMARGIN = 3.3V VMARGIN = 0V VCOMP = 0.8V to 1.2V, Load = 2kΩ, 100pF No Load (Note 6) VFB = 0V VFB = 0V VFB = 5V, VCOMP = 1.233V q q q q 4 –6 65 5 –5 90 2 2 6 –4 q q 3 4.75 – 25 7 5 1 – 10 5.25 4 – 50 5 – 25 – 10 2 U W U U WW W mA mA V V µA V kΩ % % dB MHz V mA mA V/V MHz V mA 1698f LTC1698 ELECTRICAL CHARACTERISTICS SYMBOL PARAMETER VAUX VAUX Auxiliary Supply Voltage CONDITIONS The q indicates specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VDD = 8V, unless otherwise noted. (Note 4) MIN q q q q TYP 3.320 0.05 0.05 MAX 3.465 1 1 – 23.0 – 22.5 – 120 – 80 280 370 5 UNITS V µA µA mV mV µA µA µA µA millimho dB CVAUX = 0.1µF, ILOAD = 0mA to 10mA, VDD = 7V to 12.6V VISNSGND = 0V VISNS = 0V VICOMP = 2.5V, VISNSGND = 0V VISNSGND = 0V, VISNS = – 0.3V, VICOMP = 2.5V (Note 8) 3.135 Current Limit Amplifier IISNSGND ISNSGND Input Current IISNS ISNS Input Current VILIMTH Current Limit Threshold (VISNS – VISNSGND) IICOMP ICOMP Source Current ICOMP Sink Current gmILIM GICOMP Current Limit Amplifier Transconductance Current Limit Amplifier Open-Loop DC Gain Percent Below VFB Power Good Sink Current Power Good Output Low Voltage – 27.0 – 27.5 – 280 – 370 120 80 2.2 48 – 25 – 25 – 200 – 200 200 200 3.5 60 q VISNSGND = 0V, VISNS = 0.3V, VICOMP = 2.5V (Note 8) q VISNSGND = 0V, VICOMP = 2.5V, IICOMP = ±10µA VICOMP = 2.5V, No Load q q PWRGD and OVP Comparators VPWRGD IPWRGD VOL VFB ↓, MARGIN = Open (Note 9) VFB = 2V VFB = 0V IPWRGD = 3mA, VFB = 0V VFB = VISNS = VISNSGND = 0V, OVPIN ↑ (Note 9) VOVPIN = 1.233V VFB ↑ VFB ↓ VOVPIN ↑, COPTODRV = 0.1µF q q q q q q q q q q q –9 10 –6 –3 10 0.4 % µA mA V V µA ms ms µs V V µA kHz ns ns ns VOVPREF OVPIN Threshold IOVPIN OVPIN Input Bias Current tPWRGD tOVP Power Good Response Time Power Bad Response Time Overvoltage Response Time 1.18 1 0.5 1.233 0.1 2 1 5 1.28 1 5 2.5 20 2.2 –1 50 400 90 40 SYNC and Drivers VPT SYNC Input Positive Threshold VNT ISYNC fSYNC td tSYNC tr, tf tDDIS SYNC Input Negative Threshold SYNC Input Current SYNC Frequency Range SYNC Input to Driver Output Delay Minimum SYNC Pulse Width Driver Rise and Fall Time Driver Disable Time-Out VSYNC = ± 10V CFG = CCG = 1000pF, VSYNC = ± 5V CFG = CCG = 1000pF, fSYNC = 100kHz, VSYNC = ± 5V fSYNC = 100kHz, VSYNC = ± 10V (Note 6) CFG = CCG = 1000pF, fSYNC = 100kHz, VSYNC = ± 5V, 10% to 90% CFG = CCG = 1000pF, fSYNC = 100kHz, VSYNC = ± 5V Measured from CG ↑ (Note 10) 1 – 2.2 50 1.6 –1.6 1 40 q q q q q 75 10 q 10 15 20 µs Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. All voltages refer to GND. Note 2: The LTC1698 incorporates a 5V linear regulator to power internal circuitry. Driving these pins above 5.3V may cause excessive current flow. Guaranteed by design and not subject to test. Note 3: The LTC1698E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the – 40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. For guaranteed performance to specifications over the –40°C to 85°C range, the LTC1698I is available. Note 4: All currents into device pins are positive; all currents out of the device pins are negative. All voltages are referenced to ground unless otherwise specified. For applications with VDD < 7V, refer to the Typical Performance Characteristics. Note 5: Supply current in active operation is dominated by the current needed to charge and discharge the external FET gates. This will vary with the LTC1698 operating frequency, supply voltage and the external FETs used. Note 6: This parameter is guaranteed by correlation and is not tested. Note 7: VFB is tested in an op amp feedback loop which servos VFB to the internal bandgap voltage. Note 8: The current comparator output current varies linearly with temperature. Note 9: The PWRGD and OVP comparators incorporate 10mV of hysteresis. Note 10: The driver disable time-out is proportional to the SYNC period within the frequency synchronization range. 1698f 3 LTC1698 TYPICAL PERFOR A CE CHARACTERISTICS VFB vs Temperature 1.248 VDD = 8V 1.248 1.242 1.236 VFB (V) VFB (V) VFB (V) 1.230 1.224 1.218 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G01 ISNS Threshold vs Temperature –22.5 –23.0 –23.5 VDD = 8V –22.5 –23.0 –23.5 ISNS THRESHOLD (mV) ISNS THRESHOLD (mV) –24.5 –25.0 –25.5 –26.0 –24.5 –25.0 –25.5 –26.0 gmILIM (millimho) –24.0 –26.5 –27.0 –27.5 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G04 OVPIN Threshold vs Temperature 1.28 VDD = 8V 1.26 1.26 OVPIN THRESHOLD (V) POWER GOOD THRESHOLD (V) OVPIN THRESHOLD (V) 1.24 1.22 1.20 1.18 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G07 4 UW VFB vs VDD TA = 25°C 1.295 1.282 VFB vs VMARGIN VDD = 8V TA = 25°C 5 4 3 2 1 0 –1 –2 –3 –4 0 0.33 0.66 0.99 1.32 1.65 1.98 2.31 2.64 2.97 3.3 VMARGIN (V) 1698 G03 1.242 1.270 1.258 1.236 1.245 1.233 1.221 1.208 ∆VFB (%) 1.230 1.224 1.196 1.184 1.218 5 6 7 8 9 10 VDD (V) 11 12 13 14 1.171 –5 1698 G02 ISNS Threshold vs VDD TA = 25°C 5.0 4.6 4.2 3.8 3.4 3.0 2.6 Current Limit Amplifier gm vs Temperature VDD = 8V –24.0 –26.5 –27.0 –27.5 5 6 7 8 9 10 VDD (V) 11 12 13 14 2.2 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G06 1698 G05 OVPIN Threshold vs VDD 1.28 TA = 25°C 1.196 Power Good Threshold vs Temperature VDD = 8V –3.0 1.181 –4.2 1.24 1.166 –5.4 ∆VFB (%) 1.22 1.152 –6.6 1.20 1.137 –7.8 1.18 5 6 7 8 9 10 VDD (V) 11 12 13 14 1.122 –50 –25 0 –9.0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G09 1698 G08 1698f LTC1698 TYPICAL PERFOR A CE CHARACTERISTICS VAUX vs Temperature 3.465 VDD = 8V 3.424 ILOAD = 0mA 3.383 VAUX (V) VAUX (V) 3.300 3.259 3.218 3.176 3.135 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G10 3.300 3.259 3.218 3.176 3.135 5 6 7 8 9 10 VDD (V) 11 12 13 14 VAUX (V) 3.341 VAUX Short-Circuit Current vs Temperature 0 VDD = 8V 0 VAUX SHORT-CIRCUIT CURRENT (mA) VAUX SHORT-CIRCUIT CURRENT (mA) –10 –10 OPTO DRIVER OUTPUT VOLTAGE (V) –20 –30 –40 –50 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G13 Maximum OPTO Driver Output Voltage vs Load Current MAXIMUM OPTO DRIVER OUTPUT VOLTAGE (V) MAXIMUM OPTO DRIVER OUTPUT VOLTAGE (V) OPTO DRIVER SHORT-CIRCUIT CURRENT (mA) 8 6 4 2 TA = 25°C VCOMP = 0V 0 1 2 345678 LOAD CURRENT (mA) 9 10 0 UW VDD = 10V VDD = 8V VDD = 7V VDD = 6V VDD = 5V VAUX vs Line Voltage 3.465 VDD = 8V 3.424 ILOAD = 0mA 3.383 3.341 3.465 VAUX vs Load Current VDD = 8V 3.424 TA = 25°C 3.383 3.341 3.300 3.259 3.218 3.176 3.135 0 1 2 345678 LOAD CURRENT (mA) 9 10 1698 G11 1698 G12 VAUX Short-Circuit Current vs VDD TA = 25°C 3.030 3.024 3.018 3.012 3.006 3.000 2.994 2.988 2.982 2.976 2.970 Opto Driver Load Regulation VDD = 8V TA = 25°C 1.0 0.8 0.6 0.4 PERCENT (%) –20 0.2 0 –0.2 –0.4 –0.6 –0.8 0 1 2 345678 LOAD CURRENT (mA) 9 10 –1.0 –30 –40 –50 5 6 7 8 10 VDD (V) 9 11 12 13 14 1698 G14 1698 G15 Maximum OPTO Driver Output Voltage vs Temperature 8 VDD = 10V Opto Driver Short-Circuit Current vs Temperature –10 VDD = 8V –15 VOPTODRV = 1.233V –20 –25 –30 –35 –40 –45 –50 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G16 6 VDD = 8V VDD = 7V 4 VDD = 6V VDD = 5V 2 VCOMP = 0V IOPTODRV = – 10mA 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G23 1698 G22 1698f 5 LTC1698 TYPICAL PERFOR A CE CHARACTERISTICS IVDD vs SYNC Frequency 50 45 40 35 IVDD (mA) SYNC POSITIVE THRESHOLD (V) SYNC POSITIVE THRESHOLD (V) VDD = 8V TA = 25°C CFG = CCG = 4700pF CFG = CCG = 3300pF 30 25 20 15 10 5 0 CFG = CCG = 2200pF CFG = CCG = 1000pF 50 100 150 200 250 300 350 400 450 500 fSYNC (kHz) 1698 G19 Opto Driver Short-Circuit Current vs VDD OPTO DRIVER SHORT-CIRCUIT CURRENT (mA) 0 TA = 25°C VOPTODRV = 1.233V 20 18 16 14 –10 IVDD (mA) 10 8 6 CFG = CCG = 2200pF CFG = CCG = 1000pF VUVLO (V) –20 –30 –40 –50 5 6 7 8 10 VDD (V) 9 11 12 13 14 Driver Rise, Fall and Propagation Delay vs Driver Load 90 80 70 60 TIME (ns) tf CG, FG tPHL tr VDD = 8V TA = 25°C CG, FG tPLH 90 80 70 60 DRIVER DISABLE TIME-OUT tDISS (µs) t d (ns) 50 40 30 20 10 0 0 2000 6000 4000 DRIVER LOAD (pF) 6 UW 1698 G17 SYNC Positive Threshold vs Temperature 2.25 VDD = 8V SYNC Positive Threshold vs VDD 2.20 TA = 25°C 2.00 1.96 1.75 1.72 1.50 1.48 1.25 1.24 1.00 –50 –25 1.00 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G20 5 6 7 8 9 10 VDD (V) 11 12 13 14 1698 G21 IVDD vs VDD TA = 25°C fSYNC = 100kHz 5 CFG = CCG = 4700pF 4 Undervoltage Lockout Threshold vs Temperature CFG = CCG = 3300pF 12 3 2 4 2 0 5 6 1 7 8 9 10 VDD (V) 11 12 13 14 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G24 1698 G18 SYNC Input to Driver Output Delay vs Temperature VDD = 8V CCG = CFG = 1000pF fSYNC = 100kHz 30 25 20 15 10 Driver Disable Time-Out vs SYNC Frequency VDD = 8V TA = 25°C tDISS × fSYNC 2.2 NORMALIZED DRIVER DISABLE TIME-OUT tDISS × fSYNC 2.0 1.8 1.6 1.4 tDISS 1.2 1.0 50 100 150 200 250 300 350 400 450 500 fSYNC (kHz) 1698 G27 50 40 30 20 10 CG, FG tPLH CG, FG tPHL 5 0 8000 10000 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1698 G26 1698 G25 1698f LTC1698 PI FU CTIO S VDD (Pin 1): Power Supply Input. For isolated applications, a simple rectifier from the power transformer is used to power the chip. This pin powers the opto driver, the VAUX supply and the FG and CG drivers. An internal 5V regulator powers the remaining circuitry. VDD requires an external 4.7µF bypass capacitor. CG (Pin 2): Catch Gate Driver. If SYNC slews positive, CG pulls high to drive an external N-channel MOSFET. CG draws power from the VDD pin and swings between VDD and PGND. PGND (Pin 3): Power Ground. Connect PGND to a low impedance ground plane in close proximity to the ground terminal of the external current sensing resistor. GND (Pin 4): Logic and Signal Ground. GND is referenced to the internal low power circuitry. Careful board layout techniques must be used to prevent corruption of signal ground reference. Connect GND and PGND together directly at the LTC1698. OPTODRV (Pin 5): Optocoupler Driver Output. This pin drives a ground referenced optocoupler through an external resistor. If VFB is low, OPTODRV pulls low. If VFB is high, OPTODRV pulls high. This optocoupler driver has a DC gain of 5. During overvoltage or overcurrent conditions, OPTODRV pulls high. The output is capable of sourcing 10mA of current and will drive an external 0.1µF capacitive load and is short-circuit protected. VCOMP (Pin 6): Error Amplifier Output. This error amplifier is able to drive more than 2kΩ and 100pF of load. The internal diode connected from VFB to VCOMP reduces OPTODRV recovery time under start-up conditions. MARGIN (Pin 7): Current Input to Adjust the Output Voltage Linearly. The MARGIN pin connects to an internal 16.5k resistor. The other end of this resistor is regulated to 1.65V. Connecting MARGIN to a 3.3V logic supply sources 100µA of current into the chip and moves the output voltage 5% higher. Connecting MARGIN to 0V sinks 100µA out of the pin and moves the regulated output voltage 5% lower. The MARGIN pin voltage does not affect the PWRGD and OVPIN trip points. VFB (Pin 8): Feedback Voltage. VFB senses the regulated output voltage through an external resistor divider. The VFB pin is servoed to the reference voltage of 1.233V under closed-loop conditions. An RC network from VFB to VCOMP compensates the feedback loop. If VFB goes low, VCOMP pulls high and OPTODRV goes low. OVPIN (Pin 9): Overvoltage Input. OVPIN is a high impedance input to an internal comparator. The threshold of this comparator is set to 1.233V. If the OVPIN potential is higher than the threshold voltage, OPTODRV pulls high immediately. Use an external RC lowpass filter to prevent noisy signals from triggering this comparator. PWRGD (Pin 10): Power Good Output. This is an opendrain output. PWRGD floats if VFB is above 94% of the nominal value for more than 2ms. PWRGD pulls low if VFB is below 94% of the nominal value for more than 1ms. The PWRGD threshold is independent of the MARGIN pin potential. ISNSGND (Pin 11): Current Sense Ground. Connect to the positive side of the sense resistor, normally grounded. ISNS (Pin 12): Current Sense Input. Connect to the negative side of the sense resistor through an external RC lowpass filter. This pin normally sees a negative voltage, which is proportional to the average load current. If current limit is exceeded, OPTODRV pulls high. ICOMP (Pin 13): Current Amplifier Output. An RC network at this pin compensates the current limit feedback loop. Referencing the RC to VOUT controls output voltage overshoot on start-up. This pin can float if current limit loop compensation is not required. VAUX (Pin 14): Auxiliary 3.3V Logic Supply. This pin requires a 0.1µF or greater bypass capacitor. This auxiliary power supply can power external devices and sources 10mA of current. Internal current limiting is provided. SYNC (Pin 15): Drivers Synchronization Input. A negative voltage slew at SYNC forces FG to pull high and CG to pull low. A positive voltage slew at SYNC resets the FG pin and CG pulls high. If SYNC loses its synchronization signal for more than the driver disable time-out interval, both the forward and catch drivers output are forced low. The SYNC circuit accepts pulse and square wave signals. The minimum pulse width is 75ns. The synchronization frequency range is between 50kHz to 400kHz. FG (Pin 16): Forward Gate Driver. If SYNC slews negative, FG goes high. FG draws power from VDD and swings between VDD and PGND. 1698f U U U 7 LTC1698 BLOCK DIAGRA 14 15 7 5 10 VFB 0.94VREF OPERATIO (Refer to Block Diagram) The LTC1698 is a secondary-side synchronous rectifier controller designed to work with the LT3781 primary-side synchronous controller chip to form an isolated synchronous forward converter. This chip set uses a dual transistor forward topology that is predominantly used in distributed power supply systems where isolated low voltages are needed to power complex electronic equipment. The primary stage is a current mode, fixed frequency forward converter and provides the typical PWM operation. A power transformer is used to provide the functions of input/output isolation and voltage step-down to achieve the required low output voltage. Instead of using typical 8 + – W 1 VDD VAUX AUX GEN VCC GEN VCC SYNC IN CG MARGIN RMARGIN I-TO-V CONVERTER 2 FG 16 SYNC BANDGAP VREF ± 5% VREF + OPTODRV OPTO – 100k 20k ERR + – VFB VCOMP 8 6 11 PWRGD MPWRGD + MILIM RILIM 3k PWRGD ROVP 3k OVP ILIM 25mV ISNSGND – + ISNS ICOMP 12 13 + – VREF OVPIN 9 1698 BD U Schottky diodes, synchronous rectification on the secondary offers isolation with high efficiency. It supplies high power without the need of bulky heat sinks, which is often a problem in any space constrained application. The LTC1698 not only provides synchronous drivers for the external MOSFETs, it comes with other housekeeping functions performed on the secondary side of the power supply, all within a single integrated controller. Figure 1 shows the typical chip-set application. Upon power up, the LTC1698’s VDD input is low, the gate drivers TG and BG are both at the ground potential. The secondary forward and 1698f LTC1698 OPERATIO catch MOSFETs Q3 and Q4 are off. As soon as transistors Q1 and Q2 turn on, the flux in the power transformer T1 forces the body diodes of Q3 and Q4 to conduct, and the whole circuit starts like a conventional forward converter. At the same time, the LTC1698 VDD potential ramps up quickly through the VDD bias circuitry. Once the VDD voltage exceeds 4.0V, the LTC1698 enables its drivers and enters synchronous operation. The pulse transformer T2 synchronizes the primary and secondary MOSFET drivers. In a typical conversion cycle, the primary MOSFETs Q1 and Q2 turn on simultaneously. SG goes low and generates a negative spike at the LTC1698 SYNC input through the pulse transformer. The LTC1698 forces FG to turn on and CG to turn off. Power is delivered to the load through the transformer T1 and the inductor L1. At the beginning of the next phase in which Q1 and Q2 turn off, SG goes high, SYNC sees a positive spike, the MOSFET Q3 shuts off, Q4 conducts and allows continuous current to flow through the inductor L1. The capacitor COUT filters the switching waveform to provide a steady DC output voltage for the load. The LTC1698 error amplifier ERR senses the output voltage through an external resistor divider and regulates the VFB pin potential to the 1.233V internal bandgap voltage. An external RC network across the VFB and VCOMP pins frequency compensates the error amplifier feedback. The opto driver amplifies the voltage difference between the VCOMP pin and the bandgap potential, driving the external optocoupler diode with an inverting gain of 5. The optocoupler feeds the amplified output error signal to the primary controller and closes the forward converter voltage feedback loop. Under start-up conditions, the internal diode across the LTC1698 error amplifier clamps the VCOMP pin. This speeds up the opto driver recovery time by reducing the negative slew rate excursion at the COMP pin. The forward converter output voltage can be easily adjusted. The potential at the MARGIN pin is capable of U (Refer to Block Diagram) forcing the error amplifier reference voltage to move linearly by ± 5%. The internal RMARGIN resistor converts the MARGIN voltage to a current and linearly controls the offset of the error amplifier. Connecting the MARGIN pin to 3.3V increases the VFB voltage by 5%, and connecting the MARGIN pin to 0V reduces VFB by 5%. With the MARGIN pin floating, the VFB voltage is regulated to the internal bandgap voltage. The current limit transconductance amplifier ILIM provides the secondary side average current limit function. The average voltage drops across the RSECSEN resistor is sensed and compared to the – 25mV threshold set by the internal ILIM amplifier. Once ILIM detects high output current, the current amplifier output pulls high, overrides the error amplifier, injects more current into the photo diode and forces a lower duty cycle. An RC network connected to the ICOMP pin is used to stabilize the secondary current limit loop. Alternatively, if only overcurrent fault protection is required, ICOMP can float. If under abnormal conditions the feedback path is broken, OVPIN provides another route for overvoltage fault protection. If the voltage at OVPIN is higher than the bandgap voltage, the OVP comparator forces OPTODRV high immediately. A simple external RC filter prevents a momentary overshoot at OVPIN from triggering the OVP comparator. Short OVPIN to ground if this pin is not used. The LTC1698 provides an open-drain PWRGD output. If VFB is less than 94% of its nominal value for more than 1ms, the PWRGD comparator pulls the PWRGD pin low. If VFB is higher than 94% of its nominal value for more than 2ms, the transistor MPWRGD shuts off, and an external resistor pulls the PWRGD pin high. The LTC1698 provides an auxiliary 3.3V logic power supply. This auxiliary power supply is externally compensated with a minimum 0.1µF bypass capacitor. It supplies up to 10mA of current to any external devices. 1698f 9 LTC1698 APPLICATIO S I FOR ATIO Undervoltage Lockout In UVLO (low VDD voltage) the drivers FG and CG are shut off and the pins OPTODRV, VAUX, PWRGD and ICOMP are forced low. The LTC1698 allows the bandgap and the internal bias currents to reach their steady-state values before releasing UVLO. Typically, this happens when VDD reaches approximately 4.0V. Beyond this threshold, the drivers start switching. The OPTODRV, VAUX, PWRGD and ICOMP pins return to their normal values and the chip is fully functional. However, if the VDD voltage is less than 7V, the OPTODRV and VAUX current sourcing capabilities are limited. See the OPTO driver graphs in the Typical Performance Characteristics section. VDD Regulator The bias supply for the LTC1698 is generated by peak rectifying the isolated transformer secondary winding. As shown in Figure 2, the zener diode Z1 is connected from base of Q5 to ground such that the emitter of Q5 is regulated to one diode drop below the zener voltage. RZ is selected to bring Z1 into conduction and also provide base current to Q5. A resistor (on the order of a few hundred ohms), in series with the base of Q5, may be required to surpress high frequency oscillations depending on Q5’s selection. A power MOSFET can also be used by increasing the zener diode value to offset the drop of the gate-tosource voltage. VDD supply current varies linearly with the supply voltage, driver load and clock frequency. A 4.7µF bypass capacitor for the VDD supply is sufficient for most applications. This capacitor must be large enough to provide a stable DC voltage to meet the LTC1698 VDD VSECONDARY 1Ω D3 RZ 2k 0.47µF Z1 10V RB* Q5 FZT690 VDD 4.7µF FG CG *RB IS OPTIONAL, SEE TEXT 1698 F02 Figure 2. VDD Regulator 10 U supply requirement. Under start-up conditions, it must be small enough to power up instantaneously, enabling the LTC1698 to regulate the feedback loop. Using a larger capacitor requires evaluation of the start-up performance. SYNC Input Figure 3 shows the synchronous forward converter application. The primary controller LT3781 runs at a fixed frequency and controls MOSFETs Q1 and Q2. The secondary controller LTC1698 controls MOSFETs Q3 and Q4. An inexpensive, small-size pulse transformer T2 synchronizes the primary and the secondary controllers. Figure 4 shows the pulse transformer timing waveforms. When the LT3781 synchronization output SG goes low, MOSFET L1 VOUT TG Q1 Q4 CG VIN PRIMARY CONTROLLER LT3781 D1 W UU • D2 • T1 SECONDARY CONTROLLER LTC1698 Q3 COUT SG BG Q2 FG SYNC • CSG T2 • CSYNC RSYNC 1698 F03 PRIMARY ISOLATION BARRIER SECONDARY Figure 3. Synchronization Using Pulse Transformer TG BG SG SYNC 1698 F04 Figure 4. Primary Side and Secondary Side Synchronization Waveforms 1698f LTC1698 APPLICATIO S I FOR ATIO drivers TG and BG go high. The pulse transformer T2 generates a negative slew at the SYNC pin and forces the secondary MOSFET driver FG to go high and CG to go low. When TG and BG go low, SG goes high and the secondary controller forces CG high and FG low. For a given pulse transformer, a bigger capacitor CSG generates a higher and wider SYNC pulse. The peak of this pulse should be much higher than the SYNC threshold. Amplitudes greater than ± 5V help to speed up the SYNC comparator and reduce the SYNC to FG and CG drivers propagation delay. The minimum pulse width is 75ns. Overshoot during the pulse transformer reset interval must be minimized and kept below the minimum comparator thresholds of ±1V. The amount of overshoot can be reduced by having a smaller reset resistor RSYNC. For nonisolated applications, the SYNC input can be driven directly by a square pulse. To reduce the propagation delay, make the positive and negative magnitude of the square wave much greater than the ± 2.2V maximum threshold. In addition to the simple driver synchronization, the secondary controller requires a driver disable signal. Loss of synchronization while CG is high will cause Q4 to discharge the output capacitor. This produces a negative output voltage transient and possible damage to the load circuitry connected to VOUT. To overcome this problem, the LTC1698 comes with a unique adaptive time-out circuit. It works well within the 50kHz to 400kHz frequency range. At every positive SYNC pulse, the internal timer resets. If the SYNC signal is missing, the internal timer loses its reset command, and eventually exceeds the internal time-out limit. This forces both the FG and CG drivers to go low immediately. The time-out duration varies linearly with the LT3781 primary controller clocking frequency. Upon power up, the time-out circuitry takes a few clock cycles to adapt to the input clock frequency. During this time interval, the drivers pulse width might be prematurely terminated, and the inductor current flows through the MOSFETs body diode. Once the LTC1698 timer locks to the clocking frequency, the LTC1698 drivers follow the SYNC signal without fail. Figure 5 shows the SYNC time-out wave- U SG SYNC FG CG RESET (INTERNAL) DISDRI (INTERNAL) 1698 F05 W UU Figure 5. SYNC Time-Out Waveforms forms. The time-out circuit guarantees that if the SYNC pulse is missing for more than one period, both the drivers will be shut down preventing the output voltage from going below ground. The wide synchronization frequency range adds flexibility to the forward converter and allows this converter chip set to meet different application requirements. Under normal operating conditions, the time-out circuitry adapts to the switching frequency within a few cycles. Once synchronized, internal circuitry ensures the maximum time that the Catch FET (Q4) could be left turned on is typically just over one switching period. This is particularly important with high output voltages that can generate significant negative output inductor currents if the Catch FET Q4 is left on. Poor feedback loop performance including output voltage overshoot can cause the primary controller to interrupt the synchronization pulse train. While this generally is not a problem, it is possible that low frequency interruptions could lead to a time-out period longer than a switching period, limited only by the internal timer clamp (50µs typical). Output Voltage Programming The switching regulator output voltage is programmed through a resistor feedback network (R1 and R2 in Figure 1) connected to VFB. If the output is at its nominal value, the divider output is regulated to the error amplifier threshold of 1.233V. The output voltage is thus set according to the relation: VOUT = 1.233 • (1 + R2/R1) 1698f 11 LTC1698 APPLICATIO S I FOR ATIO MARGIN Adjustment The MARGIN input is used for adjusting the programmed output voltage linearly by varying the current flowing into and out of the pin. Forcing 100µA into the pin moves the output voltage 5% higher. Forcing 100µA out of the pin moves the output voltage 5% lower. With the MARGIN pin floating, the VFB pin is regulated to the bandgap voltage of 1.233V. The MARGIN pin is a high impedance input. It is important to keep this pin away from any noise source like the inductor switching node. Any stray signal coupled to the MARGIN pin can affect the switching regulator output voltage. This pin is internally connected to a 16.5k resistor that feeds the I-V converter. The I-V converter output linearly controls the error amplifier offset voltage. The input of the I-V converter is biased at 1.65V. This allows the ± 100µA current to be obtained by connecting the MARGIN pin to the VAUX 3.3V supply (+ 5%) or GND (– 5%). For output voltage adjustment smaller than ± 5%, an external resistor REXT as shown in Figure 6 is added in series with the internal resistor to lower the current flowing into or out of the MARGIN pin. The value of REXT is calculated as follow:   5% REXT =  – 1 • 16.5k  REQUIRED %  REXT (OPTIONAL) 7 REDUCE VFB INCREASE VFB RMARGIN MARGIN I-V CONVERTER BANDGAP VREF ± 5% VREF VDD ERR VAUX 3.3V 0.1µF 14 VAUX AUX GEN + – VFB VCOMP 8 6 Figure 6. Output Voltage Adjustment Overvoltage Function The OVPIN is used for overvoltage protection and is designed to protect against an open VFB loop. Opening the 12 U VFB loop causes the error amplifier to drive the OPTODRV pin low, forcing the primary controller to increase the duty cycle. This causes the output voltage to increase to a dangerously high level. To eliminate this fault condition, the OVP comparator monitors the output voltage with a resistive divider at OVPIN. A voltage at OVPIN higher than the VREF potential forces the OPTODRV pin high and reduces the duty cycle, thus preventing the output voltage from increasing further. The OVPIN senses the output voltage through a resistor divider network (R4 and R5 in Figure 1). The divider is ratioed such that the voltage at OVPIN equals 1.233V when the output voltage rises to the overvoltage level. The overvoltage level is set following the relation: VOVERVOLTAGE = 1.233 • (1 + R5/R4) The OVP comparator is designed to respond quickly to an overvoltage condition. A small capacitor from OVPIN to ground keeps any noise spikes from coupling to the OVP pin. This simple RC filter prevents a momentary overshoot from triggering the OVP comparator. The OVP comparator threshold is independent of the potential at the MARGIN pin. If the OVP function is not used, connect OVPIN to ground. Power Good The PWRGD pin is an open-drain output for power good indication. PWRGD floats if VFB is above 94% of the nominal value for more than 2ms. An external pull-up resistor is required for PWRGD to swing high. PWRGD pulls low if VFB drops below 94% of the nominal value for more than 1ms. The PWRGD threshold is referenced to the 1.233V bandgap voltage, which remains unchanged if the MARGIN pin is exercised. Opto Feedback and Frequency Compensation For a forward converter to obtain good load and line regulation, the output voltage must be sensed and compared to an accurate reference potential. Any error voltage must be amplified and fed back to the supply’s control circuitry where the sensed error can be corrected. In an isolated supply, the control circuitry is frequently located on the primary. The output error signal in this type of 1698f 1698 F06 W UU LTC1698 APPLICATIO S I FOR ATIO supply must cross the isolation boundary. Coupling this signal requires an element that will withstand the isolation potentials and still transfer the loop error signal. Optocouplers are widely used for this function due to their ability to couple DC signals. To properly apply them, a number of factors must be considered. The gain, or current transfer ratio (CTR) through an optocoupler is loosely specified and is a strong function of the input current through the diode. It changes considerably as a function of time (aging) and temperature. The amount of aging accelerates with higher operating current. This variation directly affects the overall loop gain of the system. To be an effective optical detector, the output transistor of the optocoupler must have a large base area to collect the light energy. This gives it a large collector to base capacitance which can introduce a pole into the feedback loop. This pole varies considerably with the current and interacts with the overall loop frequency compensation network. The common collector optocoupler configuration removes the miller effect due to the parasitic capacitance and  R • CTR  (1 + s • C C • RC ) (1 + s • RK • C K ) 1 1 VC =– •5• • F • •  (s • R2 • C C )• (1 + s • RC • C FB ) VOUT R • R   RD + RK  (1 + s • rπ • C DE ) (1 + s • RF • C F ) 1+ s • C K • K D   RK + RD   where: RD = Optocoupler diode equivalent small − signal resis tan ce CTR = Optocoupler current transfer ratio C DE = Optocoupler nonlinear capacitor across base to emitter = Optocoupler small − signal resis tan ce across the base emitter rπ LTC1698 OPTODRV LT3781 VCC RK CK 100k MOC207 VCOMP CC RE RF CF CFB 1698 F07 + – VC VREF VFB Figure 7. Error Signal Feedback 1698f U increases the frequency response. Figure 7 shows the optocoupler feedback circuitry using the common collector approach. Note that the terms RD, CTR, CDE and rπ vary from part to part. They also change with bias current. The dominant pole of the opto feedback is due to RF and CF. The feedforward capacitor CK at the optocoupler creates a low frequency zero. This zero should be chosen to provide a phase boost at the loop crossover frequency. The parallel combination of RK and RD form a high frequency pole with CK. For most optocouplers, RD is 50Ω at a DC bias of 1mA, and 25Ω at a DC bias of 2mA. The CTR term is the small signal AC current transfer ratio. For the QT Optoelectronics MOC207 optocoupler used here, the AC CTR is around 1, even though the DC CTR is much lower when biased at 1mA or 2mA. The first denominator term in the VC/VOUT equation has been simplified and assumes that CFB