LTC1772B

LTC1772B Constant Frequency Current Mode Step-Down DC/DC Controller in SOT-23 FEATURES n n n n n n n n n n DESCRIPTION The LTC®1772B is a constant frequency current mode step-down DC/DC controller providing excellent AC and DC load and line regulation. The device incorporates an accurate undervoltage lockout feature that shuts down the LTC1772B when the input voltage falls below 2.0V. The LTC1772B provides a ± 2.5% output voltage accuracy and consumes only 270μA of quiescent current. In shutdown, the device draws a mere 8μA. To further maximize the life of a battery source, the external P-channel MOSFET is turned on continuously in dropout (100% duty cycle). High constant operating frequency of 550kHz allows the use of a small external inductor. The LTC1772B is available in a small footprint 6-lead TSOT-23. For a Burst Mode operation enabled version of the LTC1772B, please refer to the LTC1772 data sheet. L, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Burst Mode® Operation Disabled for Lower Output Ripple at Light Loads High Efficiency: Up to 94% High Output Currents Easily Achieved Wide VIN Range: 2.5V to 9.8V Constant Frequency 550kHz Operation Low Dropout: 100% Duty Cycle Output Voltage Down to 0.8V Current Mode Operation for Excellent Line and Load Transient Response Shutdown Mode Draws Only 8µA Supply Current Tiny 6-Lead TSOT-23 Package APPLICATIONS n n n n n One or Two Lithium-Ion-Powered Applications Cellular Telephones Wireless Devices Portable Computers Distributed 3.3V, 2.5V or 1.8V Power Systems TYPICAL APPLICATION R1 0.03Ω 1 10k 220pF ITH/RUN PGATE LTC1772B 2 3 GND VFB VIN SENSE– 5 4 D1 6 L1 M1 4.7μH C1 10μF 10V VIN 2.5V TO 9.8V 100 95 90 EFFICIENCY (%) VOUT 2.5V 2A 174k 85 80 75 70 C1: TAIYO YUDEN LMK325BJ106K-T C2A: SANYO 6TPA47M C2B: AVX 0805ZC105KAT1A D1: MOTOROLA MBRM120T3 L1: MURATA LQN6C-4R7 M1: FAIRCHILD FDC638P R1: IRC LRC-LR1206-01-R030F 80.6k 65 60 1772 F01a Efficiency vs Load Current* LTC1772 Burst Mode OPERATION + C2A 47μF 6V C2B 1μF 10V LTC1772B NON-Burst Mode OPERATION VIN = 3.6V VOUT = 2.5V 10 100 1000 10000 LOAD CURRENT (mA) *OUTPUT RIPPLE WAVEFORMS FOR THE CIRCUIT OF FIGURE 1 APPEAR IN FIGURE 2. 1772 F01b Figure 1. High Efficiency, High Output Current 2.5V/2A Regulator 1772bfa 1 LTC1772B ABSOLUTE MAXIMUM RATINGS (Note 1) PIN CONFIGURATION TOP VIEW ITH/RUN 1 GND 2 VFB 3 6 PGATE 5 VIN 4 SENSE– Input Supply Voltage (VIN) .........................– 0.3V to 10V SENSE–, PGATE Voltages ...............–0.3V to (VIN + 0.3V) VFB, ITH/RUN Voltages............................... –0.3V to 2.4V PGATE Peak Output Current (< 10µs) ........................ 1A Storage Ambient Temperature Range.....– 65°C to 150°C Operating Temperature Range (Note 2).... –40°C to 85°C Junction Temperature (Note 3) ............................ 150°C Lead Temperature (Soldering, 10 sec) .................. 300°C S6 PACKAGE 6-LEAD PLASTIC TSOT-23 TJMAX = 150°C, θJA = 230°C/W ORDER INFORMATION LEAD FREE FINISH LTC1772BES6#PBF LTC1772BIS6#PBF LEAD BASED FINISH LTC1772BES6 LTC1772BIS6 TAPE AND REEL LTC1772BES6#TRPBF LTC1772BIS6#TRPBF TAPE AND REEL LTC1772BES6#TR LTC1772BIS6#TR PART MARKING* LTVU LTVU PART MARKING* LTVU LTVU PACKAGE DESCRIPTION 6-Lead Plastic TSOT Package 6-Lead Plastic TSOT Package PACKAGE DESCRIPTION 6-Lead Plastic TSOT Package 6-Lead Plastic TSOT Package TEMPERATURE RANGE –40°C to 85°C –40°C to 85°C TEMPERATURE RANGE –40°C to 85°C –40°C to 85°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS PARAMETER Input DC Supply Current Normal Operation Shutdown UVLO Undervoltage Lockout Threshold Shutdown Threshold (at ITH/RUN) Start-Up Current Source Regulated Feedback Voltage Output Voltage Line Regulation Output Voltage Load Regulation VITH/RUN = 0V 0°C to 70°C (Note 5) –40°C to 85°C (Note 5) 2.4V ≤ VIN ≤ 9.8V (Note 5) CONDITIONS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2) MIN TYP 270 8 6 l l l l MAX 420 22 10 2.35 2.40 0.55 0.85 0.820 0.830 UNITS μA μA μA V V V μA V V mV/V mV/μA mV/μA Typicals at VIN = 4.2V (Note 4) 2.4V ≤ VIN ≤ 9.8V, PGATE Logic High 2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 0V VIN < UVLO Threshold VIN Falling VIN Rising 1.55 1.85 0.15 0.25 0.780 0.770 2.00 2.10 0.35 0.5 0.800 0.800 0.05 2.5 2.5 ITH/RUN Sinking 5μA (Note 5) ITH/RUN Sourcing 5μA (Note 5) 1772bfa 2 LTC1772B ELECTRICAL CHARACTERISTICS PARAMETER VFB Input Current Overvoltage Protect Threshold Overvoltage Protect Hysteresis Oscillator Frequency Gate Drive Rise Time Gate Drive Fall Time Peak Current Sense Voltage VFB = 0.8V VFB = 0V CLOAD = 3000pF CLOAD = 3000pF (Note 6) 500 CONDITIONS (Note 5) Measured at VFB 0.820 The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2) MIN TYP 10 0.860 20 550 120 40 40 105 650 MAX 50 0.895 UNITS nA V mV kHz kHz ns ns mV Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC1772BE is guaranteed to meet specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LTC1772BI is guaranteed to meet specifications over the full –40°C to 85°C operating temperature range. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD • θJA°C/W) Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: The LTC1772B is tested in a feedback loop that servos VFB to the output of the error amplifier. Note 6: Peak current sense voltage is reduced dependent on duty cycle to a percentage of value as given in Figure 2. 1772bfa 3 LTC1772B TYPICAL PERFORMANCE CHARACTERISTICS Reference Voltage vs Temperature 825 820 815 VFB VOLTAGE (mV) Normalized Oscillator Frequency vs Temperature 10 8 NORMALIZED FREQUENCY (%) Undervoltage Lockout Trip Voltage vs Temperature 2.24 2.20 2.16 TRIP VOLTAGE (V) VIN = 4.2V VIN = 4.2V VIN FALLING 6 4 2 0 –2 –4 –6 –8 810 805 800 795 790 785 780 775 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1772 G01 2.12 2.08 2.04 2.00 1.96 1.92 1.88 –10 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1772 G02 1.84 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1772 G03 Maximum (VIN – SENSE –) Voltage vs Duty Cycle 120 110 TRIP VOLTAGE (mV) Shutdown Threshold vs Temperature 600 560 520 ITH/RUN VOLTAGE (mV) VIN = 4.2V TA = 25°C VIN = 4.2V 100 90 80 70 60 50 40 20 30 40 50 60 70 80 DUTY CYCLE (%) 90 100 480 440 400 360 320 280 240 200 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1772 G05 1772 G04 1772bfa 4 LTC1772B PIN FUNCTIONS ITH/RUN (Pin 1): This pin performs two functions. It serves as the error amplifier compensation point as well as the run control input. Nominal voltage range for this pin is 0.85V to 1.9V. Forcing this pin below 0.35V causes the device to be shut down. In shutdown all functions are disabled and the PGATE pin is held high. GND (Pin 2): Ground Pin. VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output. SENSE – (Pin 4): The Negative Input to the Current Comparator. VIN (Pin 5): Supply Pin. Must be closely decoupled to GND Pin 2. PGATE (Pin 6): Gate Drive for the External P-channel MOSFET. This pin swings from 0V to VIN. FUNCTIONAL DIAGRAM VIN 5 SENSE– 4 + 15mV ICMP – RS1 OSC SLOPE COMP R Q S SWITCHING LOGIC AND BLANKING CIRCUIT VIN PGATE 6 SHORT-CIRCUIT DETECT 0.5μA + VIN – VOLTAGE REFERENCE GND 2 UNDERVOLTAGE LOCKOUT VREF 0.8V + FREQ FOLDBACK – OVP 0.3V + – VREF + 60mV EAMP + – VREF 0.8V VFB 3 VIN 1 ITH/RUN 0.35V + SHDN CMP SHDN UV – 1.2V 1772FD 1772bfa 5 LTC1772B OPERATION Main Control Loop The LTC1772B is a constant frequency current mode switching regulator. During normal operation, the external P-channel power MOSFET is turned on each cycle when the oscillator sets the RS latch (RS1) and turned off when the current comparator (ICMP) resets the latch. The peak inductor current at which ICMP resets the RS latch is controlled by the voltage on the ITH/RUN pin, which is the output of the error amplifier EAMP An external resistive . divider connected between VOUT and ground allows the EAMP to receive an output feedback voltage VFB. When the load current increases, it causes a slight decrease in VFB relative to the 0.8V reference, which in turn causes the ITH/RUN voltage to increase until the average inductor current matches the new load current. The main control loop is shut down by pulling the ITH/RUN pin low. Releasing ITH/RUN allows an internal 0.5μA current source to charge up the external compensation network. When the ITH/RUN pin reaches 0.35V, the main control loop is enabled with the ITH/RUN voltage then pulled up to its zero current level of approximately 0.85V. As the external compensation network continues to charge up, the corresponding output current trip level follows, allowing normal operation. Comparator OVP guards against transient overshoots > 7.5% by turning off the external P-channel power MOSFET and keeping it off until the fault is removed. VOUT Ripple for Figure 1 Circuit Using LTC1772 Burst Mode Operation. (Refer to Functional Diagram) Low Load Current Operation Under very light load current conditions, the ITH/RUN pin voltage will be very close to the zero current level of 0.85V. As the load current decreases further, an internal offset at the current comparator input will assure that the current comparator remains tripped (even at zero load current) and the regulator will start to skip cycles, as it must, in order to maintain regulation. This behavior allows the regulator to maintain constant frequency down to very light loads, resulting in less low frequency noise generation over a wide load current range. Figure 2 illustrates this result for the circuit of Figure 1 using both an LTC1772 in Burst Mode operation and an LTC1772B (non-Burst Mode operation). At an output current of 100mA, the Burst Mode operation part exhibits an output ripple of approximately 60mVP-P, whereas the non-Burst Mode operation part has an output ripple of only 20mVP-P. At lower output current levels, the improvement is even greater. This comes at a tradeoff of lower efficiency for the non-Burst Mode operation part (see Figure 1). Also notice the constant frequency operation of the LTC1772B, even at 5% of maximum output current. Dropout Operation When the input supply voltage decreases towards the output voltage, the rate of change of inductor current during the ON cycle decreases. This reduction means VOUT Ripple for Figure 1 Circuit Using LTC1772B Non-Burst Mode Operation. 20mVAC/DIV 20mVAC/DIV VIN = 3.6V VOUT = 2.5V IOUT = 100mA 5μs/DIV 1772 F02a VIN = 3.6V VOUT = 2.5V IOUT = 100mA 5μs/DIV 1772 F02b Figure 2. Output Ripple Waveforms for the Circuit of Figure 1. 1772bfa 6 LTC1772B OPERATION (Refer to Functional Diagram) that the external P-channel MOSFET will remain on for more than one oscillator cycle since the inductor current has not ramped up to the threshold set by EAMP Further . reduction in input supply voltage will eventually cause the P-channel MOSFET to be turned on 100%, i.e., DC. The output voltage will then be determined by the input voltage minus the voltage drop across the MOSFET, the sense resistor and the inductor. Undervoltage Lockout To prevent operation of the P-channel MOSFET below safe input voltage levels, an undervoltage lockout is incorporated into the LTC1772B. When the input supply voltage drops below approximately 2.0V, the P-channel MOSFET and all circuitry is turned off except the undervoltage block, which draws only several microamperes. Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator will be reduced to about 120kHz. This lower frequency allows the inductor current to safely discharge, thereby preventing current runaway. The oscillator’s fre110 100 90 SF = IOUT/IOUT(MAX) (%) 80 70 60 50 40 30 20 10 0 quency will gradually increase to its designed rate when the feedback voltage again approaches 0.8V. Overvoltage Protection As a further protection, the overvoltage comparator in the LTC1772B will turn the external MOSFET off when the feedback voltage has risen 7.5% above the reference voltage of 0.8V. This comparator has a typical hysteresis of 20mV. Slope Compensation and Inductor’s Peak Current The inductor’s peak current is determined by: IPK = VITH – 0.85 10 RSENSE ( ) when the LTC1772B is operating below 40% duty cycle. However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor current. The amount of reduction is given by the curves in Figure 3. IRIPPLE = 0.4IPK AT 5% DUTY CYCLE IRIPPLE = 0.2IPK AT 5% DUTY CYCLE VIN = 4.2V 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 1772 F03 Figure 3. Maximum Output Current vs Duty Cycle 1772bfa 7 LTC1772B APPLICATIONS INFORMATION The basic LTC1772B application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L1 and RSENSE (= R1). Next, the power MOSFET, M1 and the output diode D1 is selected followed by CIN (= C1)and COUT(= C2). RSENSE Selection for Output Current RSENSE is chosen based on the required output current. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator determines the inductor’s peak current. The output current the LTC1772B can provide is given by: 0.105 IRIPPLE IOUT = − RSENSE 2 where IRIPPLE is the inductor peak-to-peak ripple current (see Inductor Value Calculation section). A reasonable starting point for setting ripple current is IRIPPLE = (0.4)(IOUT). Rearranging the above equation, it becomes: 0.0875 RSENSE = for Duty Cycle < 40% IOUT However, for operation that is above 40% duty cycle, slope compensation effect has to be taken into consideration to select the appropriate value to provide the required amount of current. Using Figure 3, the value of RSENSE is: RSENSE = (0.0875)SF IOUT 100 The inductance value also has a direct effect on ripple current. The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VIN or VOUT. The inductor’s peak-to-peak ripple current is given by: IRIPPLE = VIN VOUT VOUT + VD f (L ) VIN + VD where f is the operating frequency. Accepting larger values of IRIPPLE allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is IRIPPLE = 0.4(IOUT(MAX)). Remember, the maximum IRIPPLE occurs at the maximum input voltage. The ripple current is normally set such that the inductor current is continuous down to approximately 1/4 of maximum load current. This results in: IRIPPLE ≤ 0.03 RSENSE This implies a minimum inductance of: LMIN = VIN VOUT VOUT + VD VIN + VD 0.03 f RSENSE (Use VIN(MAX) = VIN) A smaller value than L MIN could be used in the circuit; however, the inductor current transitioning from continuous to discontinuous will occur at a higher load current. Power MOSFET Selection An external P-channel power MOSFET must be selected for use with the LTC1772B. The main selection criteria for the power MOSFET are the threshold voltage VGS(TH) and the “on” resistance RDS(ON), reverse transfer capacitance CRSS and total gate charge. () Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use of a smaller inductor for the same amount of inductor ripple current. However, this is at the expense of efficiency due to an increase in MOSFET gate charge losses. 1772bfa 8 LTC1772B APPLICATIONS INFORMATION Since the LTC1772B is designed for operation down to low input voltages, a logic level threshold MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required for applications that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the LTC1772B is less than the absolute maximum VGS rating, typically 8V. The required minimum RDS(ON) of the MOSFET is governed by its allowable power dissipation. For applications that may operate the LTC1772B in dropout, i.e., 100% duty cycle, at its worst case the required RDS(ON) is given by: R DS(ON) = PP DC =100% Under normal load conditions, the average current conducted by the diode is: ID = VIN VOUT IOUT VIN + VD The allowable forward voltage drop in the diode is calculated from the maximum short-circuit current as: VF ≈ PD ISC(MAX) (IOUT(MAX)) (1+ p) 2 where PD is the allowable power dissipation and will be determined by efficiency and/or thermal requirements. A fast switching diode must also be used to optimize efficiency. Schottky diodes are a good choice for low forward drop and fast switching times. Remember to keep lead length short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. CIN and COUT Selection In continuous mode, the source current of the P-channel MOSFET is a square wave of duty cycle (VOUT + VD)/ (VIN + VD). To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: C IN Required IRMS ≈ IMAX where PP is the allowable power dissipation and δp is the temperature dependency of RDS(ON). (1 + δp) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but δp = 0.005/°C can be used as an approximation for low voltage MOSFETs. In applications where the maximum duty cycle is less than 100% and the LTC1772B is in continuous mode, the RDS(ON) is governed by: R DS(ON) PP (DC) IOUT2 (1+ p) where DC is the maximum operating duty cycle of the LTC1772B. Output Diode Selection The catch diode carries load current during the off-time. The average diode current is therefore dependent on the P-channel switch duty cycle. At high input voltages the diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short-circuited. Under this condition the diode must safely handle IPEAK at close to 100% duty cycle. Therefore, it is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. [V (V OUT IN − VOUT )] 1/ 2 VIN This formula has a maximum value at VIN = 2VOUT, where IRMS = IOUT /2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet the 1772bfa 9 LTC1772B APPLICATIONS INFORMATION size or height requirements in the design. Due to the high operating frequency of the LTC1772B, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (ΔVOUT) is approximated by: VOUT IRIPPLE 1 ESR + 4fC OUT Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (η1 + η2 + η3 + ...) where η1, η2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1772B circuits: 1) LTC1772B DC bias 105 100 95 90 85 80 75 2.0 VREF where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage since ΔIL increases with input voltage. Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR (size) product of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. Low Supply Operation Although the LTC1772B can function down to approximately 2.0V, the maximum allowable output current is reduced when VIN decreases below 3V. Figure 4 shows the amount of change as the supply is reduced down to 2V. Also shown in Figure 4 is the effect of VIN on VREF as VIN goes below 2.3V. Setting Output Voltage The regulated output voltage is determined by: VOUT = 0.8 1+ R2 R1 NORMALIZED VOLTAGE (%) VITH 2.2 2.4 2.6 2.8 INPUT VOLTAGE (V) 3.0 1772 F03 Figure 4. Line Regulation of VREF and VITH VOUT LTC1772B 3 VFB R2 R1 1772 F04 Figure 5. Setting Output Voltage For most applications, an 80k resistor is suggested for R1. To prevent stray pickup, locate resistors R1 and R2 close to LTC1772B. 1772bfa 10 LTC1772B APPLICATIONS INFORMATION current, 2) MOSFET gate charge current, 3) I2R losses and 4) voltage drop of the output diode. 1. The VIN current is the DC supply current, given in the electrical characteristics, that excludes MOSFET driver and control currents. VIN current results in a small loss which increases with VIN. 2. MOSFET gate charge current results from switching the gate capacitance of the power MOSFET. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN which is typically much larger than the DC supply current. In continuous mode, IGATECHG = f(Qp). 3. losses are predicted from the DC resistances of the MOSFET, inductor and current shunt. In continuous mode the average output current flows through L but is “chopped” between the P-channel MOSFET (in series with RSENSE) and the output diode. The MOSFET RDS(ON) plus RSENSE multiplied by duty cycle can be summed with the resistances of L and RSENSE to obtain I2R losses. I2R load current. For example, assuming a duty cycle of 50% with a Schottky diode forward voltage drop of 0.4V, the loss increases from 0.5% to 8% as the load current increases from 0.5A to 2A. 5. Transition losses apply to the external MOSFET and increase at higher operating frequencies and input voltages. Transition losses can be estimated from: Transition Loss = 2(VIN)2IO(MAX)CRSS(f) Other losses including CIN and COUT ESR dissipative losses, and inductor core losses, generally account for less than 2% total additional loss. Foldback Current Limiting As described in the Output Diode Selection, the worst-case dissipation occurs with a short-circuited output when the diode conducts the current limit value almost continuously. To prevent excessive heating in the diode, foldback current limiting can be added to reduce the current in proportion to the severity of the fault. Foldback current limiting is implemented by adding diodes DFB1 and DFB2 between the output and the ITH/RUN pin as shown in Figure 6. In a hard short (VOUT = 0V), the current will be reduced to approximately 50% of the maximum output current. 4. The output diode is a major source of power loss at high currents and gets worse at high input voltages. The diode loss is calculated by multiplying the forward voltage times the diode duty cycle multiplied by the LTC1772B R2 ITH/RUN VFB R1 DFB2 1772 F05 VOUT + DFB1 Figure 6. Foldback Current Limiting 1772bfa 11 LTC1772B APPLICATIONS INFORMATION PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1772B. These items are illustrated graphically in the layout diagram in Figure 7. Check the following in your layout: 1. Is the Schottky diode closely connected between ground (Pin 2) and drain of the external MOSFET? 2. Does the (+) plate of CIN connect to the sense resistor as closely as possible? This capacitor provides AC current to the MOSFET. 3. Is the input decoupling capacitor (0.1μF) connected closely between VIN (Pin 5) and ground (Pin 2)? 4. Connect the end of RSENSE as close to VIN (Pin 5) as possible. The VIN pin is the SENSE + of the current comparator. 5. Is the trace from SENSE – (Pin 4) to the Sense resistor kept short? Does the trace connect close to RSENSE? 6. Keep the switching node PGATE away from sensitive small signal nodes. 7. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1 and R2 must be connected between the (+) plate of COUT and signal ground. 1 ITH/RUN PGATE LTC1772B GND VIN SENSE– 6 RSENSE 0.1μF M1 + CIN L1 VIN RITH 2 5 VOUT + D1 COUT 3 CITH VFB 4 R1 BOLD LINES INDICATE HIGH CURRENT PATHS 1772 F06 R2 Figure 7. LTC1772B Layout Diagram (See PC Board Layout Checklist) 1772bfa 12 LTC1772B TYPICAL APPLICATIONS LTC1772B High Efficiency, Small Footprint 3.3V to 1.8V/0.5A Regulator VIN 3.3V R1 0.15Ω 1 R4 10k C3 220pF ITH/RUN PGATE LTC1772B 2 3 GND VFB 4 SENSE– VIN 5 D1 6 L1 M1 10μH C1 10μF 10V + C2 47μF 6V VOUT 1.8V 0.5A R2 100k R3 80.6k 1772 TA02 C1: TAIYO YUDEN CERAMIC L1: COILTRONICS UP1B-100 LMK325BJ106K-T M1: Si3443DV C2: SANYO POSCAP 6TPA47M R1: DALE 0.25W D1: MOTOROLA MBRM120T3 1772bfa 13 LTC1772B TYPICAL APPLICATIONS LTC1772B 5V/0.5A Flyback Regulator VIN 2.5V TO 9.8V R1 0.033Ω 1 R4 10k C3 220pF ITH/RUN PGATE LTC1772B 2 3 GND VFB VIN SENSE– 5 4 D1 T1 6 M1 C1 10μF 10V • 10μH 10μH + • C2 100μF 10V ×2 VOUT 5V 0.5A R2 52.3k C1: TAIYO YUDEN CERAMIC LMK325BJ106K-T C2: AVXTPSE107M010R0100 D1: IR10BQ015 M1: Si9803 R1: DALE 0.25W T1: COILTRONICS CTX10-4 R3 10k 1772 TA04 1772bfa 14 LTC1772B PACKAGE DESCRIPTION S6 Package 6-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1636) 2.90 BSC (NOTE 4) 0.62 MAX 0.95 REF 1.22 REF 3.85 MAX 2.62 REF 1.4 MIN 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE ID RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.95 BSC 0.30 – 0.45 6 PLCS (NOTE 3) 0.80 – 0.90 0.20 BSC 1.00 MAX DATUM ‘A’ 0.01 – 0.10 0.30 – 0.50 REF NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 0.09 – 0.20 (NOTE 3) 1.90 BSC S6 TSOT-23 0302 REV B 1772bfa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LTC1772B TYPICAL APPLICATION LTC1772B 3.3V to 5V/1A Boost Regulator R1 0.033Ω C1 47μF 16V ×2 U1 1 R4 10k C3 220pF ITH/RUN PGATE LTC1772B 2 3 GND VFB VIN SENSE– 5 4 R2 422k U1: FAIRCHILD NC7SZ04 ALSO SEE LTC1872 FOR THIS APPLICATION R3 80.6k 1772 TA03 VIN 3.3V L1 4.7μH D1 6 2 5 3 4 M1 + C2 100μF 10V ×2 VOUT 5V 1A C1: AVXTPSE476M016R0047 C2: AVXTPSE107M010R0100 D1: IR10BQ015 L1: MURATA LQN6C-4R7 M1: Si9804 R1: DALE 0.25W RELATED PARTS PART NUMBER LT1375/LT1376 LTC1622 LTC1624 LTC1625 LTC1627 LTC1649 LTC1702 LTC1735 LTC1771 LTC1772 LTC1773 LTC1872 DESCRIPTION 1.5A, 500kHz Step-Down Switching Regulators Low Input Voltage Current Mode Step-Down DC/DC Controller High Efficiency SO-8 N-Channel Switching Regulator Controller No RSENSETM Synchronous Step-Down Regulator Low Voltage, Monolithic Synchronous Step-Down Regulator 3.3V Input Synchronous Step-Down Controller 550kHz, 2 Phase, Dual Synchronous Controller Single, High Efficiency, Low Noise Synchronous Switching Controller Ultra-Low Supply Current Step-Down DC/DC Controller Constant Frequency Current Mode Step-Down DC/DC Controller in SOT-23 95% Efficient Synchronous Step-Down Controller SOT-23 Step-Up Controller COMMENTS High Frequency, Small Inductor, High Efficiency VIN 2V to 10V, IOUT Up to 4.5A, Synchronizable to 750kHz Optional Burst Mode Operation, 8-Lead MSOP N-Channel Drive, 3.5V ≤ VIN ≤ 36V 97% Efficiency, No Sense Resistor Low Supply Voltage Range: 2.65V to 8V, IOUT = 0.5A No Need for 5V Supply, Uses Standard Logic Gate MOSFETs; IOUT up to 15A Two Channels; Minimum CIN and COUT, IOUT up to 15A High Efficiency 5V to 3.3V Conversion at up to 15A 10μA Supply Current, 93% Efficiency, 1.23V ≤ VOUT ≤ 18V; 2.8V ≤ VIN ≤ 20V With Burst Mode Operation for Higher Efficiency at Light Load Current 2.65V ≤ VIN ≤ 8.5V; 0.8V ≤ VOUT ≤ VIN; Current Mode; 550kHz 2.5V ≤ VIN ≤ 9.8V; 550kHz; 90% Efficiency No RSENSE is a trademark of Linear Technology Corporation. 1772bfa 16 Linear Technology Corporation (408) 432-1900 ● FAX: (408) 434-0507 ● LT 0508 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 www.linear.com © LINEAR TECHNOLOGY CORPORATION 1999