LTC1872B

LTC1872B Constant Frequency Current Mode Step-Up DC/DC Controller in ThinSOT FEATURES s s s s s s s DESCRIPTIO s s Burst ModeTM Operation Disabled for Lower Output Ripple at Light Loads High Efficiency: Over 90% High Output Currents Easily Achieved Wide VIN Range: 2.5V to 9.8V VOUT Limited Only by External Components Constant Frequency 550kHz Operation Current Mode Operation for Excellent Line and Load Transient Response Shutdown Mode Draws Only 8µA Supply Current Low Profile (1mm) ThinSOTTM Package The LTC®1872B is a constant frequency current mode step-up DC/DC controller providing excellent AC and DC load and line regulation. The device incorporates an accurate undervoltage lockout feature that shuts down the LTC1872B when the input voltage falls below 2.0V. The LTC1872B provides a ± 2.5% output voltage accuracy and consumes only 270µA of quiescent current. In shutdown, the device draws a mere 8µA. High constant operating frequency of 550kHz allows the use of a small external inductor. The constant frequency operation is maintained down to very light loads, resulting in less low frequency noise generation over a wide load current range. The LTC1872B is available in a 6-lead low profile (1mm) ThinSOT package. For a Burst Mode operation enabled version of the LTC1872B, please refer to the LTC1872 data sheet. , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode and ThinSOT are trademarks of Linear Technology Corporation. APPLICATIO S s s s s s s Optical Communications Lithium-Ion-Powered Applications Cellular Telephones Wireless Devices Portable Computers Scanners TYPICAL APPLICATION R1 0.03Ω 147k 220pF 80.6k 1 ITH/RUN GND VFB VIN SENSE – NGATE 422k C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: MURATA GRM42-2X5R106K010AL D1: IR10BQ015 L1: MURATA LQN6C4R7M04 M1: Si2302DS R1: DALE 0.25W 5 L1 4.7µH VOUT 5V 1A C1 10µF 10V VIN 3.3V 100 95 90 EFFICIENCY (%) 85 80 75 70 65 1 10 100 LOAD CURRENT (mA) 1000 1872B TA01b LTC1872B 2 3 4 6 M1 D1 C2 4× 10µF 10V 1872B TA01 Figure 1. LTC1872B High Output Current 3.3V to 5V Boost Converter *Output ripple waveforms for the circuit of Figure 1 appear in Figure 2. U Typical Efficiency vs Load Current* VIN = 3.3V VOUT = 5V U U 1 LTC1872B ABSOLUTE MAXIMUM RATINGS (Note 1) PACKAGE/ORDER INFORMATION TOP VIEW ITH/RUN 1 GND 2 VFB 3 6 NGATE 5 VIN 4 SENSE – Input Supply Voltage (VIN).........................– 0.3V to 10V SENSE –, NGATE Voltages ............ – 0.3V to (VIN + 0.3V) VFB, ITH/RUN Voltages ..............................– 0.3V to 2.4V NGATE Peak Output Current (< 10µs) ....................... 1A Storage Ambient Temperature Range ... – 65°C to 150°C Operating Temperature Range (Note 2) .. – 40°C to 85°C Junction Temperature (Note 3) ............................. 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER LTC1872BES6 S6 PART MARKING LTXY S6 PACKAGE 6-LEAD PLASTIC SOT-23 TJMAX = 150°C, θJA = 230°C/ W Consult LTC marketing for parts specified with wider operating temperature ranges. The q denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2) PARAMETER Input DC Supply Current Normal Operation Sleep Mode Shutdown UVLO Undervoltage Lockout Threshold Shutdown Threshold (at ITH/RUN) Start-Up Current Source Regulated Feedback Voltage VFB Input Current Oscillator Frequency Gate Drive Rise Time Gate Drive Fall Time Peak Current Sense Voltage VITH/RUN = 0V 0°C to 70°C(Note 5) – 40°C to 85°C(Note 5) (Note 5) VFB = 0.8V CLOAD = 3000pF CLOAD = 3000pF (Note 6) 114 500 q q ELECTRICAL CHARACTERISTICS CONDITIONS Typicals at VIN = 4.2V (Note 4) 2.4V ≤ VIN ≤ 9.8V 2.4V ≤ VIN ≤ 9.8V 2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 0V VIN < UVLO Threshold VIN Falling VIN Rising q q MIN TYP 270 230 8 6 MAX 420 370 22 10 2.35 2.40 0.55 0.85 0.820 0.830 50 650 UNITS µA µA µA µA V V V µA V V nA kHz ns ns mV 1.55 1.85 0.15 0.25 0.780 0.770 2.00 2.10 0.35 0.5 0.800 0.800 10 550 40 40 120 Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC1872BE is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the – 40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD • θJA°C/W) Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: The LTC1872B is tested in a feedback loop that servos VFB to the output of the error amplifier. Note 6: Guaranteed by design at duty cycle = 30%. Peak current sense voltage is VREF/6.67 at duty cycle 7.5% by turning off the external N-channel power MOSFET and keeping it off until the fault is removed. Low Load Current Operation Under very light load current conditions, the ITH/RUN pin voltage will be very close to the zero current level of 0.85V. As the load current decreases further, an internal offset at the current comparator input will assure that the current comparator remains tripped (even at zero load current) and the regulator will start to skip cycles, as it must, in order to maintain regulation. This behavior allows the regulator to maintain constant frequency down to very light loads, resulting in less low frequency noise generation over a wide load current range. LTC1872B OPERATIO Figure 2 illustrates this result for the circuit of Figure 1 using both an LTC1872 in Burst Mode operation and an LTC1872B (non-Burst Mode operation). At an output current of 50mA, the Burst Mode operation part exhibits an output ripple of approximately 80mVP-P, whereas the non-Burst Mode operation part has an output ripple of ≈45mVP-P. At lower output current levels, the improvement is even greater. This comes at a trade off of slightly lower efficiency for the non-Burst Mode operation part. Also notice the constant frequency operation of the LTC1872B, even at 5% of maximum output current. Undervoltage Lockout To prevent operation of the N-channel MOSFET below safe input voltage levels, an undervoltage lockout is incorporated into the LTC1872B. When the input supply voltage drops below approximately 2.0V, the N-channel MOSFET and all circuitry is turned off except the undervoltage block, which draws only several microamperes. Overvoltage Protection The overvoltage comparator in the LTC1872B will turn the external MOSFET off when the feedback voltage has risen 7.5% above the reference voltage of 0.8V. This comparator has a typical hysteresis of 20mV. SF = IOUT/IOUT(MAX) (%) 20mV AC/DIV VIN = 3.3V VOUT = 5V IOUT = 50mA (2a) VOUT Ripple for Figure 1 Circuit Using LTC1872 Burst Mode Operation U (Refer to Functional Diagram) Slope Compensation and Inductor’s Peak Current The inductor’s peak current is determined by: IPK = VITH − 0.7 10(RSENSE ) when the LTC1872B is operating below 40% duty cycle. However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor current. The amount of reduction is given by the curves in Figure 3. 110 100 90 80 70 60 50 40 30 20 10 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 1872B F03 IRIPPLE = 0.4IPK AT 5% DUTY CYCLE IRIPPLE = 0.2IPK AT 5% DUTY CYCLE VIN = 4.2V Figure 3. Maximum Output Current vs Duty Cycle 20mV AC/DIV 5µs/DIV 1872B F02a VIN = 3.3V VOUT = 5V IOUT = 50mA 5µs/DIV 1872B F02b (2b) VOUT Ripple for Figure 1 Circuit Using LTC1872B Non-Burst Mode Operation Figure 2. Output Ripple Waveforms for the Circuit of Figure 1 5 LTC1872B OPERATIO Short-Circuit Protection Since the power switch in a boost converter is not in series with the power path from input to load, turning off the switch provides no protection from a short-circuit at the APPLICATIONS INFORMATION The basic LTC1872B application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L1 and RSENSE (= R1). Next, the power MOSFET and the output diode D1 is selected followed by CIN(= C1) and COUT(= C2). RSENSE Selection for Output Current RSENSE is chosen based on the required output current. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator determines the inductor’s peak current. The output current the LTC1872B can provide is given by:  0.12  I VIN IOUT =  − RIPPLE   RSENSE 2  VOUT + VD where IRIPPLE is the inductor peak-to-peak ripple current (see Inductor Value Calculation section) and VD is the forward drop of the output diode at the full rated output current. A reasonable starting point for setting ripple current is: IRIPPLE = (O.4)(IOUT ) VOUT + VD VIN Rearranging the above equation, it becomes: RSENSE =   1 VIN   (10)( IOUT)  VOUT + VD  for Duty Cycle < 40% However, for operation that is above 40% duty cycle, slope compensation’s effect has to be taken into consideration to select the appropriate value to provide the required amount of current. Using the scaling factor (SF, in %) in Figure 3, the value of RSENSE is: 6 U W U U U (Refer to Functional Diagram) output. External means such as a fuse in series with the boost inductor must be employed to handle this fault condition. RSENSE =   VIN   (10)(IOUT )(100)  VOUT + VD  SF Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use of a smaller inductor for the same amount of inductor ripple current. However, this is at the expense of efficiency due to an increase in MOSFET gate charge losses. The inductance value also has a direct effect on ripple current. The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VOUT. The inductor’s peak-to-peak ripple current is given by: IRIPPLE = VIN  VOUT + VD − VIN    f (L)  VOUT + VD  where f is the operating frequency. Accepting larger values of IRIPPLE allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is: V + VD  IRIPPLE = 0.4 IOUT (MAX)  OUT    VIN ( ) In Burst Mode operation, the ripple current is normally set such that the inductor current is continuous during the burst periods. Therefore, the peak-to-peak ripple current must not exceed: IRIPPLE ≤ 0.03 RSENSE LTC1872B APPLICATIONS INFORMATION This implies a minimum inductance of:  VOUT + VD − VIN  VIN LMIN =    0.03   VOUT + VD  f   RSENSE  A smaller value than L MIN could be used in the circuit; however, the inductor current will not be continuous during burst periods. Inductor Selection When selecting the inductor, keep in mind that inductor saturation current has to be greater than the current limit set by the current sense resistor. Also, keep in mind that the DC resistance of the inductor will affect the efficiency. Off the shelf inductors are available from Murata, Coilcraft, Toko, Panasonic, Coiltronics and many other suppliers. Power MOSFET Selection The main selection criteria for the power MOSFET are the threshold voltage VGS(TH), the “on” resistance RDS(ON), reverse transfer capacitance CRSS and total gate charge. Since the LTC1872B is designed for operation down to low input voltages, a logic level threshold MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required for applications that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the LTC1872B is less than the absolute maximum VGS rating, typically 8V. The required minimum RDS(ON) of the MOSFET is governed by its allowable power dissipation given by: R DS(ON) ≅ Output Diode Selection Under normal load conditions, the average current conducted by the diode in a boost converter is equal to the output load current: (DC )IIN2(1+ δp) PP where PP is the allowable power dissipation and δp is the temperature dependency of RDS(ON). (1 + δp) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but δp = 0.005/°C can be used as an approximation for low voltage MOSFETs. DC is the maximum operating duty cycle of the LTC1872B. U W U U ID(avg) = IOUT It is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. Schottky diodes are recommended for low forward drop and fast switching times. Remember to keep lead length short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. CIN and COUT Selection To prevent large input voltage ripple, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current for a boost converter is approximately equal to: CIN Required IRMS ≈ (0.3)IRIPPLE where IRIPPLE is as defined in the Inductor Value Calculation section. Note that capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet the size or height requirements in the design. Due to the high operating frequency of the LTC1872B, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (∆VOUT) is approximated by: 7 LTC1872B APPLICATIONS INFORMATION ∆VOUT  V + VD IRIPPLE  ≈  IO • OUT + •  VIN 2    1  ESR2 +   2π fC   OUT       1 2 2 where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR (size) product of any aluminum electrolytic at a somewhat higher price. The output capacitor RMS current is approximately equal to: IPK • DC − DC 2 where IPK is the peak inductor current and DC is the switch duty cycle. When using electrolytic output capacitors, if the ripple and ESR requirements are met, there is likely to be far more capacitance than required. In surface mount applications, multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. An excellent choice of tantalum capacitors is the AVX TPS and KEMET T510 series of surface mount tantalum capacitors. Also, ceramic capacitors in X5R pr X7R dielectrics offer excellent performance. Low Supply Operation Although the LTC1872B can function down to approximately 2.0V, the maximum allowable output current is reduced when VIN decreases below 3V. Figure 4 shows the NORMALIZED VOLTAGE (%) 8 U W U U amount of change as the supply is reduced down to 2V. Also shown in Figure 4 is the effect of VIN on VREF as VIN goes below 2.3V. Setting Output Voltage The LTC1872B develops a 0.8V reference voltage between the feedback (Pin 3) terminal and ground (see Figure 5). By selecting resistor R1, a constant current is caused to flow through R1 and R2 to set the overall output voltage. The regulated output voltage is determined by:  R2  VOUT = 0.8V 1 +   R1  For most applications, an 80k resistor is suggested for R1. To prevent stray pickup, locate resistors R1 and R2 close to LTC1872B. 105 100 95 90 85 80 75 2.0 VREF VITH 2.2 2.4 2.6 2.8 INPUT VOLTAGE (V) 3.0 1872B F04 Figure 4. Line Regulation of VREF and VITH VOUT LTC1872B 3 VFB R2 R1 1872B F05 Figure 5. Setting Output Voltage LTC1872B APPLICATIONS INFORMATION Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (η1 + η2 + η3 + ...) where η1, η2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1872B circuits: 1) LTC1872B DC bias current, 2) MOSFET gate charge current, 3) I2R losses and 4) voltage drop of the output diode. 1. The VIN current is the DC supply current, given in the electrical characteristics, that excludes MOSFET driver and control currents. VIN current results in a small loss which increases with VIN. 2. MOSFET gate charge current results from switching the gate capacitance of the power MOSFET. Each time a MOSFET gate is switched from low to high to low again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is a current out of VIN which is typically much larger than the contoller’s DC supply current. In continuous mode, IGATECHG = f(Qp). 3. I2R losses are predicted from the DC resistances of the MOSFET, inductor and current sense resistor. The MOSFET RDS(ON) multiplied by duty cycle times the average output current squared can be summed with I2R losses in the inductor ESR in series with the current sense resistor. 4. The output diode is a major source of power loss at high currents. The diode loss is calculated by multiplying the forward voltage by the load current. 5. Transition losses apply to the external MOSFET and increase at higher operating frequencies and input voltages. Transition losses can be estimated from: Transition Loss = 2(VIN)2IIN(MAX)CRSS(f) Other losses, including CIN and COUT ESR dissipative losses, and inductor core losses, generally account for less than 2% total additional loss. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1872B. These items are illustrated graphically in the layout diagram in Figure 6. Check the following in your layout: 1. The Schottky diode should be closely connected between the output capacitor and the drain of the external MOSFET. 2. The (+) plate of CIN should connect to the sense resistor as closely as possible. This capacitor provides AC current to the inductor. 3. The input decoupling capacitor (0.1µF) should be connected closely between VIN (Pin 5) and ground (Pin 2). 4. Connect the end of RSENSE as close to VIN (Pin 5) as possible. The VIN pin is the SENSE + of the current comparator. 5. The trace from SENSE – (Pin 4) to the Sense resistor should be kept short. The trace should connect close to RSENSE. 6. Keep the switching node NGATE away from sensitive small signal nodes. 7. The VFB pin should connect directly to the feedback resistors. The resistive divider R1 and R2 must be connected between the (+) plate of COUT and signal ground. U W U U 9 LTC1872B APPLICATIONS INFORMATION 1 ITH/RUN NGATE LTC1872B GND VIN RITH 2 3 CITH VFB 4 SENSE – R2 R1 BOLD LINES INDICATE HIGH CURRENT PATHS Figure 6. LTC1872B Layout Diagram (See PC Board Layout Checklist) TYPICAL APPLICATIO S LTC1872B 3-Cell White LED Driver VIN = 3 AA CELLS ≈ 2.7V TO 4.8V R1 0.27Ω AA 1 10k 220pF AA ITH/RUN GND VFB VIN SENSE – LTC1872B 2 3 4 6 AA NGATE C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: AVX TPSD156M035R0300 D0: MOTOROLA MBR0540 D1-D7: CMD333UWC 10 U W U U U 6 RS 0.1µF L1 D1 M1 VIN 5 + CIN + COUT VOUT 1872B F06 C1 10µF 10V 5 L1 150µH VOUT ≈ 28.8V (WITH 8 LEDs) M1 D0 + C2 15µF 35V C3 0.1µF CERAMIC 15mA D1 D2 • • • D8 1 TO 8 WHITE LEDs L1: COILCRAFT DO1608C-154 M1: Si9804 R1: DALE 0.25W 53.6Ω 1872B TA04 LTC1872B TYPICAL APPLICATIO S LTC1872B 12V/500mA Boost Converter R1 0.033Ω 1 10k 220pF ITH/RUN GND VFB VIN SENSE – NGATE 1.1M 1872B TA02 C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: AVX TPSE476M016R0150 D1: IR10BQ015 PACKAGE DESCRIPTION S6 Package 6-Lead Plastic SOT-23 (Reference LTC DWG # 05-08-1634) (Reference LTC DWG # 05-08-1636) 2.80 – 3.10 (.110 – .118) (NOTE 3) SOT-23 (Original) A A1 A2 L .90 – 1.45 (.035 – .057) .00 – 0.15 (.00 – .006) .90 – 1.30 (.035 – .051) .35 – .55 (.014 – .021) SOT-23 (ThinSOT) 1.00 MAX (.039 MAX) .01 – .10 (.0004 – .004) .80 – .90 (.031 – .035) .30 – .50 REF (.012 – .019 REF) 2.60 – 3.00 (.102 – .118) 1.50 – 1.75 (.059 – .069) (NOTE 3) PIN ONE ID .20 (.008) DATUM ‘A’ A A2 L NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE 4. DIMENSIONS ARE INCLUSIVE OF PLATING 5. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 6. MOLD FLASH SHALL NOT EXCEED .254mm 7. PACKAGE EIAJ REFERENCE IS: SC-74A (EIAJ) FOR ORIGINAL JEDEL MO-193 FOR THIN Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U U C1 10µF 10V VIN 3V TO 9.8V 5 L1 10µH LTC1872B 2 3 4 6 M1 D1 + C2 47µF 16V VOUT 12V 78.7k L1: COILTRONICS UP2B-100 M1: Si9804DV R1: DALE 0.25W .95 (.037) REF .25 – .50 (.010 – .020) (6PLCS, NOTE 2) .09 – .20 (.004 – .008) (NOTE 2) 1.90 (.074) REF A1 S6 SOT-23 0401 11 LTC1872B TYPICAL APPLICATIO S LTC1872B – 2.5V to 3.3V/0.5A Boost Converter R1 0.034Ω 1 10k 220pF ITH/RUN GND VFB VIN SENSE – NGATE 5 L1 4.7µH C2 2× 100µF 10V VOUT 3.3V 0.5A LTC1872B 2 3 4 6 M1 D1 332k 0.1µF CERAMIC + C1 100µF 10V U1 VIN –2.5V C1, C2: AVX TPSE107M010R0100 D1: MOTOROLA MBR2045CT L1: COILTRONICS UP2B-4R7 M1: Si9804DV R1: DALE 0.25W U1: PANASONIC 2SB709A LTC1872B 2.7V to 9.8V Input to 3.3V/1.2A Output SEPIC Converter RCS 0.03Ω ITH/RUN GND VFB VIN SENSE – NGATE 5 L1A 4 6 L1B D1 MBRM120 CS 4.7µF 10V M1 VIN 2.7V TO 9.8V CIN 10µF 10V, X5R CC1 220pF RC1 10k 1 + LTC1872B 2 3 Rf2 80.6k Rf1 252k CO1 180µF 4V, SP CIN, CS; TOKO, MURATA OR TAIYO YUDEN CO1: PANASONIC EEFUE0G181R L1: BH ELECTRONICS 511-1012 M1: IRLMS2002 RCS: DALE OR IRC FOR VOUT = 5V CHANGE Rf1 TO 427kΩ AND CO1 TO 150µF, 6V PANASONIC SP TYPE CAPACITOR RELATED PARTS PART NUMBER LT1304 LT1610 LT1613 LT1619 LT1680 LTC1624 LT1615 LTC1700 LTC1772 LTC1872 DESCRIPTION Micropower DC/DC Converter with Low-Battery Detector 1.7MHz, Single Cell Micropower DC/DC Converter 1.4MHz, Single Cell DC/DC Converter in 5-Lead ThinSOT Low Voltage Current Mode PWM Controller High Power DC/DC Step-Up Controller High Efficiency SO-8 N-Channel Switching Regulator Controller Micropower Step-Up DC/DC Converter in ThinSOT No RSENSE Synchronous Current Mode DC/DC Step-Up Controller Constant Frequency Current Mode Step-Down DC/DC Controller COMMENTS 120µA Quiescent Current, 1.5V ≤ VIN ≤ 8V 30µA Quiescent Current, VIN Down to 1V Internally Compensated, VIN Down to 1V 8-Lead MSOP Package, 1.9V ≤ VIN ≤ 18V Operation Up to 60V, Fixed Frequency Current Mode 8-Pin N-Channel Drive, 3.5V ≤ VIN ≤ 36V 20µA Quiescent Current, VIN Down to 1V 95% Efficient, 0.9V ≤ VIN ≤ 5V, 550kHz Operation VIN 2.5V to 9.8V, IOUT up to 4A, ThinSOT Package Constant Frequency Current Mode Step-Up DC/DC Controller in ThinSOT With Burst Mode Operation for Higher Efficiency at Light Load Current 10-Lead MSOP Package, 0.5V ≤ VIN ≤ 5V 1872bf LT/TP 0601 2K • PRINTED IN USA LTC3401/LTC3402 1A/2A, 3MHz Micropower Synchronous Boost Converter 12 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 q FAX: (408) 434-0507 q www.linear-tech.com + U 80.6k 1872B TA03 180k VOUT 3.3V/1.2A 1872B TA05 © LINEAR TECHNOLOGY CORPORATION 2001