LTC3406BES5-1.5

LTC3406B 1.5MHz, 600mA Synchronous Step-Down Regulator in ThinSOT FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO High Efficiency: Up to 96% 600mA Output Current at VIN = 3V 2.5V to 5.5V Input Voltage Range 1.5MHz Constant Frequency Operation No Schottky Diode Required Low Dropout Operation: 100% Duty Cycle Low Quiescent Current: 300µA 0.6V Reference Allows Low Output Voltages Shutdown Mode Draws < 1µA Supply Current Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Low Profile (1mm) ThinSOTTM Package The LTC ®3406B is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. The device is available in an adjustable version and fixed output voltages of 1.5V and 1.8V. Supply current with no load is 300µA and drops to 40%. However, the LTC3406B uses a patent-pending scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles. 1200 1000 800 600 400 200 0 VOUT = 1.8V VOUT = 2.5V VOUT = 1.5V 2.5 3.0 3.5 4.0 4.5 SUPPLY VOLTAGE (V) 5.0 5.5 3406B F02 Figure 2. Maximum Output Current vs Input Voltage 3406bfa 7 LTC3406B APPLICATIO S I FOR ATIO The basic LTC3406B application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Inductor Selection For most applications, the value of the inductor will fall in the range of 1µH to 4.7µH. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is ∆IL = 240mA (40% of 600mA). ∆IL = ( )( ) ⎛V⎞ VOUT ⎜ 1 − OUT ⎟ VIN ⎠ ⎝ fL 1 The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 720mA rated inductor should be enough for most applications (600mA + 120mA). For better efficiency, choose a low DC-resistance inductor. Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3406B requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3406B applications. 8 U Table 1. Representative Surface Mount Inductors PART NUMBER Sumida CDRH3D16 VALUE (µH) 1.5 2.2 3.3 4.7 2.2 3.3 4.7 3.3 4.7 1.0 2.2 4.7 DCR (Ω MAX) 0.043 0.075 0.110 0.162 0.116 0.174 0.216 0.17 0.20 0.060 0.097 0.150 MAX DC SIZE CURRENT (A) W × L × H (mm3) 1.55 1.20 1.10 0.90 0.950 0.770 0.750 1.00 0.95 1.00 0.79 0.65 3.8 × 3.8 × 1.8 Sumida CMD4D06 Panasonic ELT5KT Murata LQH3C 3.5 × 4.3 × 0.8 4.5 × 5.4 × 1.2 2.5 × 3.2 × 2.0 (1) W UU CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN required IRMS ≅ IOMAX [V ( V OUT IN − VOUT )] 1/ 2 VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). 3406bfa LTC3406B APPLICATIO S I FOR ATIO Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ∆VOUT is determined by: ⎛ 1⎞ ∆VOUT ≅ ∆IL ⎜ ESR + ⎟ 8fC OUT ⎠ ⎝ where f = operating frequency, COUT = output capacitance and ∆IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ∆IL increases with input voltage. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3406B’s control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can U couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Output Voltage Programming (LTC3406B Only) In the adjustable version, the output voltage is set by a resistive divider according to the following formula: W UU ⎛ R2⎞ VOUT = 0.6V ⎜ 1 + ⎟ ⎝ R1⎠ (2) The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 3. 0.6V ≤ VOUT ≤ 5.5V R2 VFB LTC3406B GND 3406B F03 R1 Figure 3. Setting the LTC3406B Output Voltage Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. 3406bfa 9 LTC3406B APPLICATIO S I FOR ATIO Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3406B circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 4. 1 VIN = 3.6V 0.1 POWER LOSS (W) 0.01 VOUT = 2.5V VOUT = 1.8V VOUT = 1.2V 0.001 VOUT = 1.5V 0.0001 0.1 1 10 100 LOAD CURRENT (mA) 1000 3406B F04 Figure 4. Power Lost vs Load Current 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 10 U 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Charateristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. Thermal Considerations In most applications the LTC3406B does not dissipate much heat due to its high efficiency. But, in applications where the LTC3406B is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3406B from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(θJA) where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. 3406bfa W UU LTC3406B APPLICATIO S I FOR ATIO The junction temperature, TJ, is given by: T J = TA + TR where TA is the ambient temperature. As an example, consider the LTC3406B in dropout at an input voltage of 2.7V, a load current of 600mA and an ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the P-channel switch at 70°C is approximately 0.52Ω. Therefore, power dissipated by the part is: PD = ILOAD2 • RDS(ON) = 187.2mW For the SOT-23 package, the θJA is 250°C/ W. Thus, the junction temperature of the regulator is: TJ = 70°C + (0.1872)(250) = 116.8°C which is below the maximum junction temperature of 125°C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (∆ILOAD • ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. U A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 • CLOAD). Thus, a 10µF capacitor charging to 3.3V would require a 250µs rise time, limiting the charging current to about 130mA. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3406B. These items are also illustrated graphically in Figures 5 and 6. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. 2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground. 3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 4. Keep the switching node, SW, away from the sensitive VFB node. 5. Keep the (–) plates of CIN and COUT as close as possible. 3406bfa W UU 11 LTC3406B APPLICATIO S I FOR ATIO 1 RUN VFB 5 R2 LTC3406B 2 – VOUT COUT GND 4 CFWD VOUT 2 + 3 L1 SW VIN CIN BOLD LINES INDICATE HIGH CURRENT PATHS 3406B F05a Figure 5a. LTC3406B Layout Diagram VIA TO GND R1 VIA TO VIN R2 CFWD VIN VIA TO VOUT PIN 1 VOUT L1 SW LTC3406B COUT GND CIN 3406B F06a Figure 6a. LTC3406B Suggested Layout Design Example As a design example, assume the LTC3406B is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.6A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 2.5V. With this information we can calculate L using equation (1), L= ( )( ) ⎛V⎞ VOUT ⎜ 1 − OUT ⎟ VIN ⎠ ⎝ f ∆IL 1 12 U R1 1 RUN LTC3406B-1.8 5 W UU – COUT GND VOUT SW VIN CIN + 3 L1 4 + VIN – VIN 3406B F05b BOLD LINES INDICATE HIGH CURRENT PATHS Figure 5b. LTC3406B-1.8 Layout Diagram VIA TO VOUT VIA TO VIN VIN PIN 1 VOUT L1 SW LTC3406B-1.8 COUT GND CIN 3406B F06b Figure 6b. LTC3406B-1.8 Suggested Layout Substituting VOUT = 2.5V, VIN = 4.2V, ∆IL = 240mA and f = 1.5MHz in equation (3) gives: L= ⎛ 2.5V ⎞ 2.5V ⎜1 − ⎟ = 2.81µH 1.5MHz(240mA) ⎝ 4.2V ⎠ A 2.2µH inductor works well for this application. For best efficiency choose a 720mA or greater inductor with less than 0.2Ω series resistance. CIN will require an RMS current rating of at least 0.3A ≅ ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.25Ω. In most cases, a ceramic capacitor will satisfy this requirement. (3) 3406bfa LTC3406B APPLICATIO S I FOR ATIO For the feedback resistors, choose R1 = 316k. R2 can then be calculated from equation (2) to be: ⎛V ⎞ R2 = ⎜ OUT − 1⎟ R1 = 1000k ⎝ 0.6 ⎠ VIN 2.7V TO 4.2V 4 CIN† 4.7µF CER 1 VIN SW 3 2.2µH* 22pF VOUT 2.5V COUT** 10µF CER 1M 316k 3406B F07a EFFICIENCY (%) LTC3406B RUN GND 2 VFB 5 *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JHK316BJ106ML † TAIYO YUDEN JMK212BJ475MG Figure 7a TYPICAL APPLICATIO S Efficiency vs Output Current Single Li-Ion 1.5V/600mA Regulator for High Efficiency and Small Footprint VIN 2.7V TO 4.2V 4 CIN** 4.7µF CER VIN RUN VOUT GND 2 *MURATA LQH32CN2R2M33 **TAIYO YUDEN CERAMIC JMK212BJ475MG † TAIYO YUDEN CERAMIC JMK316BJ106ML 5 3406B TA05 SW 3 2.2µH* COUT1† 10µF CER LTC3406B-1.5 1 EFFICIENCY (%) Load Step VOUT 100mV/DIV AC COUPLED IL 500mA/DIV ILOAD 500mA/DIV VIN = 3.6V 20µs/DIV VOUT = 1.5V ILOAD = 0mA TO 600mA 3406B TA07 U Figure 7 shows the complete circuit along with its efficiency curve. 100 90 80 70 60 50 40 30 20 10 0.1 VIN = 3.6V VIN = 2.7V VOUT = 2.5V VIN = 4.2V 1 100 10 OUTPUT CURRENT (mA) 1000 3406B F07b W U UU Figure 7b 100 90 VOUT 1.5V VOUT = 1.5V VIN = 2.7V 80 70 60 50 40 30 20 10 0.1 1 100 10 OUTPUT CURRENT (mA) 1000 3406B TA06 VIN = 3.6V VIN = 4.2V Load Step VOUT 100mV/DIV AC COUPLED IL 500mA/DIV ILOAD 500mA/DIV VIN = 3.6V 20µs/DIV VOUT = 1.5V ILOAD = 200mA TO 600mA 3406B TA08 3406bfa 13 LTC3406B TYPICAL APPLICATIO S Single Li-Ion 1.2V/600mA Regulator for High Efficiency and Small Footprint 100 VIN 2.7V TO 4.2V 4 CIN† 4.7µF CER 1 VIN SW 3 2.2µH* 22pF VOUT 1.2V COUT** 10µF CER 301k 301k *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JHK316BJ106ML † TAIYO YUDEN JMK212BJ475MG 3406B TA09 LTC3406B RUN GND 2 VFB 5 EFFICIENCY (%) Load Step VOUT 100mV/DIV AC COUPLED IL 500mA/DIV ILOAD 500mA/DIV VIN = 3.6V 20µs/DIV VOUT = 1.2V ILOAD = 0mA TO 600mA 3406B TA11 14 U Efficiency vs Output Current VOUT = 1.2V VIN = 2.7V 90 80 70 60 50 40 30 20 10 0.1 VIN = 3.6V VIN = 4.2V 1 100 10 OUTPUT CURRENT (mA) 1000 3406B TA10 Load Step VOUT 100mV/DIV AC COUPLED IL 500mA/DIV ILOAD 500mA/DIV VIN = 3.6V 20µs/DIV VOUT = 1.2V ILOAD = 200mA TO 600mA 3406B TA12 3406bfa LTC3406B PACKAGE DESCRIPTIO 0.62 MAX 0.95 REF 3.85 MAX 2.62 REF RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.20 BSC 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF 0.09 – 0.20 (NOTE 3) NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U S5 Package 5-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1635) 2.90 BSC (NOTE 4) 1.22 REF 1.4 MIN 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE 0.30 – 0.45 TYP 5 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.01 – 0.10 1.90 BSC S5 TSOT-23 0302 3406bfa 15 LTC3406B TYPICAL APPLICATIO VIN 5V RELATED PARTS PART NUMBER LT1616 LT1676 LTC1701/LT1701B LT1776 LTC1877 LTC1878 LTC1879 LTC3403 LTC3404 LTC3405/LTC3405A LTC3406 LTC3411 LTC3412 LTC3440 DESCRIPTION 500mA (IOUT), 1.4MHz, High Efficiency Step-Down DC/DC Converter 450mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter 750mA (IOUT), 1MHz, High Efficiency Step-Down DC/DC Converter 500mA (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 1.2A (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter with Bypass Transistor 600mA (IOUT), 1.4MHz, Synchronous Step-Down DC/DC Converter 300mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter COMMENTS 90% Efficiency, VIN = 3.6V to 25V, VOUT = 1.25V, IQ = 1.9mA, ISD =