LTC3442EDE

LTC3442 Micropower Synchronous Buck-Boost DC/DC Converter with Automatic Burst Mode Operation DESCRIPTIO The LTC®3442 is a highly efficient, fixed frequency, BuckBoost DC/DC converter, which operates from input voltages above, below, and equal to the output voltage. The topology incorporated in the IC provides a continuous transfer function through all operating modes, making the product ideal for a single Lithium-Ion or multicell alkaline applications where the output voltage is within the battery voltage range. The device includes two 0.10Ω N-channel MOSFET switches and two 0.10Ω P-channel switches. Operating frequency and average input current limit can each be programmed with an external resistor. Quiescent current is only 35µA in Burst Mode operation, maximizing battery life in portable applications. Automatic Burst Mode operation allows the user to program the load current for Burst Mode operation, or to control it manually. Other features include 1µA shutdown current, programmable soft-start, peak current limit and thermal shutdown. The LTC3442 is available in a low profile, thermally enhanced 12-lead (4mm × 3mm × 0.75mm) DFN package. , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a registered trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 6404251, 6166527. FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ Regulated Output with Input Voltages Above, Below, or Equal to the Output Single Inductor, No Schottky Diodes Required Manual or Programmable Automatic Burst Mode® Operation Programmable Average Input Current Limit Up to 1.2A Continuous Output Current from a Single Lithium-Ion Cell High Efficiency: Up to 95% Output Disconnect in Shutdown 2.4V to 5.5V Input Range 2.4V to 5.25V Output Range 35µA Quiescent Current in Burst Mode Operation Programmable Frequency from 300kHz to 2MHz VOUT) Switch D is always on and switch C is always off during this mode. When the internal control voltage, VCI, is above voltage V1, output A begins to switch. During the off-time of switch A, synchronous switch B turns on for the remainder of the time. Switches A and B will alternate similar to a typical synchronous buck regulator. As the control voltage increases, the duty cycle of switch A increases until the maximum duty cycle of the converter in buck mode reaches DMAX_BUCK, given by: DMAX_BUCK = 100 – D4SW % where D4SW = duty cycle % of the four switch range. D4SW = (150ns • f) • 100 % where f = operating frequency, Hz. Beyond this point the “four switch,” or buck/boost region is reached. PMOS D SW2 6 VOUT 8 Buck/Boost or Four Switch (VIN ~ VOUT) When the internal control voltage, VCI, is above voltage V2, switch pair AD remain on for duty cycle DMAX_BUCK, and the switch pair AC begins to phase in. As switch pair AC phases in, switch pair BD phases out accordingly. When the VCI voltage reaches the edge of the buck/boost range, at voltage V3, the AC switch pair completely phase out the BD pair, and the boost phase begins at duty cycle D4SW. The input voltage, VIN, where the four switch region begins is given by: NMOS C 3442 F01 V4 (≈ 2.05V) V3 (≈ 1.65V) V2 (≈ 1.55V) VIN = VOUT 1 – (150ns • f) The point at which the four switch region ends is given by: VIN = VOUT(1 – D) = VOUT(1 – 150ns • f) V 3442 F02 9 LTC3442 OPERATIO Boost Region (VIN < VOUT) Switch A is always on and switch B is always off during this mode. When the internal control voltage, VCI, is above voltage V3, switch pair CD will alternately switch to provide a boosted output voltage. This operation is typical to a synchronous boost regulator. The maximum duty cycle of the converter is limited to 88% typical and is reached when VCI is above V4. BURST MODE OPERATION Burst Mode operation occurs when the IC delivers energy to the output until it is regulated and then goes into a sleep mode where the outputs are off and the IC is consuming only 35µA of quiescent current from VIN. In this mode the output ripple has a variable frequency component that depends upon load current, and will typically be about 2% peak-to-peak. Burst Mode operation ripple can be reduced slightly by using more output capacitance (47µF or greater). Another method of reducing Burst Mode operation ripple is to place a small feed-forward capacitor across the upper resistor in the VOUT feedback divider network (as in Type III compensation). During the period where the device is delivering energy to the output, the peak switch current will be equal to 900mA typical and the inductor current will terminate at zero current for each cycle. In this mode the typical maximum average output current is given by: SW1 B L SW2 C IINDUCTOR 0.2 • VIN IOUT(MAX)BURST ≈ A VOUT + VIN Note that the peak efficiency during Burst Mode operation is less than the peak efficiency during fixed frequency VIN 9 A 4 SW1 B dI ≈ VIN dt L L D VOUT 8 + – SW2 C IINDUCTOR 5 GND Figure 3. Inductor Charge Cycle During Burst Mode Operation 3442fa 10 U VIN 9 A 4 dI ≈ – VOUT L dt D 6 900mA VOUT 8 – + 0mA T2 3442 F04 5 GND Figure 4. Inductor Discharge Cycle During Burst Mode Operation because the part enters full-time 4-switch mode (when servicing the output) with discontinuous inductor current as illustrated in Figures 3 and 4. During Burst Mode operation, the control loop is nonlinear and cannot utilize the control voltage from the error amp to determine the control mode, therefore full-time 4-switch mode is required to maintain the Buck/Boost function. The efficiency below 1mA becomes dominated primarily by the quiescent current. The Burst Mode operation efficiency is given by: EFFICIENCY ≅ n • ILOAD 35µA + ILOAD where n is typically 82% during Burst Mode operation. Automatic Burst Mode Operation Control Burst Mode operation can be automatic or manually controlled with a single pin. In automatic mode, the IC will enter Burst Mode operation at light load and return to fixed frequency operation at heavier loads. The load current at which the mode transition occurs is programmed using a single external resistor from the BURST pin to ground, according to the following equations: 17.6 RBURST 22.4 Leave Burst Mode: I = RBURST Enter Burst Mode: I = where RBURST is in kΩ and IBURST is the load transition 6 900mA 0mA T1 3442 F03 LTC3442 OPERATIO current in Amps. Do not use values of RBURST greater than 250k. For automatic operation, a filter capacitor should also be connected from BURST to ground to prevent ripple on BURST from causing the IC to oscillate in and out of Burst Mode operation. The equation for the minimum capacitor value is: CBURST(MIN) ≥ COUT • VOUT 60, 000 where CBURST(MIN) and COUT are in µF In the event that a load transient causes the feedback pin to drop by more than 4% from the regulation value while in Burst Mode operation, the IC will immediately switch to fixed frequency mode and an internal pull-up will be momentarily applied to BURST, rapidly charging the BURST cap. This prevents the IC from immediately reentering Burst Mode operation once the output achieves regulation. Manual Burst Mode Operation For manual control of Burst Mode operation, the RC network connected to BURST can be eliminated. To force fixed frequency mode, BURST should be connected to VOUT. To force Burst Mode operation, BURST should be grounded. When commanding Burst Mode operation manually, the circuit connected to BURST should be able to sink up to 2mA. U For optimum transient response with large dynamic loads, the operating mode should be controlled manually by the host. By commanding fixed frequency operation prior to a sudden increase in load, output voltage droop can be minimized. Note that if the load current applied during forced Burst Mode operation (BURST pin is grounded) exceeds the current that can be supplied, the output voltage will start to droop and the IC will automatically come out of Burst Mode operation and enter fixed frequency mode, raising VOUT. Once regulation is achieved, the IC will then enter Burst Mode operation once again, and the cycle will repeat, resulting in about 4% output ripple. Note that Burst Mode operation is inhibited during soft-start. Burst Mode Operation to Fixed Frequency Transient Response In Burst Mode operation, the compensation network is not used and VC is disconnected from the error amplifier. During long periods of Burst mode operation, leakage currents in the external components or on the PC board could cause the compensation capacitor to charge (or discharge), which could result in a large output transient when returning to fixed frequency mode of operation, even at the same load current. To prevent this, the LTC3442 incorporates an active clamp circuit that holds the voltage on VC at an optimal voltage during Burst Mode operation. This minimizes any output transient when returning to fixed frequency mode operation. For optimum transient 3442fa 11 LTC3442 OPERATIO response, Type 3 compensation is also recommended to broad band the control loop and roll off past the two pole response of the output LC filter. (See Closing the Feedback Loop.) Soft-Start The soft-start function is combined with shutdown. When the SHDN/SS pin is brought above 1V typical, the IC is enabled but the EA duty cycle is clamped from VC. A TO PWM COMPARATORS 12 U detailed diagram of this function is shown in Figure 5. The components RSS and CSS provide a slow ramping voltage on SHDN/SS to provide a soft-start function. To ensure that VC is not being clamped, SHDN/SS must be raised above 2.4V. To enable Burst Mode operation, SHDN/SS must be raised to within 0.5V of VIN. VIN 14µA ERROR AMP VOUT FB R1 + – SOFT-START CLAMP VCI 1.22V 12 VC 11 CP1 R2 SHDN/SS 1 CSS RSS ENABLE SIGNAL 3442 F05 + CHIP ENABLE – 1V Figure 5. Soft-Start Circuitry 3442fa LTC3442 APPLICATIO S I FOR ATIO COMPONENT SELECTION VIN 1 SHDN/SS 2 RT 3 SGND 4 SW1 5 PGND 6 SW2 FB 12 VC 11 RLIM 10 VIN 9 VOUT 8 BURST 7 VIN GND RT MULTIPLE VIAS 3442 F06 Figure 6. Recommended Component Placement. Traces Carrying High Current Should be Short and Wide. Trace Area at FB and VC Pins are Kept Low. Lead Length to Battery Should be Kept Short. VOUT and VIN Ceramic Capacitors Close to the IC Pins. Inductor Selection The high frequency operation of the LTC3442 allows the use of small surface mount inductors. The inductor ripple current is typically set to 20% to 40% of the maximum inductor current. For a given ripple the inductance terms are given as follows: L BOOST > L BUCK > VIN(MIN) • ( VOUT – VIN(MIN) ) f • ∆ IL • VOUT • ( VIN(MAX ) – VOUT ) H VOUT f • ∆ IL • VIN(MAX ) H Table 1. Inductor Vendor Information SUPPLIER Coilcraft CoEv Magnetics Murata Sumida TDK TOKO PHONE (847) 639-6400 (800) 227-7040 (814) 237-1431 (800) 831-9172 USA: (847) 956-0666 Japan: 81(3) 3607-5111 (847) 803-6100 (847) 297-0070 FAX (847) 639-1469 (650) 361-2508 (814) 238-0490 USA: (847) 956-0702 Japan: 81(3) 3607-5144 (847) 803-6296 (847) 699-7864 WEB SITE www.coilcraft.com www.circuitprotection.com/magnetics.asp www.murata.com www.sumida.com www.component.tdk.com www.tokoam.com 3442fa U where f = operating frequency, Hz ∆IL = maximum allowable inductor ripple current, A VIN(MIN) = minimum input voltage, V VIN(MAX) = maximum input voltage, V VOUT = output voltage, V VOUT W U U IOUT(MAX) = maximum output load current For high efficiency, choose a ferrite inductor with a high frequency core material to reduce core loses. The inductor should have low ESR (equivalent series resistance) to reduce the I2R losses, and must be able to handle the peak inductor current without saturating. Molded chokes or chip inductors usually do not have enough core to support the peak inductor currents in the 1A to 2A region. To minimize radiated noise, use a shielded inductor. See Table 1 for a suggested list of inductor suppliers. Output Capacitor Selection The bulk value of the output filter capacitor is set to reduce the ripple due to charge into the capacitor each cycle. The steady state ripple due to charge is given by: % RIPPLE_BOOST = IOUT(MAX) • (VOUT – VIN(MIN) ) •100 COUT • VOUT 2 • f % RIPPLE_BUCK = % IOUT(MAX) • (VIN(MAX) – VOUT ) •100 COUT • VIN(MAX) • VOUT • f % where COUT = output filter capacitor in Farads and f = switching frequency in Hz. 13 LTC3442 APPLICATIO S I FOR ATIO The output capacitance is usually many times larger than the minimum value in order to handle the transient response requirements of the converter. For a rule of thumb, the ratio of the operating frequency to the unity-gain bandwidth of the converter is the amount the output capacitance will have to increase from the above calculations in order to maintain the desired transient response. The other component of ripple is due to the ESR (equivalent series resistance) of the output capacitor. Low ESR capacitors should be used to minimize output voltage ripple. For surface mount applications, Taiyo Yuden or TDK ceramic capacitors, AVX TPS series tantalum capacitors or Sanyo POSCAP are recommended. See Table 2 for contact information. Input Capacitor Selection Since VIN is the supply voltage for the IC, as well as the input to the power stage of the converter, it is recommended to place at least a 4.7µF, low ESR ceramic bypass capacitor close to the VIN and SGND pins. It is also important to minimize any stray resistance from the converter to the battery or other power source. Optional Schottky Diodes The Schottky diodes across the synchronous switches B and D are not required (VOUT < 4.3V), but provide a lower drop during the break-before-make time (typically 15ns) improving efficiency. Use a surface mount Schottky diode such as an MBRM120T3 or equivalent. Do not use ordinary rectifier diodes, since the slow recovery times will compromise efficiency. For applications with an output voltage above 4.3V, a Schottky diode is required from SW2 to VOUT. Output Voltage < 2.4V The LTC3442 can operate as a buck converter with output Table 2. Capacitor Vendor Information SUPPLIER AVX Murata Sanyo Taiyo Yuden TDK PHONE (803) 448-9411 (814) 237-1431 (800) 831-9172 (619) 661-6322 (408) 573-4150 (847) 803-6100 FAX 14 U voltages as low as 0.4V. The part is specified at 2.4V minimum to allow operation without the requirement of a Schottky diode. Synchronous switch D is powered from VOUT and the RDS(ON) will increase at low output voltages, therefore a Schottky diode is required from SW2 to VOUT to provide the conduction path to the output. Note that Burst Mode operation is inhibited at output voltages below 1.6V typical. Output Voltage > 4.3V A Schottky diode from SW2 to VOUT is required for output voltages over 4.3V. The diode must be located as close to the pins as possible in order to reduce the peak voltage on SW2 due to the parasitic lead and trace inductance. Input Voltage > 4.5V For applications with input voltages above 4.5V which could exhibit an overload or short-circuit condition, a 2Ω/1nF series snubber is required between SW1 and GND. A Schottky diode from SW1 to VIN should also be added as close to the pins as possible. For the higher input voltages, VIN bypassing becomes more critical; therefore, a ceramic bypass capacitor as close to the VIN and SGND pins as possible is also required. Operating Frequency Selection Higher operating frequencies allow the use of a smaller inductor and smaller input and output filter capacitors, thus reducing board area and component height. However, higher operating frequencies also increase the IC’s total quiescent current due to the gate charge of the four switches, as given by: Buck: Boost: Iq = (0.8 • VIN • f) mA Iq = [0.4 • (VIN + VOUT) • f] mA Buck/Boost: Iq = [f • (1.2 • VIN + 0.4 • VOUT)] mA WEB SITE www.avxcorp.com www.murata.com www.sanyovideo.com www.t-yuden.com www.component.tdk.com 3442fa W UU (803) 448-1943 (814) 238-0490 (619) 661-1055 (408) 573-4159 (847) 803-6296 LTC3442 APPLICATIO S I FOR ATIO where f = switching frequency in MHz. Therefore frequency selection is a compromise between the optimal efficiency and the smallest solution size. Closing the Feedback Loop The LTC3442 incorporates voltage mode PWM control. The control to output gain varies with operation region (buck, boost, buck/boost), but is usually no greater than 15. The output filter exhibits a double pole response, as given by: f FILTER—POLE = (in buck mode) f FILTER—POLE = VIN 2 • VOUT • π • L • COUT Hz 1 2 • π • L • COUT Hz Figure 7. Error Amplifier with Type I Compensation (in boost mode) where L is in henries and COUT is in farads. The output filter zero is given by: Hz 2 • π • RESR • COUT where RESR is the equivalent series resistance of the output cap. A troublesome feature in boost mode is the right-half plane zero (RHP), given by: f FILTER— ZERO = 1 VIN f RHPZ = Hz 2 • π • IOUT • L • VOUT The loop gain is typically rolled off before the RHP zero frequency. A simple Type I compensation network can be incorporated to stabilize the loop, but at a cost of reduced bandwidth and slower transient response. To ensure proper phase margin using Type I compensation, the loop must be crossed over a decade before the LC double pole. The unity-gain frequency of the error amplifier with the Type I compensation is given by: 2 U fUG = 1 Hz 2 • π • R1 • CP1 referring to Figure 7. VOUT W UU + ERROR AMP 1.22V FB 12 VC 11 CP1 R2 3442 F07 R1 – Most applications demand an improved transient response to allow a smaller output filter capacitor. To achieve a higher bandwidth, Type III compensation is required, providing two zeros to compensate for the double-pole response of the output filter. Referring to Figure 8, the location of the poles and zeros are given by: Hz 2 • π • 32e3 • R1 • CP1 (which is extremely close to DC) 1 Hz 2 • π • RZ • CP1 1 fZERO2 = Hz 2 • π • R1 • CZ1 1 fPOLE2 = Hz 2 • π • RZ • CP2 fZERO1 = where resistance is in ohms and capacitance is in farads. VOUT fPOLE1 ≅ 1 + ERROR AMP 1.22V FB 12 VC 11 CP2 3442 F08 R1 CZ1 – RZ CP1 R2 Figure 8. Error Amplifier with Type III Compensation 3442fa 15 LTC3442 TYPICAL APPLICATIO S 1MHz Li-Ion to 3.3V at 1.2A Converter with Manual Mode Control (and Peak Current Limit Only) L1 3.3µH 2.5V TO 4.2V CIN 10µF Li-Ion Multi-Input 3.3V at 600mA Boost Converter for Portable Applications with Automatic Burst Mode Operation and Average Input Current Limit for USB Powered Devices 1nF L1 4.7µH Li-Ion USB/5V CIN 10µF 143k 2N7002 USB PRESENT 0.01µF RSNUB** 1Ω 1nF 143k 64.9k D1 MBRM120T3 2.5V TO 5.5V 1M SW1 VIN SW2 LTC3442 VOUT FB VC BURST PGND 200k 0.01µF 15k 340k 2.2k 220pF 470pF 200k COUT 22µF VOUT 3.3V 600mA **A SNUBBER RESISTOR IS REQUIRED TO PREVENT CIN: TAIYO YUDEN JMK212BJ106MG RINGING IF THERE IS SIGNIFICANT INPUT INDUCTANCE, COUT: TAIYO YUDEN JMK325BJ476MM L1: TDK RLF7030T-4R7M3R4 SUCH AS FROM A USB CABLE 16 U SW2 LTC3442 VOUT VIN SW1 1M SHDN/SS RLIM 0.01µF RT 43.2k SGND FB VC BURST PGND 15k 340k 2.2k 220pF 470pF 200k VOUT 3.3V 1.2A COUT 22µF BURST FIXED FREQ CIN: TAIYO YUDEN JMK212BJ106MG COUT: TAIYO YUDEN JMK325BJ226MM L1: TDK RLF7030T-3R3M4R 3442 TA02 2Ω SHDN/SS RLIM RT SGND 3442 TA03 3442fa LTC3442 TYPICAL APPLICATIO S High Efficiency Li-Ion Powered Constant Current LED Driver with Open-LED Protection R5 7.87k VIN 2.5V TO 4.2V *OFF ON 10µF RLIM VC SW1 VIN 3.3µH OPEN LED VOLTAGE LIMIT = (R4+R5)*0.95/R4 * NOTE: THE SHDN/SS VOLTAGE MUST BE NO MORE THAN 0.5V BELOW VIN WHEN ENABLED. EFFICIENCY (%) U SW2 VOUT LTC3442 VOUT ILED = 500mA SHDN/SS FB 1nF R2 200k 4.7µF LHXL-PW03 RT R4 2k SGND 57.6k BURST PGND R3 95.3k 47pF R1 301k 3442 TA04a R2 = R1/1.5 ILED = 24 • (R1+R2+R3)/(R1 • R3) AMPS LED Driver Efficiency vs LED Current 100 98 96 94 92 90 88 86 84 82 80 0.1 LED CURRENT (A) 3442 TA04b VIN = 3.6V 750kHz 1.0 3442fa 17 LTC3442 TYPICAL APPLICATIO S High Current LED Driver with Low/High Current Range for Pulsed Applications; LED Current is 0.5A with 1.5A Pulse R5 7.87k VIN 2.7V TO 4.2V *OFF ON 10µF 6.3V RLIM SW1 VIN R4 2k OPEN LED VOLTAGE LIMIT = (R4+R5) • 0.95/R4 LOW HI * NOTE: THE SHDN/SS VOLTAGE MUST BE NO MORE THAN 0.5V BELOW VIN WHEN ENABLED. 18 U U 3.3µH SW2 VOUT LTC3442 VOUT ILED = 500mA/1.5A SHDN/SS FB 1nF VC R2 200k R2 20k 10µF 6.3V LHXL-PW03 RT 57.6k SGND BURST PGND 95.3k 1nF R1 301k 40.2k 2N7002 R2 = R1/1.5 ILED = 24 • (R1+R2+R3)/(R1 • R3) AMPS (OR: ILED = 40/R3 + .08) 3442 TA05 3442fa LTC3442 PACKAGE DESCRIPTIO 3.50 ± 0.05 1.70 ± 0.05 2.20 ± 0.05 (2 SIDES) 0.25 ± 0.05 3.30 ± 0.05 (2 SIDES) 0.50 BSC RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS 4.00 ± 0.10 (2 SIDES) R = 0.20 TYP 3.00 ± 0.10 (2 SIDES) 1.70 ± 0.10 (2 SIDES) PIN 1 NOTCH (UE12/DE12) DFN 0603 PIN 1 TOP MARK (NOTE 6) 0.200 REF NOTE: 1. DRAWING PROPOSED TO BE A VARIATION OF VERSION (WGED) IN JEDEC PACKAGE OUTLINE M0-229 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U UE/DE Package 12-Lead Plastic DFN (4mm × 3mm) (Reference LTC DWG # 05-08-1695) 0.65 ± 0.05 PACKAGE OUTLINE 7 R = 0.115 TYP 0.38 ± 0.10 12 0.75 ± 0.05 6 0.25 ± 0.05 3.30 ± 0.10 (2 SIDES) 1 0.50 BSC 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD 3442fa 19 LTC3442 RELATED PARTS PART NUMBER LT 1613 LT1618 LT1930/LT1930A LT1935 LT1946/LT1946A LT1961 ® DESCRIPTION 550mA (ISW), 1.4MHz, High Efficiency Step-Up DC/DC Converter 1.5A (ISW), 1.25MHz, High Efficiency Step-Up DC/DC Converter 1A (ISW), 1.2MHz/2.2MHz, High Efficiency Step-Up DC/DC Converter 2A (ISW), 1.2MHz, 38V Step-Up DC/DC Converter 1.5A (ISW), 1.2MHz/2.7MHz, High Efficiency Step-Up DC/DC Converter 1.5A (ISW), 1.25MHz, High Efficiency Step-Up DC/DC Converter COMMENTS VIN: 0.9V to 10V, VOUT(MAX) = 34V, IQ = 3mA, ISD < 1µA, ThinSOTTM Package VIN: 1.6V to 18V, VOUT(MAX) = 35V, IQ = 1.8mA, ISD < 1µA, MS10 Package VIN: 2.6V to 16V, VOUT(MAX) = 34V, IQ = 4.2mA/5.5mA, ISD < 1µA, ThinSOT Package VIN: 2.3V to 16V, VOUT(MAX) = 38V, IQ = 3mA, ISD < 1µA, ThinSOT Package VIN: 2.45V to 16V, VOUT(MAX) = 34V, IQ = 3.2mA, ISD < 1µA, MS8 Package VIN: 3V to 25V, VOUT(MAX) = 35V, IQ = 0.9mA, ISD = 6µA, MS8E Package VIN: 0.85V to 5V, VOUT(MAX) = 5V, IQ = 19µA/300µA, ISD < 1µA, ThinSOT Package VIN: 0.5V to 5V, VOUT(MAX) = 6V, IQ = 38µA, ISD < 1µA, MS Package VIN: 2.7V to 6V, VOUT(MIN) = 0.8V, IQ = 20µA, ISD ≤ 1µA, MS10 Package VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20µA, ISD ≤ 1µA, ThinSOT Package VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA, ISD ≤ 1µA, MS Package VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA, ISD ≤ 1µA, MS Package VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA, ISD ≤ 1µA, TSSOP16E Package VIN: 0.5V to 4.5V, VOUT(MAX) = 5.25V, IQ = 12µA, ISD < 1µA, QFN Package VIN: 0.5V to 4.5V, VOUT(MAX) = 5.25V, IQ = 12µA, ISD < 1µA, QFN Package VIN: 0.5V to 4.4V, VOUT(MIN) = 5V, IQ = 20µA, ISD < 1µA, QFN Package VIN: 3V to 25V, VOUT(MAX) = 34V, IQ = 0.9mA, ISD < 6µA, TSSOP-16E Package VIN: 2.5V to 5.5V, VOUT(MIN) = 5.5V, IQ = 25µA, ISD < 1µA, MS, DFN Packages VIN: 2.5V to 5.5V, VOUT(MIN) = 5.5V, IQ = 25µA, ISD < 1µA, DFN Package VIN: 2.4V to 5.5V, VOUT(MIN) = 5.25V, IQ = 28µA, ISD < 1µA, MS Package VIN: 2.6V to 16V, VOUT(MAX) = 40V, IQ = 1.2mA, ISD < 1µA, ThinSOT Package LTC3400/LTC3400B 600mA (ISW), 1.2MHz Synchronous Step-Up DC/DC Converter LTC3401/LTC3402 1A/2A (ISW), 3MHz Synchronous Step-Up DC/DC Converter LTC3405/LTC3405A 300mA (IOUT), 1.5MHz Synchronous Step-Down DC/DC Converter LTC3406/LTC3406B 600mA (IOUT), 1.5MHz Synchronous Step-Down DC/DC Converter LTC3407 LTC3411 LTC3412 LTC3421 LTC3425 LTC3429 LTC3436 LTC3440 LTC3441 LTC3443 LT3467 600mA (IOUT), 1.5MHz Dual Synchronous Step-Down DC/DC Converter 1.25A (IOUT), 4MHz Synchronous Step-Down DC/DC Converter 2.5A (IOUT), 4MHz Synchronous Step-Down DC/DC Converter 3A (ISW), 3MHz Synchronous Step-Up DC/DC Converter 5A (ISW), 8MHz Multiphase Synchronous Step-Up DC/DC Converter 600mA (ISW), 500kHz Synchronous Step-Up DC/DC Converter 3A (ISW), 1MHz, 34V Step-Up DC/DC Converter 600mA (IOUT), 2MHz Synchronous Buck-Boost DC/DC Converter 600mA (IOUT), 2MHz Synchronous Buck-Boost DC/DC Converter 1.2A (IOUT), 600kHz Synchronous Buck-Boost DC/DC Converter 1.1A (ISW), 1.3MHz, High Efficiency Step-Up DC/DC Converter ThinSOT is a trademark of Linear Technology Corporation. 3442fa 20 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● LT 0306 REV A • PRINTED IN USA www.linear.com © LINEAR TECHNOLOGY CORPORATION 2004