LTC3564ES5-TRPBF

LTC3564 2.25MHz, 1.25A Synchronous Step-Down Regulator FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO High Efficiency: Up to 96% Very Low Quiescent Current: Only 20μA 1.25A Output Current 2.5V to 5.5V Input Voltage Range 2.25MHz Constant Frequency Operation No Schottky Diode Required Low Dropout Operation: 100% Duty Cycle 0.6V Reference Allows Low Output Voltages Shutdown Mode Draws ≤ 1μA Supply Current Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Low Profile (1mm) ThinSOTTM and 6-Lead (2mm × 3mm) DFN Packages The LTC ®3564 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current during operation is only 20μA, dropping to ≤ 1μA in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3564 ideally suited for single Li-Ion battery-powered or 3.3V to 5V input voltage applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. Automatic Burst Mode® operation increases efficiency at light loads, further extending battery runtime. Switching frequency is internally set at 2.25MHz, allowing the use of small surface mount inductors and capacitors. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. Low output voltages are easily supported with the 0.6V feedback reference voltage. The LTC3564 is available in low profile (1mm) ThinSOT and 6-Lead (2mm × 3mm) DFN packages. , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. Burst Mode is a registered trademark of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents including 5481178, 6580258, 6304066, 6498466, 6611131. APPLICATIO S ■ ■ ■ ■ ■ ■ Cellular Telephones Wireless and DSL Modems Digital Still Cameras Media Players Portable Instruments Point of Load Regulation TYPICAL APPLICATIO 100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 VOUT = 1.8V 1.1μH VIN CIN 22μF CER VIN SW 22pF COUT 22μF CER 634k 316k 3564 TA01a VOUT 1.8V LTC3564 RUN GND VFB 0 0.1 1 10 100 1000 OUTPUT CURRENT (mA) U 10 1 POWER LOSS (W) 0.1 0.01 0.001 VIN = 2.7V VIN = 3.6V VIN = 4.2V 0.0001 0.00001 10000 3564 TA01b U U 3564f 1 LTC3564 ABSOLUTE AXI U RATI GS (Note 1) Operating Junction Temperature Range (Notes 2, 3, 6) ...................................... – 40°C to 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C Input Supply Voltage .................................. – 0.3V to 6V RUN, VFB Voltages ..................................... – 0.3V to VIN SW Voltage (DC) ......................... – 0.3V to (VIN + 0.3V) PI CO FIGURATIO TOP VIEW VFB 1 GND 2 VIN 3 5 RUN 4 SW S5 PACKAGE 5-LEAD PLASTIC TSOT-23 TJMAX = 125°C, θJA = 215°C/ W, θJC = 50°C/ W ORDER I FOR ATIO LEAD FREE FINISH LTC3564ES5#PBF LTC3564IS5#PBF LTC3564EDCB#PBF LTC3564IDCB#PBF TAPE AND REEL LTC3564ES5#TRPBF LTC3564IS5#TRPBF LTC3564EDCB#TRPBF LTC3564IDCB#TRPBF Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on nonstandard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ The ● denotes specifications which apply over the full operating junction temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified. SYMBOL IVFB VFB ΔVFB IPK VLOADREG VIN PARAMETER Feedback Current Regulated Feedback Voltage Reference Voltage Line Regulation Peak Inductor Current Output Voltage Load Regulation Input Voltage Range ● ELECTRICAL CHARACTERISTICS CONDITIONS ● (Note 4) VIN = 2.5V to 5.5V (Note 4) VIN = 3V, VFB = 0.5V, Duty Cycle < 35% 2 U U WW U W W U U U TOP VIEW SW 1 PGND 2 VIN 3 7 6 RUN 5 SGND 4 VFB DCB PACKAGE 6-LEAD (2mm × 3mm) PLASTIC DFN TJMAX = 125°C, θJA = 64°C/ W, θJC = 10.6°C/ W EXPOSED PAD (PIN 7) IS SGND, MUST BE SOLDERED TO PCB PART MARKING* LTCYJ LTCYJ LDTQ LDTQ PACKAGE DESCRIPTION 5-Lead Plastic TSOT-23 5-Lead Plastic TSOT-23 6-Lead (2mm × 3mm) Plastic DFN 6-Lead (2mm × 3mm) Plastic DFN TEMPERATURE RANGE –40°C to 125°C –40°C to 125°C –40°C to 125°C –40°C to 125°C MIN ● TYP 0.6 0.04 2.0 0.5 MAX ± 30 0.6120 0.4 2.5 5.5 UNITS nA V %/V A % V 0.5880 1.5 2.5 3564f LTC3564 The ● denotes specifications which apply over the full operating junction temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified. SYMBOL IS PARAMETER Input DC Bias Current Active Mode Sleep Mode Shutdown Oscillator Frequency RDS(ON) of P-Channel FET RDS(ON) of N-Channel FET SW Leakage RUN Threshold RUN Leakage Current Soft-Start Time VFB from 10% to 90% Full Scale CONDITIONS (Note 5) VFB = 0.5V or VOUT = 90%, ILOAD = 0A VFB = 0.62V or VOUT = 103%, ILOAD = 0A VRUN = 0V, VIN = 4.2V VFB = 0.6V or VOUT = 100% S5 Package DCB Package S5 Package DCB Package VRUN = 0V, VSW = 0V or 5V, VIN = 5V ● ● ● ELECTRICAL CHARACTERISTICS MIN TYP 300 20 0.1 MAX 400 35 1 2.7 0.2 0.2 ±1 1.5 ±1 1.2 UNITS μA μA μA MHz Ω Ω Ω Ω μA V μA ms fOSC RPFET RNFET ILSW VRUN IRUN tSOFTSTART 1.8 2.25 0.15 0.15 0.15 0.15 ± 0.01 0.3 0.6 1 ± 0.01 0.9 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliabilty and lifetime. Note 2: The LTC3564E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction termperature range are assured by design, characterization and correlation with statistical process controls. The LTC3564I is guaranteed over the full –40°C to 125°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125°C. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3564ES5: TJ = TA + (PD)(215°C/W) LTC3564EDCB: TJ = TA + (PD)(64°C/W) Note 4: The LTC3564 is tested in a proprietary test mode. Note 5: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 6: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. 3564f 3 LTC3564 TYPICAL PERFOR A CE CHARACTERISTICS Efficiency vs Input Voltage 100 IOUT = 100mA 90 EFFICIENCY (%) EFFICIENCY (%) EFFICIENCY (%) 80 70 60 50 40 IOUT = 1.25A IOUT = 10mA IOUT = 1mA IOUT = 0.1mA VOUT = 1.8V 2.5 3.0 3.5 4.0 4.5 INPUT VOLTAGE (V) 5.0 5.5 3564 G01 Reference Voltage vs Temperature 615 OSCILLATOR FREQUENCY (MHz) 2.30 2.25 2.20 2.15 2.10 2.05 FREQUENCY VARIATION (%) 610 REFERENCE VOLTAGE (mV) 605 600 595 590 585 –50 –25 50 25 75 0 TEMPERATURE (°C) Load Regulation 1.00 0.75 VOUT ERROR (%) VOUT = 1.8V VOUT ERROR (%) 0.50 0.25 0 –0.25 –0.50 0 200 400 600 800 1000 1200 1400 OUTPUT CURRENT (mA) 3564 G07 RDS(ON) (Ω) 4 UW 100 TA = 25°C, VIN = 3.6V, unless otherwise specified. Efficiency vs Output Current 100 90 80 70 60 50 40 30 20 10 0 0.1 1 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10 100 1000 OUTPUT CURRENT (mA) 10000 3564 G02 Efficiency vs Output Current 100 90 80 70 60 50 40 30 20 10 0 0.1 1 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10 100 1000 OUTPUT CURRENT (mA) 10000 3564 G04 VOUT = 1.2V VOUT = 1.5V Oscillator Frequency vs Temperature 2.40 2.35 6 4 2 0 –2 –4 –6 Frequency Variation vs VIN 125 2.00 –50 –25 0 50 75 25 TEMPERATURE (°C) 100 125 –8 2.5 3.0 3.5 4.0 VIN (V) 4.5 5.0 5.5 3564 G06 3564 G04 3564 G05 Line Regulation 0.6 0.4 0.2 0 –0.2 0.05 VOUT = 1.8V ILOAD = 400mA 0.25 RDS(ON) vs Input Voltage 0.20 MAIN SWITCH 0.15 0.10 SYNCHRONOUS SWITCH –0.4 –0.6 0 2.5 3.0 3.5 4.0 4.5 INPUT VOLTAGE (V) 5.0 5.5 3564 G08 2.5 3.0 3.5 4.0 4.5 INPUT VOLTAGE (V) 5.0 5.5 3564 G09 3564f LTC3564 TYPICAL PERFOR A CE CHARACTERISTICS RDS(ON) vs Temperature 0.30 0.25 SUPPLY CURRENT (μA) 35 0.20 RDS(ON) (Ω) SUPPLY CURRENT (μA) MAIN SWITCH 0.15 0.10 0.05 0 –50 –25 SYNCHRONOUS SWITCH 50 25 75 0 TEMPERATURE (°C) Switch Leakage vs Temperature 600 SYNCHRONOUS SWITCH MAIN SWITCH 500 SWITCH LEAKAGE (nA) SWITCH LEAKAGE (pA) 400 300 200 100 0 –50 –25 50 25 75 0 TEMPERATURE (°C) UW 100 3564 G10 TA = 25°C, VIN = 3.6V, unless otherwise specified. Supply Current vs Supply Voltage 40 35 30 25 20 15 Supply Current vs Temperature VIN = 3.6V RUN = VIN ILOAD = 0A 30 25 20 15 10 125 2.5 3.0 3.5 4.0 4.5 SUPPLY VOLTAGE (V) 5.0 5.5 3564 G11 10 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 125 3564 G12 Switch Leakage vs Input Voltage 2500 RUN = 0V 2000 MAIN SWITCH 1500 1000 SYNCHRONOUS SWITCH 500 100 125 0 0 1 2 3 4 INPUT VOLTAGE (V) 5 6 3564 G14 3564 G13 3564f 5 LTC3564 TYPICAL PERFOR A CE CHARACTERISTICS Burst Mode Operation RUN 2V/DIV SW 2V/DIV VOUT 50mV/DIV AC COUPLED IL 200mA/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 40mA 2.5μs/DIV Load Step VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV VIN = 3.6V 20μs/DIV VOUT = 1.8V ILOAD = 0A TO 1.25A 3564 G18 6 UW TA = 25°C, VIN = 3.6V, unless otherwise specified. Start-Up from Shutdown RUN 2V/DIV VOUT 1V/DIV Start-Up from Shutdown VOUT 1V/DIV IL 500mA/DIV 3564 G15 IL 1A/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0mA 400μs/DIV 3564 G16 VIN = 3.6V VOUT = 1.8V ILOAD = 1.25A 400μs/DIV 3564 G17 Load Step VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV VIN = 3.6V 20μs/DIV VOUT = 1.8V ILOAD = 50mA TO 1.25A 3564 G19 Load Step VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV VIN = 3.6V 20μs/DIV VOUT = 1.8V ILOAD = 0.25A TO 1.25A 3564 G20 3564f LTC3564 PI FU CTIO S VFB (Pin 1/Pin 4) : Feedback Pin. Receives the feedback voltage from an external resistive divider across the output. GND (Pin 2/NA): Ground Pin. VIN (Pin 3/Pin 3): Main Supply Pin. Must be closely decoupled to GND, Pin 2, with a 10μF or greater ceramic capacitor. SW (Pin 4/Pin 1): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. FU CTIO AL DIAGRA SLOPE COMP OSC OSC FREQ SHIFT VFB 0.6V EA VIN RUN 0.6V REF SHUTDOWN – IRCMP + W – + U U U U U (S5/DCB) RUN (Pin 5/Pin 6): Run Control Input. Forcing this pin above 1.5V enables the part. Forcing this pin below 0.3V shuts down the device. In shutdown, all functions are disabled drawing 40%. However, the LTC3564 uses a patented scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles. 3564f LTC3564 APPLICATIO S I FOR ATIO L VIN CIN VIN SW CF LTC3564 R2 RUN GND VFB R1 3564 F01 Figure 1. LTC3564 General Schematic The basic LTC3564 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Inductor Selection For most applications, the value of the inductor will fall in the range of 0.47μH to 2.2μH. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is ΔIL = 500mA (40% of 1.25A). ΔIL = ⎛ ⎞ V 1 VOUT ⎜ 1− OUT ⎟ VIN ⎠ ( f )(L ) ⎝ (1) Table 1. Representative Surface Mount Inductors MANUFATURER Toko Sumida PART NUMBER A915AY-1R1M-DC53LC 1070AS-1R0N-DB3020C CDRH4D18C/LD-1R1 CDRH3D14-1R2 CR5D11-1R0 CDRH2D18/HP-2R2 FDK Coilcraft Vishay MIPW3226D0R9M LPO6610-122ML LPS4018-222ML IHLP1616ABERR47M01 IHLP1616ABER1R0M01 VALUE (μH) 1.1 1 1.1 1.2 1 2.2 0.9 1.2 2.2 0.47 1 MAX DC CURRENT (A) 3.25 1.9 2.1 2.2 2.2 1.6 1.4 2.1 2.5 5 4 DCR (mΩ) 16 47 24 36 40 48 70 80 70 20 45 HEIGHT (mm) 3 2 2 1.5 1.2 2 1 1 1.8 1.2 1.2 U VOUT COUT W U U The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 1.5A rated inductor should be enough for most applications (1.25A + 250mA). For better efficiency, choose a low DC-resistance inductor. The inductor value also has an effect on Burst Mode operation. The transition to low current operation begins when the inductor current peaks fall to approximately 300mA. Lower inductor values (higher ΔIL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase. Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3564 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3564 applications. 3564f 9 LTC3564 APPLICATIO S I FOR ATIO CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: [VOUT (VIN − VOUT )]1/ 2 CIN required IRMS ≅ IOMAX VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ΔVOUT is determined by: ⎛ 1⎞ ΔVOUT ≅ ΔIL ⎜ ESR + ⎟ ⎝ 8fC OUT ⎠ where f = operating frequency, COUT = output capacitance and ΔIL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ΔIL increases with input voltage. 10 U Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum capacitors. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3564’s control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. 3564f W U U LTC3564 APPLICATIO S I FOR ATIO Output Voltage Programming ⎛ R2 ⎞ VOUT = 0.6 V ⎜ 1 + ⎟ ⎝ R1⎠ POWER LOSS (W) In the adjustable version, the output voltage is set by a resistive divider according to the following formula: (2) The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 2. 0.6V ≤ VOUT ≤ 5.5V R2 VFB LTC3564 GND 3564 F02 R1 Figure 2. Setting the LTC3564 Output Voltage Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3564 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 3. U 1 VIN = 3.6V 0.1 0.01 0.001 VOUT = 1.2V VOUT = 1.5V VOUT = 1.8V 1 10 100 1000 LOAD CURRENT (A) 10000 3564 F03 W U U 0.0001 0.1 Figure 3. Power Lost vs Load Current 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC) 3564f 11 LTC3564 APPLICATIO S I FOR ATIO The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses which generally account for less than 2% total additional loss. Thermal Considerations In most applications the LTC3564 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3564 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3564 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(θJA) where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: T J = TA + TR where TA is the ambient temperature. As an example, consider the LTC3564 in dropout at an input voltage of 2.7V, a load current of 1.2A and an ambient temperature of 70°C. From the typical perfor- 12 U mance graph of switch resistance, the RDS(ON) of the P-channel switch at 70°C is approximately ~0.2Ω. Therefore, power dissipated by the part is: PD = ILOAD2 • RDS(ON) = 288mW For the SOT-23 package, the θJA is 215°C/ W. Thus, the junction temperature of the regulator is: TJ = 70°C + (0.288)(215) = 131.9°C which is above the maximum junction temperature of 125°C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (ΔILOAD • ESR), where ESR is the effective series resistance of COUT. ΔILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. A second, more severe transient is caused by switching in loads with large (>1μF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 • CLOAD). Thus, a 10μF capacitor charging to 3.3V would require a 250μs rise time, limiting the charging current to about 130mA. 3564f W U U LTC3564 APPLICATIO S I FOR ATIO R1 VIN Figure 4a. LTC3564 TSOT-23 Layout Diagram VOUT + – VIN Figure 4b. LTC3564 DFN Layout Diagram U 1 VFB GND 4 L1 CFWD RUN 5 R2 LTC3564 2 CIN 3 VIN SW W U U – COUT + VOUT 3564 F04a BOLD LINES INDICATE HIGH CURRENT PATH CFWD R2 L1 COUT CIN 3 VIN VFB 4 1 SW RUN 6 LTC3564 2 PGND SGND 5 R1 3564 F04b BOLD LINES INDICATE HIGH CURRENT PATH 3564f 13 LTC3564 APPLICATIO S I FOR ATIO VIA TO GND VIN VIA TO VOUT R2 CFWD Figure 5a. LTC3564 TSOT-23 Suggested Layout VOUT COUT SW PGND CIN R2 VIN VIA TO VOUT 3564 F05b Figure 5b. LTC3564 DFN Suggested Layout 14 U R1 VIA TO VIN PIN 1 LTC3564 SW L1 VOUT CIN GND 3564 F05a W U U COUT L1 VIA TO VIN 1 2 3 7 6 5 SGND R1 4 VFB CFWD 3564f LTC3564 APPLICATIO S I FOR ATIO PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3564. These items are also illustrated graphically in Figures 4 and 5. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. 2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground. 3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 4. Keep the switching node, SW, away from the sensitive VFB node. 5. Keep the (–) plates of CIN and COUT as close as possible. Design Example As a design example, assume the LTC3564 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 1.25A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low EFFICIENCY (%) VIN 2.7V TO 4.2V 3 CIN** 22μF CER 5 VIN SW 4 1.1μH* 22pF LTC3564 RUN GND 2 VFB 1 1M 316k 3564 F06a *TOKO A915AY-1R1M (D53LC SERIES) ** TAIYO YUDEN JMK316BJ226ML Figure 6a. Typical Application U and high load currents is important. Output voltage is 2.5V. With this information we can calculate L using equation (1), W U U L= ⎛V⎞ 1 VOUT ⎜ 1 − OUT ⎟ ( f)(ΔIL ) ⎝ VIN ⎠ (3) Substituting VOUT = 2.5V, VIN = 4.2V, ΔIL = 500mA and f = 2.25MHz in equation (3) gives: L= 2.5V ⎛ 2 . 5V ⎞ 1− = 0 . 9μ H ⎝ ⎠ 2.25MHz(500mA) ⎜ 4 . 2V ⎟ A 1μH or 1.1μH inductor works well for this application. For best efficiency choose a 1.5A or greater inductor with less than 0.1Ω series resistance. CIN will require an RMS current rating of at least 0.6A ≅ ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.125Ω. In most cases, a ceramic capacitor will satisfy this requirement. For the feedback resistors, choose R1 = 316k. R2 can then be calculated from equation (2) to be: ⎛V ⎞ R2 = ⎜ OUT − 1⎟ R1 = 1000k ⎝ 0.6 ⎠ Figure 6 shows the complete circuit along with its efficiency curve. 100 95 90 85 80 75 70 65 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1000 3564 F06b VOUT 2.5V COUT** 22μF CER VOUT = 2.5V 60 1 0.1 10 100 OUTPUT CURRENT (mA) Figure 6b. Efficiency vs Output Current 3564f 15 LTC3564 TYPICAL APPLICATIO S Single Li-Ion 1.8V/1.25A Regulator for High Efficiency and Small Footprint VIN 2.7V TO 5.5V 100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10000 VOUT = 1.8V 0 1 10 100 1000 0.1 OUTPUT CURRENT (mA) 16 U 4 CIN** 10μF CER 1 VIN SW 3 1μH* 22pF VOUT 1.8V COUT† 22μF CER LTC3564 RUN GND 2 VFB 5 806k 402k *MURATA LQH32CN2R2M33 ** TAIYO YUDEN JMK316BJ106ML † TAIYO YUDEN JMK316BJ226ML-BR 3564 TA02a VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV VIN = 3.6V 20μs/DIV VOUT = 1.8V ILOAD = 100mA TO 1.25A 3564 TA02c 3564 TA02b 3564f LTC3564 TYPICAL APPLICATIO S Single Li-Ion 1.5V/1.25A Regulator for High Efficiency and Low Profile,