LTC3619BEDD-TRPBF

FeaTures n n n n n n n n n n n n n n Programmable Average Input Current Limit: ±5% Accuracy Dual Step-Down Outputs: Up to 96% Efficiency Low Noise Pulse-Skipping Operation at Light Loads Input Voltage Range: 2.5V to 5.5V Output Voltage Range: 0.6V to 5V 2.25MHz Constant-Frequency Operation Power Good Output Voltage Monitor for Each Channel Low Dropout Operation: 100% Duty Cycle Independent Internal Soft-Start for Each Channel Current Mode Operation for Excellent Line and Load Transient Response ±2% Output Voltage Accuracy Short-Circuit Protected Shutdown Current ≤ 1μA Available in Small Thermally Enhanced 10-Lead MS and 3mm × 3mm DFN Packages High Peak Load Current Applications USB Powered Devices Supercapacitor Charging Radio Transmitters and Other Handheld Devices LTC3619B 400mA/800mA Synchronous Step-Down DC/DC with Average Input Current Limit DescripTion The LTC®3619B is a dual monolithic synchronous buck regulator using a constant frequency current mode architecture. The input supply voltage range is 2.5V to 5.5V, making it ideal for Li-Ion and USB powered applications. 100% duty cycle capability provides low dropout operation, extending the run time in battery-operated systems. Low output voltages are supported with the 0.6V feedback reference voltage. Channel 1 and channel 2 can supply 400mA and 800mA output current, respectively. The LTC3619B’s programmable average input current limit is ideal for USB applications and for point-of-load power supplies because the LTC3619B’s limited input current will still allow its output to deliver high peak load currents without collapsing the input supply. When the sum of both channels’ currents exceeds the input current limit, channel 2 is current limited while channel 1 remains regulated. The operating frequency is internally set at 2.25MHz allowing the use of small surface mount inductors. Internal soft-start reduces in-rush current during start-up. The LTC3619B is available in small MSOP and 3mm × 3mm DFN packages. The LTC3619B is also available in a low quiescent current, high efficiency Burst Mode® version, LTC3619. applicaTions n n n n L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S.Patents, including 5481178, 6127815, 6304066, 6498466, 6580258, 6611131. Typical applicaTion Dual Monolithic Buck Regulator in 10-Lead 3mm × 3mm DFN VIN 3.4V TO 5.5V 10µF RUN2 VIN RUN1 PGOOD2 PGOOD1 VOUT2 3.4V AT 800mA 1.5µH LTC3619B SW2 1190k 2.2mF 2 SuperCap 255k SW1 3.3µH 22pF VOUT1 1.8V AT 400mA VOUT 200mV/DIV VIN AC-COUPLED 1V/DIV IOUT 500mA/DIV IIN 500mA/DIV 1ms/DIV 3619B TA01 3619B TA01b GSM Pulse Load + VFB2 VFB1 RLIM GND 255k 511k 10µF 1000pF 116k ILIM = 475mA VIN = 5V, 500mA COMPLIANT ILOAD = 0A to 2.2A, CHANNEL 1 UNLOADED 3619bfa  LTC3619B absoluTe MaxiMuM raTings (Note 1) Input Supply Voltage (VIN) ............................. –0.3 to 6V VFB1, VFB2 ........................................ –0.3V to VIN + 0.3V RUN1, RUN2, RLIM .......................... –0.3V to VIN + 0.3V SW1, SW2........................................ –0.3V to VIN + 0.3V PGOOD1, PGOOD2 ........................... –0.3V to VIN + 0.3V P-channel SW Source Current (DC) (Note 2) Channel 1 ........................................................ 600mA Channel 2 ................................................................1A N-channel SW Source Current (DC) (Note 2) Channel 1 ........................................................ 600mA Channel 2 ................................................................1A Peak SW Source and Sink Current (Note 2) Channel 1 ........................................................ 900mA Channel 2 ............................................................. 2.7A Operating Junction Temperature Range (Notes 3, 6, 8) ........................................–40 to 125°C Storage Temperature Range .................. –65°C to 125°C Lead Temperature (Soldering, 10 sec) MSOP Package ................................................. 300°C Reflow Peak Body Temperature ............................ 260°C pin conFiguraTion TOP VIEW VFB1 RUN1 RLIM PGOOD1 SW1 1 2 3 4 5 11 GND 10 VFB2 9 RUN2 8 PGOOD2 7 SW2 6 VIN TOP VIEW VFB1 RUN1 RLIM PGOOD1 SW1 1 2 3 4 5 11 GND 10 9 8 7 6 VFB2 RUN2 PGOOD2 SW2 VIN DD PACKAGE 10-LEAD (3mm 3mm) PLASTIC DFN TJMAX = 125°C, θJA = 40°C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB MSE PACKAGE 10-LEAD PLASTIC MSOP TJMAX = 125°C, θJA = 45°C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB orDer inForMaTion LEAD FREE FINISH LTC3619BEDD#PBF LTC3619BIDD#PBF LTC3619BEMSE#PBF LTC3619BIMSE#PBF TAPE AND REEL LTC3619BEDD#TRPBF LTC3619BIDD#TRPBF LTC3619BEMSE#TRPBF LTC3619BIMSE#TRPBF PART MARKING* LFFH LFFH LTFFJ LTFFJ PACKAGE DESCRIPTION 10-Lead (3mm × 3mm) Plastic DFN 10-Lead (3mm × 3mm) Plastic DFN 10-Lead Plastic MSOP 10-Lead Plastic MSOP TEMPERATURE RANGE –40°C to 125°C –40°C to 125°C –40°C to 125°C –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25°C (Note 3) SYMBOL VIN VUV PARAMETER VIN Operating Voltage Range VIN Undervoltage Lockout VIN Low to High CONDITIONS l l elecTrical characTerisTics MIN 2.5 TYP 2.1 MAX 5.5 2.5 UNITS V V 3619bfa  LTC3619B elecTrical characTerisTics SYMBOL IFB VFBREG ΔVLINEREG ΔVLOADREG IS fOSC ILIM(PEAK) IINLIM RDS(ON) PARAMETER Feedback Pin Input Current Feedback Voltage (Channels 1, 2) VFB Line Regulation VFB Load Regulation (Channel 1) VFB Load Regulation (Channel 2) Supply Current Active Mode (Note 4) Shutdown Oscillator Frequency Peak Switch Current Limit Channel 1 (400mA) Channel 2 (800mA) Input Average Current Limit LTC3619BE, –40°C < TJ < 85°C (Note 7) LTC3619BI, –40°C < TJ < 125°C (Note 7) VIN = 2.5V to 5.5V (Note 7) ILOAD = 0mA to 400mA (Note 7) ILOAD = 0mA to 800mA (Note 7) VFB1 = VFB2 = 0.95 × VFBREG VRUN1 = VRUN2 = 0V, VIN = 5.5V VFB = VFBREG VIN = 5V, VFB < VFBREG , Duty Cycle 100pF at the RLIM pin. Each application input current limit will call for different CLIM value to optimize its response time. Using a large CLIM capacitor requires longer time for the RLIM pin voltage to charge. For example, consider the application 500mA input current limit, 5V input and 1A, 2.5V output with a 50% duty cycle. When an instantaneous 1A output pulse is applied, the current out of the RLIM pin becomes 1A/55k = 18.2µA during the 50% on-time or 9.1µA full duty cycle. With a , CLIM capacitor of 1µF RLIM of 116k, and using I = CdV/dt, it will take 110ms for CLIM to charge from 0V to 1V. This is the time after which the LTC3619B will start input current 3619bfa  LTC3619B operaTion limiting. Any current within this time must be considered in each application to determine if it is tolerable. Figure 1a shows VIN (IIN) current below input current limit . with a CLIM capacitor of 0.1µF Channel 1 is unloaded to simplify calculations. When the load pulse is applied, under the specified condition, ILIM current is 1.1A/55k • 0.66 = 13.2µA, where 0.66 is the duty cycle. It will take a little more than 7.5ms to charge the CLIM capacitor from 0V to 1V, after which the LTC3619B begins to limit input current. The IIN current is not limited during this 7.5ms time and is more than 725mA. This current transient may cause the input supply to temporarily droop if the supply current compliance is exceeded, but recovers after the input current limit engages. The output will continue to deliver the required current load while the output voltage droops to allow the input voltage to remain regulated during input current limit. For applications with short load pulse duration, a smaller CLIM capacitor may be the better choice as in the example shown in Figure 1b. Channel 1 is unloaded for simplification. In this example, a 577µs, 0A to 2A output pulse VOUT 2V/DIV IIN 500mA/DIV VRLIM 1V/DIV IL 1A/DIV 50ms/DIV 3619B F01a is applied once every 4.7ms. A CLIM capacitor of 2.2nF requires 92µs for VRLIM to charge from 0V to 1V. During this 92µs, the input current limit is not yet engaged and the output must deliver the required current load. This may cause the input voltage to droop if the current compliance is exceeded. Depending on how long this time is, the VIN supply decoupling capacitor can provide some of this current before VIN droops too much. In applications with a bigger VIN supply decoupling capacitor and where VIN supply is allow to droop closer to dropout, the CLIM capacitor can be increased slightly. This will delay the start of input current limit and artificially regulated VOUT before input current limit is engaged. In this case, within the 577µs load pulse, the VOUT voltage will stay artificially regulated for 92µs out of the total 577µs before the input current limit activates. This approach may be used if a faster recovery on the output is desired. Selecting a very small CLIM will speed up response time but it can put the device within threshold of interfering with normal operation and input current limit in every few switching cycles. This may be undesirable in terms of noise. Use 2πRC >> 100/clock frequency (2.25MHz) as VOUT 200mV/DIV VIN AC-COUPLED 1V/DIV IOUT 500mA/DIV IIN 500mA/DIV 1ms/DIV 3619B F01b VIN = 5V, 500mA COMPLIANT RLIM = 116k, CLIM = 0.1µF , ILOAD = 0A to 1.1A, COUT = 2.2mF VOUT = 3.3V ILIM = 475mA, CHANNEL 1 NOT LOADED VIN = 5V, 500mA COMPLIANT RLIM = 116k, CLIM = 2200pF , ILOAD = 0A to 2A, COUT = 2.2mF VOUT = 3.3V ILIM = 475mA, CHANNEL 1 NOT LOADED Figure 1a. Input Current Limit Within 100ms Load Pulses Figure 1b. Input Current Limit Within 577µs, 2A Repeating Load Pulses 3619bfa 0 LTC3619B applicaTions inForMaTion A general LTC3619B application circuit is shown in Figure 2. External component selection is driven by the load requirement, and begins with the selection of the inductor L. Once the inductor is chosen, CIN and COUT can be selected. Inductor Selection Although the inductor does not influence the operating frequency, the inductor value has a direct effect on ripple current. The inductor ripple current ΔIL decreases with higher inductance and increases with higher VIN or VOUT : ΔIL = VOUT  VOUT  • 1− fO • L  VIN    (1) Inductor Core Selection Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price versus size requirements, and any radiated field/EMI requirements, than on what the LTC3619B requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3619B applications. Table 1. Representative Surface Mount Inductors MANUFACTURER Coilcraft PART NUMBER LPS4012-152ML LPS4012-222ML LPS4012-332ML LPS4012-472ML LPS4018-222ML LPS4018-332ML LPS4018-472ML FDKMIPF2520D FDKMIPF2520D FDKMIPF2520D ELT5KT4R7M CDRH2D18/LD CDH38D11SNP3R3M CDH38D11SNP2R2M MAX DC VALUE CURRENT 1.5µH 2.2µH 3.3µH 4.7µH 2.2µH 3.3µH 4.7µH 4.7µH 3.3µH 2.2µH 4.7µH 4.7µH 3.3μH 2.2μH 2.2µH 2.2µH 3.3µH 2.2µH 4.7µH 4.7µH 3.3µH 2.2µH 2.2µH 3.3μH 2.2µH 2200mA 1750mA 1450mA 1450mA 2300mA 2000mA 1800mA 1100mA 1200mA 1300mA 450mA 950mA 630mA 1560mA 1900mA 510mA 530mA 410mA 1100mA 750mA 700mA 870mA 1000mA 1500mA 1700mA 2300mA DCR 0.070Ω 0.100Ω 0.100Ω 0.170Ω 0.070Ω 0.080Ω 0.125Ω 0.11Ω 0.1Ω 0.08Ω 0.2Ω 0.2Ω 0.086Ω 0.115Ω 0.082Ω 0.13Ω 0.33Ω 0.27Ω 0.1Ω 0.19Ω 0.28Ω 0.17Ω 0.12Ω 0.076Ω 0.095Ω 0.059Ω HEIGHT 1.2mm 1.2mm 1.2mm 1.2mm 1.8mm 1.8mm 1.8mm 1mm 1mm 1mm 2mm 1.2mm 2mm 1.2mm 1.2mm 1.6mm 1.25mm 1.6mm 1mm 1mm 1mm 1mm 1mm 1.2mm 1.2mm 1.4mm Accepting larger values of ΔIL allows the use of low inductances, but results in higher output voltage ripple, greater core losses, and lower output current capability. A reasonable starting point for setting ripple current is 40% of the maximum output load current. So, for a 800mA regulator, ΔIL = 320mA (40% of 800mA). The inductor value will also have an effect on Burst Mode operation. The transition to low current operation begins when the peak inductor current falls below a level set by the internal burst clamp. Lower inductor values result in higher ripple current which causes the transition to occur at lower load currents. This causes a dip in efficiency in the upper range of low current operation. Furthermore, lower inductance values will cause the bursts to occur with increased frequency. VIN 2.5V TO 5.5V C1 RUN2 VIN RUN1 PGOOD2 PGOOD1 VOUT2 L2 CF2 LTC3619B SW2 SW1 L1 CF1 VOUT1 FDK Murata Panasonic Sumida LQH32CN4R7M23 4.7µH Taiyo Yuden CB2016T2R2M CB2012T2R2M CB2016T3R3M NR30102R2M NR30104R7M TDK VLF3010AT4R7MR70 VLF3010AT3R3MR87 VLF3010AT2R2M1R0 VLF4012AT-2R2 M1R5 VLF5012ST-3R3 M1R7 VLF5014ST-2R2 M2R3 COUT2 R4 R3 VFB2 RLIM CLIM VFB1 GND RLIM R1 R2 COUT1 3619B F02 Figure 2. LTC3619B General Schematic 3619bfa  LTC3619B applicaTions inForMaTion Input Capacitor (CIN) Selection In continuous mode, the input current of the converter is a square wave with a duty cycle of approximately VOUT / VIN . To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: IRMS ≈ IMAX VOUT (VIN − VOUT ) VIN capacitor types include Sanyo POSCAP Kemet T510 and , T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3619B control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. For more information, see Application Note 88. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Setting the Output Voltage The LTC3619B regulates the VFB1 and VFB2 pins to 0.6V during regulation. Thus, the output voltage is set by a resistive divider, Figure 2, according to the following formula:  R2  VOUT = 0.6V  1+   R1 (2) Where the maximum average output current IMAX equals the peak current minus half the peak-to-peak ripple current, IMAX = ILIM – ΔIL /2. This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case is commonly used to design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet the size or height requirements of the design. An additional 0.1µF to 1µF ceramic capacitor is also recommended on VIN for high frequency decoupling when not using an all-ceramic capacitor solution. Output Capacitor (COUT) Selection The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ΔVOUT is determined by:  1 ΔVOUT ≈ ΔIL  ESR + 8fOCOUT    where fO = operating frequency, COUT = output capacitance and ΔIL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ΔIL increases with input voltage. If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other Keeping the current small (< 10µA) in these resistors maximizes efficiency, but making it too small may allow stray capacitance to cause noise problems or reduce the phase margin of the error amp loop. To improve the frequency response of the main control loop, a feedback capacitor (CF) may also be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. 3619bfa  LTC3619B applicaTions inForMaTion Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ΔILOAD • ESR, where ESR is the effective series resistance of COUT. ΔILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second order overshoot/DC ratio cannot be used to determine the phase margin. In addition, feedback capacitors (CF1 and CF2) can be added to improve the high frequency response, as shown in Figure 2. Capacitor CF provides phase lead by creating a high frequency zero with R2 which improves the phase margin. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Application Note 76. In some applications, a more severe transient can be caused by switching in loads with large (>1µF) input capacitors. The discharged input capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot Swap™ controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection, and soft-starting. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: % Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc., are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four sources usually account for the losses in LTC3619B circuits: 1) VIN quiescent current, 2) switching losses, 3) I2R losses, 4) other system losses. 1. The VIN current is the DC supply current given in the Electrical Characteristics which excludes MOSFET driver and control currents. VIN current results in a small (