LTC3783

LTC3783 PWM LED Driver and Boost, Flyback and SEPIC Controller DESCRIPTIO The LTC®3783 is a current mode LED driver and boost, flyback and SEPIC controller that drives both an N-channel power MOSFET and an N-channel load PWM switch. When using an external load switch, the PWMIN input not only drives PWMOUT, but also enables controller GATE switching and error amplifier operation, allowing the controller to store load current information while PWMIN is low. This feature (patent pending) provides extremely fast, true PWM load switching with no transient overvoltage or undervoltage issues; LED dimming ratios of 3000:1 can be achieved digitally, avoiding the color shift normally associated with LED current dimming. The FBP pin allows analog dimming of load current, further increasing the effective dimming ratio by 100:1 over PWM alone. In applications where output load current must be returned to VIN, optional constant current/constant voltage regulation controls either output (or input) current or output voltage and provides a limit for the other. ILIM provides a 10:1 analog dimming ratio. For low- to medium-power applications, No RSENSE mode can utilize the power MOSFET’s on-resistance to eliminate the current-sense resistor, thereby maximizing efficiency. The IC’s operating frequency can be set with an external resistor over a 20kHz to 1MHz range and can be synchronized to an external clock using the SYNC pin. The LTC3783 is available in the 16-lead DFN and TSSOP packages. FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ True Color PWMTM Delivers Constant Color with 3000:1 Dimming Ratio Fully Integrated Load FET Driver for PWM Dimming Control of High Power LEDs 100:1 Dimming from Analog Inputs Wide FB Voltage Range: 0V to 1.23V Constant Current or Constant Voltage Regulation Low Shutdown Current: IQ = 20µA 1% 1.23V Internal Voltage Reference 2% RUN Pin Threshold with 100mV Hysteresis Programmable Operating Frequency (20kHz to 1MHz) with One External Resistor Synchronizable to an External Clock Up to 1.3fOSC Internal 7V Low Dropout Voltage Regulator Programmable Output Overvoltage Protection Programmable Soft-Start Can be Used in a No RSENSETM Mode for VDS < 36V 16-Lead DFN and TSSOP Packages APPLICATIO S ■ ■ ■ ■ ■ High Voltage LED Arrays Telecom Power Supplies 42V Automotive Systems 24V Industrial Controls IP Phone Power Supplies , LTC and LT are registered trademarks of Linear Technology Corporation. True Color PWM and No RSENSE are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Patent pending. TYPICAL APPLICATIO VIN 6V TO 16V (< TOTAL VF OF LEDs) 1M LTC3783 RUN VIN PWMIN OV/FB ITH PWMOUT SS ILIM GATE VREF FBP SENSE FBN INTVCC GND FREQ SYNC 350mA PWM LED Boost Application 10µF ×2 2.2µH ZETEX ZLL51000 VOUT 2.5V), connect this pin to the negative side of the current-regulating resistor. Nominal voltage for this pin in regulation is either VFBP or (VFBP – 100mV) for VILIM = 1.23V, depending on operational mode (voltage or constant current/constant voltage) set by the voltage at VFBP. FBP (Pin 2): Error Amplifier Noninverting Input/Positive Current Sense Pin. This pin voltage determines the control loop’s feedback mode (voltage or constant current/constant voltage), the threshold of which is approximately 2V. In voltage mode (VFBP ≤ VREF), this pin represents the desired voltage which the regulated loop will cause FBN to follow. In constant current/constant voltage mode (VFBP > 2.5V), connect this pin to the positive side of the load current-sensing resistor. The acceptable input ranges for this pin are 0V to 1.23V (voltage mode) and 2.5V to 36V (constant current/constant voltage mode). ILIM(Pin 3): Current Limit Pin. Sets current sense resistor offset voltage (VFBP – VFBN) in constant current mode regulation (i.e., when VFBP > 2.5V). Offset voltage is 100mV when VILIM = 1.23V and decreases proportionally with VILIM. Nominal voltage range for this pin is 0.1V to 1.23V. VREF (Pin 4): Reference Voltage Pin. Provides a buffered version of the internal bandgap voltage, which can be connected to FBP either directly or with attenuation. Nominal voltage for this pin is 1.23V. This pin should never be bypassed by a capacitor to GND. Instead, a 10k resistor to GND should be used to lower pin impedance in noisy systems. FREQ (Pin 5): A resistor from the FREQ pin to ground programs the operating frequency of the chip. The nominal voltage at the FREQ pin is 0.615V. SYNC (Pin 6): This input allows for synchronizing the operating frequency to an external clock and has an internal 100k pull-down resistor. PWMIN (Pin 7): PWM Gate Driver Input. Internal 100k pull-up resistor. While PWMIN is low, PWMOUT is low, GATE stops switching and the external ITH network is disconnected, saving the ITH state. PWMOUT (Pin 8): PWM Gate Driver Output. Used for constant current dimming (LED load) or for output disconnect (step-up power supply). GATE (Pin 9): Main Gate Driver Output for the Boost Converter. INTVCC (Pin 10): Internal 7V Regulator Output. The main and PWM gate drivers and control circuits are powered from this voltage. Decouple this pin locally to the IC ground with a minimum of 4.7µF low ESR ceramic capacitor. VIN (Pin 11): Main Supply Pin. Must be closely decoupled to ground. SENSE (Pin 12): Current Sense Input for the Control Loop. Connect this pin to the drain of the main power MOSFET for VDS sensing and highest efficiency for VSENSE ≤ 36V. Alternatively, the SENSE pin may be connected to a resistor in the source of the main power MOSFET. Internal leading-edge blanking is provided for both sensing methods. SS (Pin 13): Soft-Start Pin. Provides a 50µA pull-up current, enabled and reset by RUN, which charges an optional external capacitor. This voltage ramp translates into a corresponding current limit ramp through the main MOSFET. OV/FB (Pin 14): Overvoltage Pin/Voltage Feedback Pin. In voltage mode (VFBP ≤ VREF), this input, connected to VOUT through a resistor network, sets the output voltage at which GATE switching is disabled in order to prevent an overvoltage situation. Nominal threshold voltage for the OV pin is 1.32V (VREF + 7%) with 20mV hysteresis. In current/voltage mode (VFBP > 2.5V), this pin senses VOUT through a resistor divider and brings the loop into voltage regulation such that pin voltage approaches VREF = 1.23V, provided the loop is not regulating the load current (e.g., [VFBP – VFBN] < 100mV for ILIM = 1.23V). U U U 3783f 7 LTC3783 PI FU CTIO S ITH (Pin 15): Error Amplifier Output/Compensation Pin. The current comparator input threshold increases with this control voltage, which is the output of the gm type error amplifier. Nominal voltage range for this pin is 0V to 1.40V. RUN (Pin 16): The RUN pin provides the user with an accurate means for sensing the input voltage and programming the start-up threshold for the converter. The falling RUN pin threshold is nominally 1.248V and the comparator has 100mV hysteresis for noise immunity. When the RUN pin is grounded, the IC is shut down and the VIN supply current is kept to a low value (20µA typ). Exposed Pad (Pin 17): Ground Pin. Solder to PCB ground for electrical contact and rated thermal performance. BLOCK DIAGRA 5 FREQ V-TO-I 0.615V SYNC OSC CLK S Q R SS_RESET LOGIC GATE 9 6 1.9V ILIM FBP FBN OV/FB – + 3 2 1 14 A VREF 13 SS IMAX 15 ITH 7 PWMIN EN 10 INTVCC 8 W U U U VREF SLOPE COMP BIAS VREF GND 4 17 IVMODE OV/FB EA VREF + – OV TEMP SENSOR (165°C) OT ITRIP – – + + – + – SENSE 12 1S 0.2V + – SLEEP 0 50µA + – 0.15V V-TO-I PWMOUT 8 LDO VREF 2.23V + – UV BIAS AND START-UP + – RUN VREF VIN 16 11 3738 BD 3783f LTC3783 OPERATIO Main Control Loop The LTC3783 is a constant frequency, current mode controller for PWM LED as well as DC/DC boost, SEPIC and flyback converter applications. In constant current LED applications, the LTC3783 provides an especially wide PWM dimming range due to its unique switching scheme, which allows PWM pulse widths as short as several converter switching periods. For voltage feedback circuit operation (defined by VFBP ≤ 1.23V), please refer to the Block Diagram of the IC and the Typical Application on the first page of this data sheet. In normal operation with PWMIN high, the power MOSFET is turned on (GATE goes high) when the oscillator sets the PWM latch, and is turned off when the ITRIP current comparator resets the latch. Based on the error voltage represented by (VFBP – VFBN), the error amplifier output signal at the ITH pin sets the ITRIP current comparator input threshold. When the load current increases, a fall in the FBN voltage relative to the reference voltage at FBP causes the ITH pin to rise, causing the ITRIP current comparator to trip at a higher peak inductor current value. The average inductor current will therefore rise until it equals the load current, thereby maintaining output regulation. When PWMIN goes low, PWMOUT goes low, the ITH switch opens and GATE switching is disabled. Lowering PWMOUT and disabling GATE causes the output capacitor COUT to hold the output voltage constant in the absence of load current. Opening the ITH switch stores the correct load current value on the ITH capacitor CITH. As a result, when PWMIN goes high again, both ITH and VOUT are instantly at the appropriate levels. In voltage feedback operation, an overvoltage comparator, OV, senses when the OV/FB pin exceeds the reference voltage by 7% and provides a reset pulse to the main RS latch. Because this RS latch is reset-dominant, the power MOSFET is actively held off for the duration of an output overvoltage condition. U For constant current/constant voltage regulation operation (defined by VFBP > 2.5V), please refer to the Block Diagram of the IC and Figure 11. Loop operation is similar to the voltage feedback, except FBP and FBN now sense the voltage across sense resistor RL in series with the load. The ITH pin now represents the error from the desired differential set voltage, from 10mV to 100mV, for ILIM values of 0.123V to 1.23V. That is, with VILIM = 1.23V, the loop will regulate such that VFBP – VFBN = 100mV; lower values of ILIM attenuate the difference proportionally. PWMIN is still functional as above, but will only work properly if load current can be disconnected by the PWMOUT signal. In constant current/constant voltage operation, the OV/FB pin becomes a voltage feedback pin, which causes the loop to regulate such that VOV/FB = 1.23V, provided the above current-sense voltage is not reached. In this way, the loop regulates either voltage or current, whichever parameter hits its preset limit first. The nominal operating frequency of the LTC3783 is programmed using a resistor from the FREQ pin to ground and can be controlled over a 20kHz to 1MHz range. In addition, the internal oscillator can be synchronized to an external clock applied to the SYNC pin and can be locked to a frequency between 100% and 130% of its nominal value. When the SYNC pin is left open, it is pulled low by an internal 100k resistor. With no load, or an extremely light one, the controller will skip pulses in order to maintain regulation and prevent excessive output ripple. The RUN pin controls whether the IC is enabled or is in a low current shutdown state. A micropower 1.248V reference and RUN comparator allow the user to program the supply voltage at which the IC turns on and off (the RUN comparator has 100mV of hysteresis for noise immunity). With the RUN pin below 1.248V, the chip is off and the input supply current is typically only 20µA. 3783f 9 LTC3783 OPERATIO The SS pin provides a soft-start current to charge an external capacitor. Enabled by RUN, the soft-start current is 50µA, which creates a positive voltage ramp on V SS to which the internal ITH is limited, avoiding high peak currents on start-up. Once VSS reaches 1.23V, the full ITH range is established. The LTC3783 can be used either by sensing the voltage drop across the power MOSFET or by connecting the SENSE pin to a conventional shunt resistor in the source of the power MOSFET, as shown in the Typical Application on the first page of this data sheet. Sensing the voltage across the power MOSFET maximizes converter efficiency and minimizes the component count, but limits the output voltage to the maximum rating for this pin (36V). By connecting the SENSE pin to a resistor in the source of the power MOSFET, the user is able to program output voltages significantly greater than 36V, limited only by other components’ breakdown voltages. Externally Synchronized Operation When an external clock signal drives the SYNC pin at a rate faster than the chip’s internal oscillator, the oscillator will synchronize to it. When the oscillator’s internal logic circuitry detects a synchronizing signal on the SYNC pin, the internal oscillator ramp is terminated early and the slope compensation is increased by approximately 25%. As a result, in applications requiring synchronization, it is recommended that the nominal operating frequency of the IC be programmed to be about 80% of the external clock frequency. Attempting to synchronize to too high an external frequency (above 1.3fOSC) can result in inadequate slope compensation and possible subharmonic oscillation (or jitter). The external clock signal must exceed 2V for at least 25ns, and should have a maximum duty cycle of 80%, as shown in Figure 1. The MOSFET turn-on will synchronize to the rising edge of the external clock signal. Programming the Operating Frequency The choice of operating frequency and inductor value is a tradeoff between efficiency and component size. Low frequency operation improves efficiency by reducing RT (kΩ) 10 U 2V TO 7V MODE/ SYNC tMIN = 25ns 0.8T T T = 1/fO GATE D = 40% IL 3783 F01 Figure 1. MODE/SYNC Clock Input and Switching Waveforms for Synchronized Operation MOSFET and diode switching losses. However, lower frequency operation requires more inductance for a given amount of load current. The LTC3783 uses a constant frequency architecture that can be programmed over a 20kHz to 1MHz range with a single external resistor from the FREQ pin to ground, as shown in the application on the first page of this data sheet. The nominal voltage on the FREQ pin is 0.615V, and the current that flows out of the FREQ pin is used to charge and discharge an internal oscillator capacitor. The oscillator frequency is trimmed to 300kHz with RT = 20k. A graph for selecting the value of RT for a given operating frequency is shown in Figure 2. 1000 100 10 1 1 10 100 1000 FREQUENCY (kHz) 10000 3783 G09 Figure 2. Timing Resistor (RT) Value 3783f LTC3783 OPERATIO INTVCC Regulator Bypassing and Operation An internal, P-channel low dropout voltage regulator produces the 7V supply which powers the gate drivers and logic circuitry within the LTC3783 as shown in Figure 3. The INTVCC regulator can supply up to 50mA and must be bypassed to ground immediately adjacent to the IC pins with a minimum of 4.7µF low ESR or ceramic capacitor. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate driver. For input voltages that don’t exceed 8V (the absolute maximum rating for INTVCC is 9V), the internal low dropout regulator in the LTC3783 is redundant and the INTVCC pin can be shorted directly to the VIN pin. With the INTVCC pin shorted to VIN, however, the divider that programs the regulated INTVCC voltage will draw 15µA from the input supply, even in shutdown mode. For applications that require the lowest shutdown mode input supply current, do not connect the INTVCC pin to VIN. Regardless of whether the INTVCC pin is shorted to VIN or not, it is always necessary to have the driver circuitry bypassed with a 4.7µF low ESR ceramic capacitor to ground immediately adjacent to the INTVCC and GND pins. 1.230V R2 Figure 3. Bypassing the LDO Regulator and Gate Driver Supply + – U In an actual application, most of the IC supply current is used to drive the gate capacitance of the power MOSFET. As a result, high input voltage applications in which a large power MOSFET is being driven at high frequencies can cause the LTC3783 to exceed its maximum junction temperature rating. The junction temperature can be estimated using the following equations: IQ(TOT) = IQ + f • QG PIC = VIN • (IQ + f • QG) TJ = TA + PIC • θJA The total quiescent current IQ(TOT) consists of the static supply current (IQ) and the current required to charge and discharge the gate of the power MOSFET. The 16-lead FE package has a thermal resistance of θJA = 38°C/W and the DHD package has an θJA = 43°C/W As an example, consider a power supply with VIN = 12V and VOUT = 25V at IOUT = 1A. The switching frequency is 300kHz, and the maximum ambient temperature is 70°C. The power MOSFET chosen is the Si7884DP, which has a maximum RDS(ON) of 10mΩ (at room temperature) and a VIN INPUT SUPPLY 6V TO 36V P-CH CIN R1 7V INTVCC CVCC 4.7µF X5R M1 LOGIC DRIVER GATE 6V-RATED POWER MOSFET GND 3783 F03 GND PLACE AS CLOSE AS POSSIBLE TO DEVICE PINS 3783f 11 LTC3783 OPERATIO maximum total gate charge of 35nC (the temperature coefficient of the gate charge is low). IQ(TOT) = 1.2mA + 35nC • 300kHz = 12mA PIC = 12V • 12mA = 144mW TJ = 70°C + 110°C/W • 144mW = 86°C This demonstrates how significant the gate charge current can be when compared to the static quiescent current in the IC. To prevent the maximum junction temperature from being exceeded, the input supply current must be checked when operating in a continuous mode at high VIN. A tradeoff between the operating frequency and the size of the power MOSFET may need to be made in order to maintain a reliable IC junction temperature. Prior to lowering the operating frequency, however, be sure to check with the power MOSFET manufacturers for the latest low QG, low RDS(ON) devices. Power MOSFET manufacturing technologies are continually improving, with newer and betterperforming devices being introduced almost monthly. Output Voltage Programming In constant voltage mode, in order to regulate the output voltage, the output voltage is set by a resistor divider according to the following formula: R2 ⎞ ⎛ VOUT = VFBP • ⎜ 1+ ⎝ ⎠ R1⎟ where 0 ≤ VFBP ≤ 1.23V. The external resistor divider is connected to the output as shown in Figure 4, allowing remote voltage sensing. The resistors R1 and R2 are typically chosen so that the error caused by the 500nA input bias current flowing out of the FBN pin during normal operation is less than 1%, which translates to a maximum R1 value of about 25k at VFBP = 1.23V. For lower FBP voltages, R1 must be reduced accordingly to maintain accuracy, e.g., R1 < 2k for 1% accuracy when VFBP = 100mV. More accuracy can be achieved with lower resistances, at the expense of increased dissipation and decreased light load efficiency. 12 U A similar analysis applies to the VFBP resistive divider, if one is used: VFBP = VREF • R3 R3 + R4 where R3 is subject to a similar 500nA bias current. LTC3783 RUN VIN PWMIN OV/FB ITH PWMOUT SS ILIM GATE VREF FBP SENSE FBN INTVCC GND FREQ SYNC VIN 3V TO 36V VOUT R2 R4 R3 R1 GND 3783 F04 Figure 4. LTC3783 Boost Application Programming Turn-On and Turn-Off Thresholds with the RUN Pin The LTC3783 contains an independent, micropower voltage reference and comparator detection circuit that remains active even when the device is shut down, as shown in Figure 5. This allows users to accurately program an input voltage at which the converter will turn on and off. The falling threshold on the RUN pin is equal to the internal reference voltage of 1.248V. The comparator has 100mV of hysteresis to increase noise immunity. The turn-on and turn-off input voltage thresholds are programed using a resistor divider according to the following formulas: ⎛ R2 ⎞ VIN(OFF) = 1.248 V • ⎜ 1 + ⎟ ⎝ R1⎠ ⎛ R2 ⎞ VIN(ON) = 1.348 V • ⎜ 1 + ⎟ ⎝ R1⎠ The resistor R1 is typically chosen to be less than 1M. 3783f LTC3783 OPERATIO EXTERNAL LOGIC CONTROL Figure 5b. On/Off Control Using External Logic For applications where the RUN pin is only to be used as a logic input, the user should be aware of the 7V Absolute Maximum Rating for this pin! The RUN pin can be connected to the input voltage through an external 1M resistor, as shown in Figure 5c, for “always on” operation. Soft-Start Capacitor Selection For proper soft-start operation, the LTC3783 should have a sufficiently large soft-start capacitor, CSS, attached to the SS pin. The minimum soft-start capacitor size can be estimated on the basis of output voltage, capacitor size and load current. In addition, PWM operation reduces the effective SS capacitor value by the dimming ratio. CSS(MIN) > 2 • dimmin g ratio • 50µA • COUT • VOUT • RDS(ON)/SENSE 150mV • 1.2V U + R2 RUN 6V INPUT SUPPLY VIN RUN COMPARATOR BIAS AND START-UP CONTROL + – OPTIONAL FILTER CAPACITOR R1 1.248V µPOWER REFERENCE GND 3783 F05a – Figure 5a. Programming the Turn-On and Turn-Off Thresholds Using the RUN Pin + R2 1M RUN COMPARATOR RUN 6V 1.248V VIN RUN COMPARATOR RUN 6V + – INPUT SUPPLY + – 3483 F05b 3483 F05c – GND 1.248V Figure 5c. External Pull-Up Resistor on RUN Pin for “Always On” Operation assuming 50% ripple current, where RDS(ON)/SENSE represents either the RDS(ON) of the switching MOSFET or RSENSE, whichever is used on the SENSE pin. Dimming ratio is described by 1/DPWM as shown in Figure 6. Application Circuits A basic LTC3783 PWM-dimming LED application is shown on the first page of this data sheet. Operating Frequency and PWM Dimming Ratio The minimum operating frequency, fOSC, required for proper operation of a PWM dimming application depends on the minimum PWM frequency, fPWM, the dimming ratio 1/DPWM, and N, the number of fOSC cycles per PWM cycle: fOSC > N • fPWM DPWM 3783f 13 LTC3783 OPERATIO Figure 6 illustrates these various quantities in relation to one another. Typically, in order to avoid visible flicker, fPWM should be greater than 120Hz. Assuming inductor and capacitor sizing which is close to discontinuous operation, 2 fOSC cycles are sufficient for proper PWM operation. Thus, within the 1MHz rated maximum fOSC, a dimming ratio of 1/DPWM = 3000 is possible. 1/fPWM DPWM/fPWM PWMIN #=N GATE 3783 F06 1/fOSC Figure 6. PWM Dimming Parameters Boost Converter: Duty Cycle Considerations For a boost converter operating in a continuous conduction mode (CCM), the duty cycle of the main switch is: D= VOUT + VD – VIN VOUT + VD where VD is the forward voltage of the boost diode. For converters where the input voltage is close to the output voltage, the duty cycle is low, and for converters that develop a high output voltage from a low input voltage, the duty cycle is high. The maximum output voltage for a boost converter operating in CCM is: VOUT(MAX) = VIN(MIN) – VD 1 – DMAX The maximum duty cycle capability of the LTC3783 is typically 90%. This allows the user to obtain high output voltages from low input supply voltages. Boost Converter: The Peak and Average Input Currents The control circuit in the LTC3783 is measuring the input current (either by using the RDS(ON) of the power MOSFET or by using a sense resistor in the MOSFET source), so the 14 U output current needs to be reflected back to the input in order to dimension the power MOSFET properly. Based on the fact that, ideally, the output power is equal to the input power, the maximum average input current is: IIN(MAX) = IOUT(MAX) 1 – DMAX The peak input current is: ⎛ χ ⎞ IOUT(MAX ) IIN(PEAK ) = ⎜ 1+ ⎟ • ⎝ 2 ⎠ 1– DMAX The maximum duty cycle, DMAX, should be calculated at minimum VIN. Boost Converter: Ripple Current ∆IL and the ‘χ’ Factor The constant ‘χ’ in the equation above represents the percentage peak-to-peak ripple current in the inductor, relative to its maximum value. For example, if 30% ripple current is chosen, then χ = 0.3, and the peak current is 15% greater than the average. For a current mode boost regulator operating in CCM, slope compensation must be added for duty cycles above 50% in order to avoid subharmonic oscillation. For the LTC3783, this ramp compensation is internal. Having an internally fixed ramp compensation waveform, however, does place some constraints on the value of the inductor and the operating frequency. If too large an inductor is used, the resulting current ramp (∆IL) will be small relative to the internal ramp compensation (at duty cycles above 50%), and the converter operation will approach voltage mode (ramp compensation reduces the gain of the current loop). If too small an inductor is used, but the converter is still operating in CCM (near critical conduction mode), the internal ramp compensation may be inadequate to prevent subharmonic oscillation. To ensure good current mode gain and to avoid subharmonic oscillation, it is recommended that the ripple current in the inductor fall in the range of 20% to 40% of the maximum average current. For example, if the maximum average input current is 1A, choose a ∆IL between 0.2A and 0.4A, and correspondingly a value ‘χ’ between 0.2 and 0.4. 3783f LTC3783 OPERATIO Boost Converter: Inductor Selection Given an operating input voltage range, and having chosen the operating frequency and ripple current in the inductor, the inductor value can be determined using the following equation: ⎛ VIN(MIN) ⎞ • DMAX L=⎜ ⎠ ⎝ ∆IL • f ⎟ where : ∆IL = χ • IOUT(MAX ) 1 – DMAX Remember that most boost converters are not shortcircuit protected. Under a shorted output condition, the inductor current is limited only by the input supply capability. For applications requiring a step-up converter that is short-circuit protected, please refer to the applications section covering SEPIC converters. The minimum required saturation current of the inductor can be expressed as a function of the duty cycle and the load current, as follows: ⎛ χ ⎞ IOUT(MAX ) IL(SAT) > ⎜ 1+ ⎟ • ⎝ 2 ⎠ 1 – DMAX The saturation current rating for the inductor should be checked at the minimum input voltage (which results in the highest inductor current) and maximum output current. Boost Converter: Operating in Discontinuous Mode Discontinuous mode operation occurs when the load current is low enough to allow the inductor current to run out during the off-time of the switch, as shown in Figure 7. Once the inductor current is near zero, the switch and diode capacitances resonate with the inductance to form damped ringing at 1MHz to 10MHz. If the off-time is long enough, the drain voltage will settle to the input voltage. Depending on the input voltage and the residual energy in the inductor, this ringing can cause the drain of the power MOSFET to go below ground where it is clamped by the body diode. This ringing is not harmful to the IC and it has U OUTPUT VOLTAGE 200mV/DIV INDUCTOR CURRENT 1A/DIV MOSFET DRAIN VOLTAGE 20V/DIV 1µs/DIV 3783 F07 Figure 7. Discontinuous Mode Waveforms not been shown to contribute significantly to EMI. Any attempt to damp it with a snubber will degrade the efficiency. Boost Converter: Power MOSFET Selection The power MOSFET can serve two purposes in the LTC3783: it represents the main switching element in the power path, and its RDS(ON) can represent the current sensing element for the control loop. Important parameters for the power MOSFET include the drain-to-source breakdown voltage BVDSS, the threshold voltage VGS(TH), the onresistance RDS(ON) versus gate-to-source voltage, the gate-to-source and gate-to-drain charges QGS and QGD, respectively, the maximum drain current ID(MAX) and the MOSFET’s thermal resistances θJC and θJA. The gate drive voltage is set by the 7V INTVCC low drop regulator. Consequently, 6V rated MOSFETs are required in most high voltage LTC3783 applications. If low input voltage operation is expected (e.g., supplying power from a lithium-ion battery or a 3.3V logic supply), then sublogiclevel threshold MOSFETs should be used. Pay close attention to the BVDSS specifications for the MOSFETs relative to the maximum actual switch voltage in the application. Many logic-level devices are limited to 30V or less, and the switch node can ring during the turn-off of the MOSFET due to layout parasitics. Check the switching waveforms of the MOSFET directly across the drain and source terminals using the actual PC board layout for excessive ringing. 3783f 15 LTC3783 OPERATIO During the switch on-time, the IMAX comparator limits the absolute maximum voltage drop across the power MOSFET to a nominal 150mV, regardless of duty cycle. The peak inductor current is therefore limited to 150mV/RDS(ON). The relationship between the maximum load current, duty cycle, and the RDS(ON) of the power MOSFET is: RDS(ON) < 150mV • The ρT term accounts for the temperature coefficient of the RDS(ON) of the MOSFET, which is typically 0.4%/°C. Figure 8 illustrates the variation of normalized RDS(ON) over temperature for a typical power MOSFET. Another method of choosing which power MOSFET to use is to check what the maximum output current is for a given RDS(ON), since MOSFET on-resistances are available in discrete values. IO(MAX ) = 150mV • It is worth noting that the 1 - DMAX relationship between IO(MAX) and RDS(ON) can cause boost converters with a wide input range to experience a dramatic range of maximum input and output currents. This should be taken into consideration in applications where it is important to limit 2.0 ρT NORMALIZED ON RESISTANCE 1.5 1.0 0.5 0 – 50 Figure 8. Normalized RDS(ON) vs Temperature 16 U the maximum current drawn from the input supply, and also to avoid triggering the 150mV IMAX comparator, as this condition can result in excessive noise. Calculating Power MOSFET Switching and Conduction Losses and Junction Temperatures In order to calculate the junction temperature of the power MOSFET, the power dissipated by the device must be known. This power dissipation is a function of the duty cycle, the load current, and the junction temperature itself (due to the positive temperature coefficient of its RDS(ON)). As a result, some iterative calculation is normally required to determine a reasonably accurate value. Since the controller is using the MOSFET as both a switching and a sensing element, care should be taken to ensure that the converter is capable of delivering the required load current over all operating conditions (line voltage and temperature), and for the worst-case specifications for VSENSE(MAX) and the RDS(ON) of the MOSFET listed in the manufacturer’s data sheet. The power dissipated by the MOSFET in a boost converter is: ⎛ IOUT(MAX) ⎞ PFET = ⎜ ⎟ • RDS(ON) • DMAX • ρT + ⎝ 1 – DMAX ⎠ ⎛ IOUT(MAX) ⎞ k • VOUT1.85 • ⎜ ⎟ • CRSS • f ⎝ 1 – DMAX ⎠ 2 1– DMAX ⎛ χ⎞ ⎜ 1+ 2 ⎟ • IOUT(MAX ) • ρT ⎝ ⎠ 1 – DMAX ⎛ χ⎞ ⎜ 1+ 2 ⎟ • RDS(ON) • ρT ⎝ ⎠ The first term in the equation above represents the I2R losses in the device, and the second term, the switching losses. The constant k = 1.7 is an empirical factor inversely related to the gate drive current and has the dimension of 1/current. From a known power dissipated in the power MOSFET, its junction temperature can be obtained using the following formula: TJ = TA + PFET • θJA The θJA to be used in this equation normally includes the θJC for the device plus the thermal resistance from the case to the ambient temperature (θCA). This value of TJ can then be compared to the original, assumed value used in the iterative calculation process. 3783f 50 100 0 JUNCTION TEMPERATURE (°C) 150 3783 F08 LTC3783 OPERATIO Boost Converter: Output Diode Selection To maximize efficiency, a fast switching diode with low forward drop and low reverse leakage is desired. The output diode in a boost converter conducts current during the switch off-time. The peak reverse voltage that the diode must withstand is equal to the regulator output voltage. The average forward current in normal operation is equal to the output current, and the peak current is equal to the peak inductor current. VOUT (AC) ∆VCOUT ∆VESR 3783 F09 ⎛ χ ⎞ IOUT(MAX ) ID(PEAK ) = IL(PEAK ) = ⎜ 1+ ⎟ • ⎝ 2 ⎠ 1 – DMAX The power dissipated by the diode is: PD = IOUT(MAX) • VD and the diode junction temperature is: TJ = TA + PD • θJA The θJA to be used in this equation normally includes the θJC for the device plus the thermal resistance from the board to the ambient temperature in the enclosure. Remember to keep the diode lead lengths short and to observe proper switch-node layout (see Board Layout Checklist) to avoid excessive ringing and increased dissipation. Boost Converter: Output Capacitor Selection Contributions of ESR (equivalent series resistance), ESL (equivalent series inductance) and the bulk capacitance must be considered when choosing the correct component for a given output ripple voltage. The effects of these three parameters (ESR, ESL and bulk C) on the output voltage ripple waveform are illustrated in Figure 9 for a typical boost converter. The choice of component(s) begins with the maximum acceptable ripple voltage (expressed as a percentage of the output voltage), and how this ripple should be divided between the ESR step and the charging/discharging ∆V. For the purpose of simplicity we will choose 2% for the maximum output ripple, to be divided equally between the U RINGING DUE TO TOTAL INDUCTANCE (BOARD + CAP) Figure 9. Output Ripple Voltage ESR step and the charging/discharging ∆V. This percentage ripple will change, depending on the requirements of the application, and the equations provided below can easily be modified. For a 1% contribution to the total ripple voltage, the ESR of the output capacitor can be determined using the following equation: ESRCOUT < 0.01 • where : ⎛ χ ⎞ IOUT(MAX ) IIN(PEAK ) = ⎜ 1+ ⎟ • ⎝ 2 ⎠ 1 – DMAX For the bulk C component, which also contributes 1% to the total ripple: VOUT IIN(PEAK ) COUT > IOUT(MAX ) 0.01• VOUT • f For many designs it is possible to choose a single capacitor type that satisfies both the ESR and bulk C requirements for the design. In certain demanding applications, however, the ripple voltage can be improved significantly by connecting two or more types of capacitors in parallel. For example, using a low ESR ceramic capacitor can minimize the ESR setup, while an electrolytic capacitor can be used to supply the required bulk C. Once the output capacitor ESR and bulk capacitance have been determined, the overall ripple voltage waveform should be verified on a dedicated PC board (see Board 3783f 17 LTC3783 OPERATIO Layout section for more information on component placement). Lab breadboards generally suffer from excessive series inductance (due to inter-component wiring), and these parasitics can make the switching waveforms look significantly worse than they would be on a properly designed PC board. The output capacitor in a boost regulator experiences high RMS ripple currents. The RMS output capacitor ripple current is: IRMS(COUT) IOUT(MAX ) • VOUT – VIN(MIN) VIN(MIN) Note that the ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be placed in parallel to meet size or height requirements in the design. Boost Converter: Input Capacitor Selection The input capacitor of a boost converter is less critical than the output capacitor, due to the fact that the inductor is in series with the input, and hence, the input current waveform is continuous (see Figure 10). The input voltage source impedance determines the size of the input capacitor, which is typically in the range of 10µF to 100µF. A low ESR capacitor is recommended, although it is not as critical as for the output capacitor. The RMS input capacitor ripple current for a boost converter is: IRMS(CIN) 0.3 • IL Figure 10. Inductor and Input Currents 18 U Please note that the input capacitor can see a very high surge current when a battery is suddenly connected to the input of the converter, and solid tantalum capacitors can fail catastrophically under these conditions. Be sure to specify surge-tested capacitors! Boost Converter Design Example The design example given here will be for the circuit shown in Figure 1. The input voltage is 12V, and the output voltage is 25V at a maximum load current of 0.7A (1A peak). 1. The duty cycle is: D= VOUT + VD – VIN 25 + 0.4 – 12 = = 53% VOUT + VD 25 + 0.4 2. The operating frequency is chosen to be 1MHz to maximize the PWM dimming range. From Figure 2, the resistor from the FREQ pin to ground is 6k. 3. An inductor ripple current of 40% of the maximum load current is chosen, so the peak input current (which is also the minimum saturation current) is: 0.7 ⎛ χ ⎞ IOUT(MAX ) = 1.8 A IIN(PEAK ) = ⎜ 1+ ⎟ • = 1.2 • ⎝ 2 ⎠ 1 – DMAX 1 – 0.53 The inductor ripple current is: ∆IL = χ • IOUT(MAX ) 1− DMAX = 0.4 • 0.7 = 0.6 A 1− 0.53 And so the inductor value is: L= VIN(MIN) ∆ IL • f • DMAX = 12V • 0.53 = 11µH 0.6 A • 1MHz VIN(MIN) L•f • DMAX IIN 4. RSENSE should be: RSENSE = 0.5 • VSENSE(MAX ) IIN(PEAK ) = 0.5 • 150mV = 42mΩ 1.8 A 3783f LTC3783 OPERATIO 5. The diode for this design must handle a maximum DC output current of 0.7A and be rated for a minimum reverse voltage of VOUT, or 25V. A 1A, 40V diode from Zetex was chosen for its specifications, especially low leakage at higher temperatures, which is important for maintaining dimming range. 6. Voltage and value permitting, the output capacitor usually consists of some combination of low ESR ceramics. Based on a maximum output ripple voltage of 1%, or 250mV, the bulk C needs to be greater than: COUT > IOUT(MAX ) 0.01 • VOUT • f = 0.7 A = 3µF 0.01 • 25V • 1MHz The RMS ripple current rating for this capacitor needs to exceed: IRMS(COUT ) = IOUT(MAX ) • VOUT – VIN(MIN) VIN(MIN) 25V – 12V = 0.7 A = 0.7 A • 12V Based on value and ripple current, and taking physical size into account, a surface mount ceramic capacitor is a good choice. A 4.7µF TDK C5750X7R1H475M will satisfy all requirements in a compact package. 7. The soft-start capacitor should be: CSS(MIN) > 2 • dimmin g ratio • 50µA • COUT • VOUT • RDS(ON)/ SENSE 150mV • 1.2V 2 • 3000 • 50µA • 4.7µF • 25V • 42mΩ = 8µF > 150mV • 1.2V 8. The choice of an input capacitor for a boost converter depends on the impedance of the source supply and the amount of input ripple the converter will safely tolerate. For this particular design and lab setup, 20µF was found to be satisfactory. PC Board Layout Checklist 1. In order to minimize switching noise and improve output load regulation, the GND pad of the LTC3783 should be connected directly to 1) the negative terminal of U the INTVCC decoupling capacitor, 2) the negative terminal of the output decoupling capacitors, 3) the bottom terminals of the sense resistors or the source of the power MOSFET, 4) the negative terminal of the input capacitor, and 5) at least one via to the ground plane immediately under the exposed pad. The ground trace on the top layer of the PC board should be as wide and short as possible to minimize series resistance and inductance. 2. Beware of ground loops in multiple layer PC boards. Try to maintain one central ground node on the board and use the input capacitor to avoid excess input ripple for high output current power supplies. If the ground plane is to be used for high DC currents, choose a path away from the small-signal components. 3. Place the CVCC capacitor immediately adjacent to the INTVCC and GND pins on the IC package. This capacitor carries high di/dt MOSFET gate-drive currents. A low ESR and ESL 4.7µF ceramic capacitor works well here. 4. The high di/dt loop from the bottom terminal of the output capacitor, through the power MOSFET, through the boost diode and back through the output capacitors should be kept as tight as possible to reduce inductive ringing. Excess inductance can cause increased stress on the power MOSFET and increase HF noise on the output. If low ESR ceramic capacitors are used on the output to reduce output noise, place these capacitors close to the boost diode in order to keep the series inductance to a minimum. 5. Check the stress on the power MOSFET by measuring its drain-to-source voltage directly across the device terminals (reference the ground of a single scope probe directly to the source pad on the PC board). Beware of inductive ringing which can exceed the maximum specified voltage rating of the MOSFET. If this ringing cannot be avoided and exceeds the maximum rating of the device, either choose a higher voltage device or specify an avalanche-rated power MOSFET. 6. Place the small-signal components away from high frequency switching nodes. All of the small-signal components should be placed on one side of the IC and all of the power components should be placed on the other. This also allows the use of a pseudo-Kelvin connection for the signal ground, where high di/dt gate driver currents flow 3783f 19 LTC3783 OPERATIO out of the IC ground pad in one direction (to bottom plate of the INTVCC decoupling capacitor) and small-signal currents flow in the other direction. 7. If a sense resistor is used in the source of the power MOSFET, minimize the capacitance between the SENSE pin trace and any high frequency switching nodes. The LTC3783 contains an internal leading-edge blanking time of approximately 160ns, which should be adequate for most applications. 8. For optimum load regulation and true remote sensing, the top of the output resistor should connect independently to the top of the output capacitor (Kelvin connection), staying away from any high dV/dt traces. Place the divider resistors near the LTC3783 in order to keep the high impedance FBN node short. 9. For applications with multiple switching power converters connected to the same input supply, make sure that the input filter capacitor for the LTC3783 is not shared with any other converters. AC input current from another converter could cause substantial input voltage ripple, and this could interfere with the operation of the LTC3783. A few inches of PC trace or wire (L ~ 100nH) between the CIN of the LTC3783 and the actual source VIN should be sufficient to prevent current-sharing problems. PWM 5V AT 0Hz TO 10Hz 100k 1µF 4.7µF 20k Figure 11. Single Inductor Buck-Boost Application with Analog Dimming and Low Frequency PWM Dimming 20 U Returning the Load to VIN: A Single Inductor Buck-Boost Application As shown in Figure 11, due to its available high side current sensing mode, the LTC3783 is also well-suited to a boost converter in which the load current is returned to VIN, hence providing a load voltage (VOUT – VIN) which can be greater or less than the input voltage VIN. This configuration allows for complete overlap of input and output voltages, with the disadvantages that only the load current, and not the load voltage, can be tightly regulated. The switch must be rated for a VDS(MAX) equal to VIN + VLOAD. The design of this circuit resembles that of the boost converter above, and the procedure is much the same, except VOUT is now (VIN + VLOAD), and the duty cycles and voltages must be adjusted accordingly. Similar to the boost converter, which can be dimmed via the digital PWMIN input or the analog FBP pin, the buckboost can be dimmed via the PWMIN pin or the analog ILIM pin, which adjusts the offset voltage to which the loop will drive (VFBP – VFBN). In the case of the buckboost, however, the dimming ratio cannot be as high as in the boost converter, since there is no load switch to preserve the VOUT level while PWMIN is low. VIN 1M 10µF, 50V ×2 UMK432C106MM 10µH SUMIDA CDRH8D28-100 PMEG6010 40.2k 0V TO 1.23V RL 9V TO 26V 0.28Ω LED STRING 1-4 EA LUMILEDS LHXL-BW02 EACH LED IS 3V TO 4.2V AT 350mA VOUT LTC3783 RUN VIN PWMIN OV/FB ITH PWMOUT SS ILIM GATE VREF FBP SENSE FBN INTVCC GND FREQ SYNC FAIRCHILD FDN5630 10µF, 50V C5750X7R1H106M CERAMIC 4.7µF 0.05Ω 1k GND 3783 F11 3783f LTC3783 OPERATIO Using the LTC3783 for Buck Applications As shown in Figure 12, high side current sensing also allows the LTC3783 to control a functional buck converter when load voltage is always sufficiently less than VIN 6V TO 36V U VIN. In this scheme the input voltage to the inductor is lowered by the load voltage. The boost converter now sees a VIN’ = VIN – VLOAD, meaning the controller is now boosting from (VIN – VLOAD) to VIN. LED STRING LTC3783 RUN VIN PWMIN OV/FB ITH PWMOUT SS ILIM GATE VREF FBP SENSE FBN INTVCC GND FREQ SYNC GND 3783 F12 Figure 12. LED Buck Application 3783f 21 LTC3783 PACKAGE DESCRIPTIO 4.50 ± 0.05 3.10 ± 0.05 2.44 ± 0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS 5.00 ± 0.10 (2 SIDES) R = 0.20 TYP R = 0.115 TYP 9 16 0.40 ± 0.10 PIN 1 TOP MARK (SEE NOTE 6) 8 0.200 REF 0.75 ± 0.05 4.34 ± 0.10 (2 SIDES) BOTTOM VIEW—EXPOSED PAD 1 0.25 ± 0.05 0.50 BSC NOTE: 1. DRAWING PROPOSED TO BE MADE VARIATION OF VERSION (WJGD-2) IN JEDEC PACKAGE OUTLINE MO-229 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 22 U DHD Package 16-Lead Plastic DFN (5mm × 4mm) (Reference LTC DWG # 05-08-1707) 0.70 ± 0.05 PACKAGE OUTLINE 0.25 ± 0.05 0.50 BSC 4.34 ± 0.05 (2 SIDES) 4.00 ± 0.10 (2 SIDES) 2.44 ± 0.10 (2 SIDES) PIN 1 NOTCH (DHD16) DFN 0504 0.00 – 0.05 3783f LTC3783 PACKAGE DESCRIPTIO 3.58 (.141) 6.60 ± 0.10 4.50 ± 0.10 SEE NOTE 4 0.65 BSC RECOMMENDED SOLDER PAD LAYOUT 4.30 – 4.50* (.169 – .177) 0.09 – 0.20 (.0035 – .0079) 0.50 – 0.75 (.020 – .030) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U FE Package 16-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663) Exposed Pad Variation BC 4.90 – 5.10* (.193 – .201) 3.58 (.141) 16 1514 13 12 1110 9 2.94 (.116) 0.45 ± 0.05 1.05 ± 0.10 6.40 2.94 (.252) (.116) BSC 12345678 1.10 (.0433) MAX 0° – 8° 0.25 REF 0.65 (.0256) BSC 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE16 (BC) TSSOP 0204 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3783f 23 LTC3783 RELATED PARTS PART NUMBER LT 1618 LTC1871 LT3477 LTC3780 LTC3782 ® DESCRIPTION Monolithic 1.4MHz Boost Regulator Boost, Flyback, SEPIC Controller 3A DC/DC LED Driver with Rail-to-Rail Current Sense High Power Buck-Boost Controller 2-Phase Boost Controller COMMENTS Constant-Current/Constant-Voltage, 1A Switch No RSENSE, 2.5V ≤ VIN ≤ 36V, 92% Duty Cycle 2.5V ≤ VIN ≤ 25V: Buck, Buck-Boost and Boost Topologies 4-Switch, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 30V High Power, 6V ≤ VIN ≤ 40V, 150kHz to 500kHz 2-Phase, 80µA IQ, 0.8V ≤ VOUT ≤ 10V, 4V ≤ VIN ≤ 36V 1- and 2-Cell, 4.7V ≤ VIN ≤ 22V, 3 Hour Timer LTC3827/LTC3827-1 Low IQ Current Dual Controllers LTC4002 Standalone 2A Li-Ion Battery Charger 3783f 24 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● LT 1105 • PRINTED IN THE USA www.linear.com © LINEAR TECHNOLOGY CORPORATION 2005