LTC3803HS6-5

LTC3803-5 Constant Frequency Current Mode Flyback DC/DC Controller in ThinSOT FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO VIN and VOUT Limited Only by External Components 4.8V Undervoltage Lockout Threshold Operating Junction Temperature from –40°C to 150°C Adjustable Slope Compensation Internal Soft-Start Constant Frequency 200kHz Operation ±1.5% Reference Accuracy Current Mode Operation for Excellent Line and Load Transient Response No Minimum Load Requirement Low Quiescent Current: 240µA Low Profile (1mm) SOT-23 Package The LTC®3803-5 is a constant frequency current mode flyback controller optimized for driving 4.5V and 6V-rated N-channel MOSFETs in high input voltage applications. The LTC3803-5 operates from inputs as low as 5V. Constant frequency operation is maintained down to very light loads, resulting in less low frequency noise generation over a wide range of load currents. Slope compensation can be programmed with an external resistor. The LTC3803-5 provides ±1.5% output voltage accuracy and consumes only 240µA of quiescent current. Groundreferenced current sensing allows LTC3803-5-based converters to accept input supplies beyond the LTC3803-5’s absolute maximum VCC. For simplicity, the LTC3803-5 can be powered from a high VIN through a resistor, due to its internal 8V shunt regulator. An internal undervoltage lockout shuts down the IC when the input voltage falls below 3.2V, guaranteeing at least 3.2V of gate drive to the external MOSFET. The LTC3803-5 is available in a low profile (1mm) 6-lead SOT-23 (ThinSOTTM) package. APPLICATIO S ■ ■ ■ ■ 42V and 12V Automotive Power Supplies Telecom Power Supplies Auxiliary/Housekeeping Power Supplies Power Over Ethernet , LTC and LT are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. ThinSOT is a trademark of Linear Technology Corporation. TYPICAL APPLICATIO VIN 6V TO 50V PDZ9.1B 22k Dual Output Wide Input Range Converter VPH5-0155 1µF 100V 3x MMBTA42 10nF LTC3803-5 ITH/RUN NGATE GND 8.06k VFB VCC SENSE 0.1µF 57.6k ALL CAPACITORS ARE X7R, TDK 38035 TA01 10MQ100N 13V/0.3A 20mA MIN LOAD 90 85 VIN = 12V EFFICIENCY (%) 22µF 10V 1µF 100V 1µF 100V 80 VIN = 24V 75 70 VIN = 48V 65 60 VIN = 12V 0 2 6 8 4 OUTPUT POWER (W) 10 7.5k PHM25NQ10T 4.7k B3100 0.012Ω 6.5V/1.2A 47µF 10V U Efficiency and Power Loss vs Output Power VIN = 8V 3.0 2.5 U U POWER LOSS (W) 38035f 2.0 1.5 1.0 0.5 0 12 38035 TA01b 1 LTC3803-5 ABSOLUTE MAXIMUM RATINGS (Note 1) PACKAGE/ORDER INFORMATION TOP VIEW ITH/RUN 1 GND 2 VFB 3 6 NGATE 5 VCC 4 SENSE VCC to GND (Current Fed) .................... 25mA into VCC* NGATE Voltage ......................................... – 0.3V to VCC VFB, ITH/RUN Voltages ..............................– 0.3V to 3.5V SENSE Voltage ........................................... – 0.3V to 1V NGATE Peak Output Current ( VITHSHDN VITH/RUN < VITHSHDN ITH/RUN AND VCC > VTURNON (NOMINALLY 0.28V) (NOMINALLY 4.8V) LTC3803-5 ENABLED 38035 F01 Figure 1. Start-Up/Shutdown State Diagram 38035f 7 LTC3803-5 OPERATIO The undervoltage lockout (UVLO) mechanism prevents the LTC3803-5 from trying to drive a MOSFET with insufficient VGS. The voltage at the VCC pin must exceed VTURNON (nominally 4.8V) at least momentarily to enable LTC3803-5 operation. The VCC voltage is then allowed to fall to VTURNOFF (nominally 4V) before undervoltage lockout disables the LTC3803-5. The ITH/RUN pin can be driven below VSHDN (nominally 0.28V) to force the LTC3803-5 into shutdown. An internal 0.3µA current source always tries to pull this pin towards VCC. When the ITH/RUN pin voltage is allowed to exceed VSHDN, and VCC exceeds VTURNON, the LTC3803-5 begins to operate and an internal clamp immediately pulls the ITH/RUN pin up to about 0.7V. In operation, the ITH/RUN pin voltage will vary from roughly 0.7V to 1.9V to represent current comparator thresholds from zero to maximum. Internal Soft-Start An internal soft-start feature is enabled whenever the LTC3803-5 comes out of shutdown. Specifically, the ITH/RUN voltage is clamped and is prevented from reaching maximum until roughly 0.7ms has passed. This allows the input and output currents of LTC3803-5based power supplies to rise in a smooth and controlled manner on start-up. 8 U Powering the LTC3803-5 In the simplest case, the LTC3803-5 can be powered from a high voltage supply through a resistor. A built-in shunt regulator from the VCC pin to GND will draw as much current as needed through this resistor to regulate the VCC voltage to around 8V as long as the VCC pin is not forced to sink more than 25mA. This shunt regulator is always active, even when the LTC3803-5 is in shutdown, since it serves the vital function of protecting the VCC pin from seeing too much voltage. The VCC pin must be bypassed to ground immediately adjacent to the IC pins with a ceramic or tantalum capacitor. Proper supply bypassing is necessary to supply the high transient currents required by the MOSFET gate driver. 10µF is a good starting point. Adjustable Slope Compensation The LTC3803-5 injects a 5µA peak current ramp out through its SENSE pin which can be used for slope compensation in designs that require it. This current ramp is approximately linear and begins at zero current at 6.5% duty cycle, reaching peak current at 80% duty cycle. Additional details are provided in the Applications Information section. 38035f LTC3803-5 APPLICATIO S I FOR ATIO Many LTC3803-5 application circuits can be derived from the topology shown in Figure 2. The LTC3803-5 itself imposes no limits on allowed power output, input voltage VIN or desired regulated output voltage VOUT; these are all determined by the ratings on the external power components. The key factors are: Q1’s maximum drain-source voltage (BVDSS), on-resistance (RDS(ON)) and maximum drain current, T1’s saturation flux level and winding insulation breakdown voltages, CIN and COUT’s maximum working voltage, ESR, and maximum ripple current ratings, and D1 and RSENSE’s power ratings. VIN T1 RVCC 5 1 CC 2 VCC ITH/RUN NGATE LTC3803-5 GND VFB R1 3 R2 38035 F02 D1 • CIN LPRI LSEC COUT • CVCC 6 4 RSL Q1 SENSE RSENSE Figure 2. Typical LTC3803-5 Application Circuit SELECTING FEEDBACK RESISTOR DIVIDER VALUES The regulated output voltage is determined by the resistor divider across VOUT (R1 and R2 in Figure 2). The ratio of R2 to R1 needed to produce a desired VOUT can be calculated: R2 = VOUT – 0.8 V • R1 0.8 V Choose resistance values for R1 and R2 to be as large as possible in order to minimize any efficiency loss due to the static current drawn from VOUT, but just small enough so that when VOUT is in regulation, the error caused by the nonzero input current to the VFB pin is less than 1%. A good rule of thumb is to choose R1 to be 80k or less. U TRANSFORMER DESIGN CONSIDERATIONS Transformer specification and design is perhaps the most critical part of applying the LTC3803-5 successfully. In addition to the usual list of caveats dealing with high frequency power transformer design, the following should prove useful. Turns Ratios Due to the use of the external feedback resistor divider ratio to set output voltage, the user has relative freedom in selecting transformer turns ratio to suit a given application. Simple ratios of small integers, e.g., 1:1, 2:1, 3:2, etc. can be employed which yield more freedom in setting total turns and mutual inductance. Simple integer turns ratios also facilitate the use of “off-the-shelf” configurable transformers such as the Coiltronics VERSA-PACTM series in applications with high input to output voltage ratios. For example, if a 6-winding VERSA-PAC is used with three windings in series on the primary and three windings in parallel on the secondary, a 3:1 turns ratio will be achieved. Turns ratio can be chosen on the basis of desired duty cycle. However, remember that the input supply voltage plus the secondary-to-primary referred version of the flyback pulse (including leakage spike) must not exceed the allowed external MOSFET breakdown rating. Leakage Inductance Transformer leakage inductance (on either the primary or secondary) causes a voltage spike to occur after the output switch (Q1) turn-off. This is increasingly prominent at higher load currents, where more stored energy must be dissipated. In some cases a “snubber” circuit will be required to avoid overvoltage breakdown at the MOSFET’s drain node. Application Note 19 is a good reference on snubber design. A bifilar or similar winding technique is a good way to minimize troublesome leakage inductances. However, remember that this will limit the primary-to-secondary breakdown voltage, so bifilar winding is not always practical. VERSA-PAC is a trademark of Coiltronics, Inc. 38035f W UU VOUT 9 LTC3803-5 APPLICATIO S I FOR ATIO CURRENT SENSE RESISTOR CONSIDERATIONS The external current sense resistor (RSENSE in Figure 2) allows the user to optimize the current limit behavior for the particular application. As the current sense resistor is varied from several ohms down to tens of milliohms, peak switch current goes from a fraction of an ampere to several amperes. Care must be taken to ensure proper circuit operation, especially with small current sense resistor values. For example, a peak switch current of 5A requires a sense resistor of 0.020Ω. Note that the instantaneous peak power in the sense resistor is 0.5W and it must be rated accordingly. The LTC3803-5 has only a single sense line to this resistor. Therefore, any parasitic resistance in the ground side connection of the sense resistor will increase its apparent value. In the case of a 0.020Ω sense resistor, one milliohm of parasitic resistance will cause a 5% reduction in peak switch current. So the resistance of printed circuit copper traces and vias cannot necessarily be ignored. PROGRAMMABLE SLOPE COMPENSATION The LTC3803-5 injects a ramping current through its SENSE pin into an external slope compensation resistor (RSL in Figure 2). This current ramp starts at zero right after the NGATE pin has been high for the LTC3803-5’s minimum duty cycle of 6.5%. The current rises linearly towards a peak of 5µA at the maximum duty cycle of 80%, shutting off once the NGATE pin goes low. A series resistor (RSL) connecting the SENSE pin to the current sense resistor (RSENSE) thus develops a ramping voltage drop. From the perspective of the SENSE pin, this ramping voltage adds to the voltage across the sense resistor, effectively reducing the current comparator threshold in proportion to duty cycle. This stabilizes the control loop VIN RVCC LTC3803-5 VCC CVCC GND 38035 F03 Figure 3. Powering the LTC3803-5 Via the Internal Shunt Regulator 10 U against subharmonic oscillation. The amount of reduction in the current comparator threshold (∆VSENSE) can be calculated using the following equation: ∆VSENSE = Duty Cycle – 6.5% • 5µA • RSL 73.5% Note: LTC3803-5 enforces 6.5% < Duty Cycle < 80%. A good starting value for RSL is 5.9k, which gives a 30mV drop in current comparator threshold at 80% duty cycle. Designs not needing slope compensation may replace RSL with a short circuit. VCC SHUNT REGULATOR An internal shunt regulator allows the LTC3803-5 to be powered through a single dropping resistor from VIN to VCC, in conjunction with a bypass capacitor, CVCC, that closely decouples VCC to GND (see Figure 3). The shunt regulator can draw up to 25mA through the VCC pin to GND to drop enough voltage across RVCC to regulate VCC to around 8V. For applications where VIN is low enough such that the static power dissipation in RVCC is acceptable, using the VCC shunt regulator is the simplest way to power the LTC3803-5. EXTERNAL PREREGULATOR The circuit in Figure 4 shows another way to power the LTC3803-5. An external series preregulator consisting of series pass transistor Q1, Zener diode D1, and bias resistor RB brings VCC above the VCC turn-on threshold, enabling the LTC3803-5. 8V TO 75 VIN RB 100k D1 6.8V Q1 MMBTA42 CVCC 0.1µF LTC3803-5 VCC GND 38035 F04 W UU Figure 4. Powering the LTC3803-5 with an External Preregulator 38035f LTC3803-5 TYPICAL APPLICATIO S 2W Isolated Housekeeping Telecom Converter PRIMARY SIDE 10V, 100mA OUTPUT BAS516 T1 VIN 36V TO 75V 9.2k 1nF BAS516 1k 220k 806Ω PACKAGE DESCRIPTIO 0.62 MAX 0.95 REF 3.85 MAX 2.62 REF RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.20 BSC 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U U • 2.2µF 1µ F BAS516 • 2.2µF • FDC2512 SECONDARY SIDE 10V, 100mA OUTPUT 22k 1 2 3 LTC3803-5 6 ITH/RUN NGATE GND VFB VCC SENSE 5 4 5.6k 1µ F PRIMARY GROUND SECONDARY SIDE GROUND T1: PULSE ENGINEERING PA0648 OR TYCO TTI8698 0.1Ω 38035 TA03 S6 Package 6-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1636) 2.90 BSC (NOTE 4) 1.22 REF 1.4 MIN 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE ID 0.95 BSC 0.30 – 0.45 6 PLCS (NOTE 3) 0.80 – 0.90 0.01 – 0.10 0.09 – 0.20 (NOTE 3) 1.90 BSC S6 TSOT-23 0302 38035f 11 LTC3803-5 TYPICAL APPLICATIO S 90% Efficient Synchronous Flyback Converter VIN 36V TO 72V 270k CIN T1 VOUT* 3.3V 1.5A CO • 1n D1 ITH/RUN GATE Q1 560 5k 38035 TA04a • 0.1µF EFFICIENCY (%) EFFICIENCY (%) 33k 1 2 6 LTC3803-5 5 VCC GND VFB SENSE 4 1µF 10V 8.06k 3 25.5k* RFB VOUT RCS T1: PULSE ENGINEERING PA1006 Q1: FAIRCHILD FDC2512 Q2: VISHAY Si9803 RCS: VISHAY OR IRC, 80mΩ D1: PHILIPS BAS516 CIN: TDK 1µF, 100V, X5R *FOR 5V OUTPUT CHANGE CO: TDK 100µF, 6.3V, X5R RFB TO 42.2k RELATED PARTS PART NUMBER LT®1425 LT1725 LTC1772 LTC1871 LTC1872 LT1950 LT1952 LT3420 LT3468/LT3468-1 LTC3806 LTC4441 DESCRIPTION Isolated Flyback Switching Regulator with No External Power Devices General Purpose Isolated Flyback Controller SOT-23 Constant Frequency Current Mode Step-Down DC/DC Controller Wide Input Range, No RSENSETM Current Mode Flyback, Boost and SEPIC Controller SOT-23 Constant Frequency Current Mode Boost DC/DC Controller Current Mode PWM Controller Current Mode PWM Controller Photoflash Capacitor Charger with Automatic Refresh Photoflash Capacitor Charger in 5-Pin SOT-23 Synchronous Flyback Controller 6A N-Channel MOSFET Driver COMMENTS No Optoisolator or “Third Winding” Required, Up to 6W Output No Optoisolator Required, VIN and VOUT Limited Only by External Power Components 550kHz Switching Frequency, 2.4V to 9.8V VIN Range Adjustable Switching Frequency, Programmable Undervoltage Lockout, Optional Burst Mode® Operation at Light Load 550kHz Switching Frequency, 2.4V to 9.8V VIN Range Controller for Forward Converters from 30W to 300W Synchronous Controller for Forward Converters from 30W to 500W Specialized Flyback Charges High Voltage Photoflash Capacitors Quickly and Efficiently Minimal Component Count, Uses Small Transformers; VIN from 2.5V to 16V High Efficiency (89%); Multiple Output with Excellent Cross Regulation Gate Drive Adjustable from 5V to 8V, Adjustable Blanking Prevents Ringing, 10-Lead MSSOP Package Burst Mode is a registered trademark of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation. 12 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● U Synchronous Flyback 3.3VOUT 91 92 91 90 Synchronous Flyback 5VOUT • Q2 90 89 88 87 86 89 88 0.5 1.5 1.0 OUTPUT CURRENT (A) 2.0 38035 TA04b 85 0.5 1.0 1.5 2.0 OUTPUT CURRENT (A) 2.5 38035 TA04c 38035f LT/TP 1104 1K • PRINTED IN THE USA www.linear.com © LINEAR TECHNOLOGY CORPORATION 2004