LTC3809-1

LTC3809-1 No RSENSETM, Low Input Voltage, Synchronous DC/DC Controller with Output Tracking FEATURES n n n n n n n n n n n n n n DESCRIPTION The LTC®3809-1 is a synchronous step-down switching regulator controller that drives external complementary power MOSFETs using few external components. The constant frequency current mode architecture with MOSFET VDS sensing eliminates the need for a current sense resistor and improves efficiency. Optional Burst Mode operation provides high efficiency operation at light loads. 100% duty cycle provides low dropout operation, extending operating time in batterypowered systems. Burst Mode is inhibited when the MODE pin is pulled low to reduce noise and RF interference. The LTC3809-1 allows either coincident or ratiometric output voltage tracking. Switching frequency is fixed at 550kHz. Fault protection is provided by an overvoltage comparator and a short-circuit current limit comparator. The LTC3809-1 is available in tiny footprint thermally enhanced DFN and 10-lead MSOP packages. , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. Burst is a registered trademark of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents including 5481178, 5929620, 6580258, 6304066, 5847554, 6611131, 6498466. Other Patents pending. Programmable Output Voltage Tracking No Current Sense Resistor Required Constant Frequency Current Mode Operation for Excellent Line and Load Transient Response Wide VIN Range: 2.75V to 9.8V Wide VOUT Range: 0.6V to VIN 0.6V ±1.5% Reference Low Dropout Operation: 100% Duty Cycle Selectable Burst Mode®/Pulse-Skipping/Forced Continuous Operation Auxiliary Winding Regulation Internal Soft-Start Circuitry Selectable Maximum Peak Current Sense Threshold Output Overvoltage Protection Micropower Shutdown: IQ = 9μA Tiny Thermally Enhanced Leadless (3mm × 3mm) DFN and 10-lead MSOP Packages APPLICATIONS n n n n 1- or 2-Cell Lithium-Ion Powered Devices Notebook and Palmtop Computers, PDAs Portable Instruments Distributed DC Power Systems TYPICAL APPLICATION High Efficiency, 550kHz Step-Down Converter 10μF IPRG MODE 59k 15k 187k 470pF VFB ITH RUN GND 38091 TA01 Efficiency and Power Loss vs Load Current 100 VIN 2.75V TO 9.8V 90 EFFICIENCY (%) EFFICIENCY VIN = 3.3V VIN = 5V 80 TYPICAL POWER LOSS (VIN = 4.2V) VIN = 4.2V 100 1k POWER LOSS (mW) 10k VIN TG 2.2μH SW BG 47μF VOUT 2.5V 2A LTC3809-1 70 10 60 FIGURE 8 CIRCUIT VOUT = 2.5V 1 10 100 1k LOAD CURRENT (mA) 1 50 0.1 10k 38091fc 38091 TA02 1 LTC3809-1 ABSOLUTE MAXIMUM RATINGS (Note 1) Input Supply Voltage (VIN) ........................ –0.3V to 10V RUN, TRACK/SS, MODE, IPRG Voltages ............................... –0.3V to (VIN + 0.3V) VFB, ITH Voltages ...................................... –0.3V to 2.4V SW Voltage ......................... –2V to VIN + 1V (10V Max) TG, BG Peak Output Current ( 5V) may work fine at lower voltages (e.g., 3.3V). Selecting the N-channel MOSFET is typically easier, since for a given RDS(ON), the gate charge and turn-on and turn-off delays are much smaller than for a P-channel MOSFET. Inductor Value Calculation Given the desired input and output voltages, the inductor value and operating frequency, fOSC , directly determine the inductor’s peak-to-peak ripple current: IRIPPLE = VOUT VIN – VOUT • VIN fOSC • L The corresponding average current depends on the amount of ripple current. Lower inductor values (higher IRIPPLE) will reduce the load current at which Burst Mode operation begins. The ripple current is normally set so that the inductor current is continuous during the burst periods. Therefore, IRIPPLE ≤ IBURST(PEAK) This implies a minimum inductance of: L MIN ≤ VIN – VOUT V • OUT fOSC • IBURST(PEAK ) VIN Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors and output voltage ripple. Thus, highest efficiency operation is obtained at low frequency with a small ripple current. Achieving this, however, requires a large inductor. A reasonable starting point is to choose a ripple current that is about 40% of IOUT(MAX). Note that the largest ripple current occurs at the highest input voltage. To guarantee that ripple current does not exceed a specified maximum, the inductor should be chosen according to: L≥ VIN – VOUT VOUT • fOSC • IRIPPLE VIN A smaller value than LMIN could be used in the circuit, although the inductor current will not be continuous during burst periods, which will result in slightly lower efficiency. In general, though, it is a good idea to keep IRIPPLE comparable to IBURST(PEAK). Inductor Core Selection Once the value of L is known, the type of inductor must be selected. Actual core loss is independent of core size for a fixed inductor value, but is very dependent on the inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard”, which means that inductance collapses abruptly when the peak design current is exceeded. Core saturation results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Burst Mode Operation Considerations The choice of RDS(ON) and inductor value also determines the load current at which the LTC3809-1 enters Burst Mode operation. When bursting, the controller clamps the peak inductor current to approximately: 1 ΔVSENSE(MAX ) IBURST(PEAK ) = • 4 RDS(ON) 38091fc 14 LTC3809-1 APPLICATIONS INFORMATION Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar characteristics. The choice of which style inductor to use mainly depends on the price vs size requirements and any radiated field/EMI requirements. New designs for surface mount inductors are available from Coiltronics, Coilcraft, Toko and Sumida. Schottky Diode Selection (Optional) The schottky diode D in Figure 9 conducts current during the dead time between the conduction of the power MOSFETs. This prevents the body diode of the bottom N-channel MOSFET from turning on and storing charge during the dead time, which could cost as much as 1% in efficiency. A 1A Schottky diode is generally a good size for most LTC3809-1 applications, since it conducts a relatively small average current. Larger diode results in additional transition losses due to its larger junction capacitance. This diode may be omitted if the efficiency loss can be tolerated. CIN and COUT Selection In continuous mode, the source current of the P-channel MOSFET is a square wave of duty cycle (VOUT /VIN). To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN Re quiredIRMS ≈ IMAX • VOUT • ( VIN – VOUT ) VIN 1/ 2 This formula has a maximum value at VIN = 2VOUT, where IRMS = IOUT /2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet the size or height requirements in the design. Due to the high operating frequency of the LTC3809-1, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. The selection of COUT is driven by the effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (ΔVOUT) is approximated by: ⎛ ⎞ 1 ΔVOUT ≈ IRIPPLE • ⎜ ESR + ⎟ 8 • f • COUT ⎠ ⎝ where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage since IRIPPLE increase with input voltage. Setting Output Voltage The LTC3809-1 output voltage is set by an external feedback resistor divider carefully placed across the output, as shown in Figure 3. The regulated output voltage is determined by: ⎛ R⎞ VOUT = 0.6 V • ⎜ 1 + B ⎟ ⎝ RA ⎠ 38091fc 15 LTC3809-1 APPLICATIONS INFORMATION For most applications, a 59k resistor is suggested for RA. In applications where minimizing the quiescent current is critical, RA should be made bigger to limit the feedback divider current. If RB then results in very high impedance, it may be beneficial to bypass RB with a 50pF to 100pF capacitor CFF. VOUT RB CFF LTC3809-1 VFB Once the controller is enabled, the start-up of VOUT is controlled by the state of the TRACK/SS pin. If the TRACK/SS pin is connected to VIN, the start-up of VOUT is controlled by internal soft-start, which slowly ramps the positive reference to the error amplifier from 0V to 0.6V, allowing VOUT to rise smoothly from 0V to its final value. The default internal soft-start time is around 0.74ms. The soft-start time can be changed by placing a capacitor between the TRACK/SS pin and GND. In this case, the soft-start time will be approximately: tSS = CSS • 600mV 1μA RA 38091 F03 Figure 3. Setting Output Voltage where 1μA is an internal current source which is always on. When the voltage on the TRACK/SS pin is less than the internal 0.6V reference, the LTC3809-1 regulates the VFB voltage to the TRACK/SS pin voltage instead of 0.6V. Therefore the start-up of VOUT can ratiometrically track an external voltage VX, according to a ratio set by a resistor divider at TRACK/SS pin (Figure 5a). The ratiometric relation between VOUT and VX is (Figure 5c): VOUT R TA R A + RB = • VX R A R TA + R TB VOUT VX LTC3809-1 Run and Soft-Start/Tracking Functions The LTC3809-1 has a low power shutdown mode which is controlled by the RUN pin. Pulling the RUN pin below 1.1V puts the LTC3809-1 into a low quiescent current shutdown mode (IQ = 9μA). Releasing the RUN pin, an internal 0.7μA (at VIN = 4.2V) current source will pull the RUN pin up to VIN, which enables the controller. The RUN pin can be driven directly from logic as showed in Figure 4. 3.3V OR 5V LTC3809-1 RUN LTC3809-1 RUN 38091 F04 RB RA RTB RTA VFB TRACK/SS 38091 F5a Figure 4. RUN Pin Interfacing Figure 5a. Using the TRACK/SS Pin to Track VX 38091fc 16 LTC3809-1 APPLICATIONS INFORMATION VX OUTPUT VOLTAGE OUTPUT VOLTAGE VX VOUT VOUT 38091 F05b,c TIME TIME (5b) Coincident Tracking (5c) Ratiometric Tracking Figure 5b and 5c. Two Different Modes of Output Voltage Tracking For coincident tracking (VOUT = VX during start-up), RTA = RA, RTB = RB VX should always be greater than VOUT when using the tracking function of TRACK/SS pin. The internal current source (1μA), which is for external soft-start, will cause a tracking error at VOUT. For example, if a 59k resistor is chosen for RTA, the RTA current will be about 10μA (600mV/59k). In this case, the 1μA internal current source will cause about 10% (1μA/10μA • 100%) tracking error, which is about 60mV (600mV • 10%) referred to VFB. This is acceptable for most applications. If a better tracking accuracy is required, the value of RTA should be reduced. Table 1 summarizes the different states in which the TRACK/SS can be used. Table 1. The States of the TRACK/SS Pin TRACK/SS Pin Capacitor CSS VIN Resistor Divider FREQUENCY External Soft-Start Internal Soft-Start VOUT Tracking an External Voltage VX Auxiliary Winding Control Using the MODE Pin The MODE pin can be used as an auxiliary feedback to provide a means of regulating a flyback winding output. When this pin drops below its ground-referenced 0.4V threshold, continuous mode operation is forced. During continuous mode, current flows continuously in the transformer primary side. The auxiliary winding draws current only when the bottom synchronous N-channel MOSFET is on. When primary load currents are low and/ or the VIN /VOUT ratio is close to unity, the synchronous MOSFET may not be on for a sufficient amount of time to transfer power from the output capacitor to the auxiliary load. Forced continuous operation will support an auxiliary winding as long as there is a sufficient synchronous MOSFET duty factor. The MODE input pin removes the requirement that power must be drawn from the transformer primary side in order to extract power from the auxiliary winding. With the loop in continuous mode, the auxiliary output may nominally be loaded without regard to the primary output load. 38091fc 17 LTC3809-1 APPLICATIONS INFORMATION The auxiliary output voltage VAUX is normally set, as shown in Figure 6, by the turns ratio N of the transformer: VAUX = (N + 1) • VOUT VAUX 1μF VOUT NORMALIZED VOLTAGE OR CURRENT (%) In a hard short (VOUT = 0V), the top P-channel MOSFET is turned off and kept off until the short-circuit condition is cleared. In this case, there is no current path from input supply (VIN) to either VOUT or GND, which prevents excessive MOSFET and inductor heating. 105 100 95 90 85 80 75 2.0 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0 INPUT VOLTAGE (V) 38091 F07 VIN LTC3809-1 R6 MODE R5 BG TG SW L1 1:N + VREF + COUT 38091 F06 MAXIMUM SENSE VOLTAGE Figure 6. Auxiliary Output Loop Connection However, if the controller goes into pulse-skipping operation and halts switching due to a light primary load current, then VAUX will droop. An external resistor divider from VAUX to the MODE sets a minimum voltage VAUX(MIN): ⎛ R6 ⎞ VAUX(MIN) = 0.4 V • ⎜ 1 + ⎟ ⎝ R5 ⎠ If VAUX drops below this value, the MODE voltage forces temporary continuous switching operation until VAUX is again above its minimum. Fault Condition: Short-Circuit and Current Limit If the LTC3809-1’s load current exceeds the short-circuit current limit (ISC), which is set by the short-circuit sense threshold (ΔVSC) and the on resistance (RDS(ON)) of bottom N-channel MOSFET, the top P-channel MOSFET is turned off and will not be turned on at the next clock cycle unless the load current decreases below ISC. In this case, the controller’s switching frequency is decreased and the output is regulated by short-circuit (current limit) protection. Figure 7. Line Regulation of VREF and Maximum Sense Voltage Low Supply Voltage Although the LTC3809-1 can function down to below 2.4V, the maximum allowable output current is reduced as VIN decreases below 3V. Figure 7 shows the amount of change as the supply is reduced down to 2.4V. Also shown is the effect on VREF. Minimum On-Time Considerations Minimum on-time, tON(MIN) is the smallest amount of time that the LTC3809-1 is capable of turning the top P-channel MOSFET on. It is determined by internal timing delays and the gate charge required to turn on the top MOSFET. Low duty cycle and high frequency applications may approach the minimum on-time limit and care should be taken to ensure that: tON(MIN) < VOUT fOSC • VIN 38091fc 18 LTC3809-1 APPLICATIONS INFORMATION If the duty cycle falls below what can be accommodated by the minimum on-time, the LTC3809-1 will begin to skip cycles (unless forced continuous mode is selected). The output voltage will continue to be regulated, but the ripple current and ripple voltage will increase. The minimum ontime for the LTC3809-1 is typically about 210ns. However, as the peak sense voltage (IL(PEAK) • RDS(ON)) decreases, the minimum on-time gradually increases up to about 260ns. This is of particular concern in forced continuous applications with low ripple current at light loads. If forced continuous mode is selected and the duty cycle falls below the minimum on time requirement, the output will be regulated by overvoltage protection. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + …) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC3809-1 circuits: 1) LTC3809-1 DC bias current, 2) MOSFET gate-charge current, 3) I2R losses and 4) transition losses. 1) The VIN (pin) current is the DC supply current, given in the Electrical Characteristics, which excludes MOSFET driver currents. VIN current results in a small loss that increases with VIN. 2) MOSFET gate-charge current results from switching the gate capacitance of the power MOSFET. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN, which is typically much larger than the DC supply current. In continuous mode, IGATECHG = f • QP. 3) I2R losses are calculated from the DC resistances of the MOSFETs, inductor and/or sense resistor. In continuous mode, the average output current flows through L but is “chopped” between the top P-channel MOSFET and the bottom N-channel MOSFET. The MOSFET RDS(ON) multiplied by duty cycle can be summed with the resistance of L to obtain I2R losses. 4) Transition losses apply to the external MOSFET and increase with higher operating frequencies and input voltages. Transition losses can be estimated from: Transition Loss = 2 • VIN2 • IO(MAX) • CRSS • f Other losses, including CIN and COUT ESR dissipative losses and inductor core losses, generally account for less than 2% total additional loss. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (ΔILOAD) • (ESR), where ESR is the effective series resistance of COUT. ΔILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. OPTI-LOOP compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. The ITH series RC-CC filter (see Functional Diagram) sets the dominant pole-zero loop compensation. The ITH external components showed in the figure on the first page of this data sheet will provide adequate compensation for most applications. The values can be modified slightly (from 0.2 to 5 times their suggested values) to optimize transient response once the final PC layout is done and the particular output capacitor type and value have been determined. The output capacitor needs to be decided upon because the various types and values determine the loop feedback factor gain and phase. An output current 38091fc 19 LTC3809-1 APPLICATIONS INFORMATION pulse of 20% to 100% of full load current having a rise time of 1μs to 10μs will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability. The gain of the loop will be increased by increasing RC and the bandwidth of the loop will be increased by decreasing CC. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Application Note 76. A second, more severe transient is caused by switching in loads with large (>1μF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25) • (CLOAD). Thus a 10μF capacitor would be require a 250μs rise time, limiting the charging current to about 200mA. Design Example As a design example, assume VIN will be operating from a maximum of 4.2V down to a minimum of 2.75V (powered by a single lithium-ion battery). Load current requirement is a maximum of 2A, but most of the time it will be in a standby mode requiring only 2mA. Efficiency at both low and high load currents is important. Burst Mode operation at light loads is desired. Output voltage is 1.8V. The IPRG pin will be left floating, so the maximum current sense threshold ΔVSENSE(MAX) is approximately 125mV. Maximum Duty Cycle = From Figure 1, SF = 82%. ΔVSENSE(MAX ) 5 RDS(ON)MAX = • 0.9 • SF • = 0.032Ω 6 IOUT(MAX ) • ρT VOUT = 65.5% VIN(MIN) A 0.032Ω P-channel MOSFET in Si7540DP is close to this value. The N-channel MOSFET in Si7540DP has 0.017Ω RDS(ON). The short-circuit current is: ISC = 90mV = 5.3A 0.017Ω So the inductor current rating should be higher than 5.3A. The LTC3809-1 operates at a frequency of 550kHz. For continuous Burst Mode operation with 600mA IRIPPLE, the required minimum inductor value is: LMIN = ⎛ 1.8 V 1.8 V ⎞ • ⎜ 1− ⎟ = 1.88μH 550kHz • 600mA ⎝ 2.75V ⎠ A 6A 2.2μH inductor works well for this application. CIN will require an RMS current rating of at least 1A at temperature. A COUT with 0.1Ω ESR will cause approximately 60mV output ripple. PC Board Layout Checklist When laying out the printed circuit board, use the following checklist to ensure proper operation of the LTC3809-1. • The power loop (input capacitor, MOSFET, inductor, output capacitor) should be as small as possible and isolated as much as possible from LTC3809-1. • Put the feedback resistors close to the VFB pins. The ITH compensation components should also be very close to the LTC3809-1. • The current sense traces should be Kelvin connections right at the P-channel MOSFET source and drain. • Keeping the switch node (SW) and the gate driver nodes (TG, BG) away from the small-signal components, especially the feedback resistors, and ITH compensation components. 38091fc 20 LTC3809-1 TYPICAL APPLICATIONS VIN 2.75V TO 8V 1 10μF MODE VIN 6 CITH 220pF RITH 15k IPRG LTC3809EDD-1 ITH TRACK/SS SW TG 9 8 MP Si7540DP L 1.5μH VOUT 2.5V (5A AT 5VIN) 4 10 2 187k BG 7 MN Si7540DP 3 59k VFB GND 11 RUN 5 COUT 150μF + 100pF L: VISHAY IHLP-2525CZ-01 COUT: SANYO 4TPB150MC 38091 F08 Figure 8. 550kHz, Synchronous DC/DC Converter with Internal Soft-Start VIN 2.75V TO 8V 10μF 1 MODE VIN 6 470pF IPRG TG LTC3809EDD-1 ITH TRACK/SS SW 9 8 MP Si3447BDV L 1.5μH 15k 4 10 10nF 2 118k BG 7 MN Si3460DV COUT 22μF x2 VOUT 1.8V 2A 3 59k VFB GND 11 RUN 5 D (OPT) 100pF L: VISHAY IHLP-2525CZ-01 D: ON SEMI MBRM120LT3 (OPTIONAL) 38091 F09 Figure 9. 550kHz, Synchronous DC/DC Converter with External Soft-Start, Ceramic Output Capacitor 38091fc 21 LTC3809-1 TYPICAL APPLICATIONS Synchronous DC/DC Converter with Output Tracking 1 VIN 2.75V TO 8V MODE VIN 6 220pF IPRG LTC3809EDD-1 ITH SW TG 9 8 MP Si7540DP L 1.5μH VOUT 1.8V (5A AT 5VIN) 10μF 15k 4 10 1.18k Vx 590Ω 118k 2 TRACK/SS BG 7 MN Si7540DP COUT 150μF + 3 59k VFB GND 11 RUN 5 100pF L: VISHAY IHLP-2525CZ-01 COUT: SANYO 4TPB150MC VOUT < Vx 38091 TA03 PACKAGE DESCRIPTION DD Package 10-Lead Plastic DFN (3mm × 3mm) (Reference LTC DWG # 05-08-1698) R = 0.115 TYP 6 0.675 0.05 0.38 10 0.10 3.50 0.05 1.65 0.05 2.15 0.05 (2 SIDES) PACKAGE OUTLINE 0.25 0.05 0.50 BSC 2.38 0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS PIN 1 TOP MARK (SEE NOTE 6) 3.00 0.10 (4 SIDES) 1.65 0.10 (2 SIDES) (DD10) DFN 1103 5 0.200 REF 0.75 0.05 2.38 0.10 (2 SIDES) 1 0.25 0.05 0.50 BSC 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 38091fc 22 LTC3809-1 PACKAGE DESCRIPTION MSE Package 10-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1664 Rev C) BOTTOM VIEW OF EXPOSED PAD OPTION 2.794 (.110 0.102 .004) 0.889 (.035 0.127 .005) 1 2.06 0.102 (.081 .004) 1.83 0.102 (.072 .004) 0.29 REF 5.23 (.206) MIN 2.083 (.082 0.102 3.20 – 3.45 .004) (.126 – .136) 0.05 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY NO MEASUREMENT PURPOSE 0.497 0.076 (.0196 .003) REF 10 0.50 0.305 0.038 (.0197) (.0120 .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT 3.00 0.102 (.118 .004) (NOTE 3) 10 9 8 7 6 4.90 0.152 (.193 .006) 0.254 (.010) GAUGE PLANE 0.53 0.152 (.021 .006) DETAIL “A” 0.18 (.007) SEATING PLANE 1.10 (.043) MAX DETAIL “A” 0 – 6 TYP 12345 3.00 0.102 (.118 .004) (NOTE 4) 0.86 (.034) REF 0.17 – 0.27 (.007 – .011) TYP NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 0.50 (.0197) BSC 0.1016 (.004 0.0508 .002) MSOP (MSE) 0908 REV C 38091fc Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 23 LTC3809-1 RELATED PARTS PART NUMBER LTC1628/LTC3728 LTC1735 LTC1773 LTC1778 LTC1872 LTC3411 LTC3412 LTC3416 LTC3418 LTC3701 LTC3708 DESCRIPTION Dual High Efficiency, 2-Phase Synchronous Step Down Controllers High Efficiency Synchronous Step-Down Controller Synchronous Step-Down Controller No RSENSE , Synchronous Step-Down Controller Constant Frequency Current Mode Step-Up Controller 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 4A, 4MHz, Monolithic Synchronous Step-Down Regulator 8A, 4MHz, Monolithic Synchronous Regulator 2-Phase, Low Input Voltage Dual Step-Down DC/DC Controller 2-Phase, No RSENSE , Dual Synchronous Controller with Output Tracking COMMENTS Constant Frequency, Standby, 5V and 3.3V LDOs, VIN to 36V, Burst Mode Operation, 16-Pin Narrow SSOP Fault Protection, , 3.5V ≤ VIN ≤ 36V 2.65V ≤ VIN ≤ 8.5V, IOUT Up to 4A, 10-Lead MSOP Current Mode Operation Without Sense Resistor, Fast Transient Response, 4V ≤ VIN ≤ 36V 2.5V ≤ VIN ≤ 9.8V, SOT-23 Package, 550kHz 95% Efficiency, VIN: 2.5V to 5.5V, VOUT = 0.8V, IQ = 60μA, ISD =