LTC3854IMSETRPBF

FeaTures n n n n n n n n n LTC3854 Small Footprint, Wide VIN Range Synchronous Step-Down DC/DC Controller DescripTion The LTC3854® is a high performance synchronous stepdown switching DC/DC controller that drives an all Nchannel synchronous power MOSFET stage. The LTC3854 features a 400kHz constant frequency current mode architecture. The LTC3854 operates from a 4.5V to 38V (40V absolute maximum) input voltage range and regulates the output voltage from 0.8V to 5.5V. The RUN/SS pin provides both soft-start and enable features. OPTI-LOOP compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. Current foldback limits MOSFET dissipation during short circuit conditions. Current foldback functions are disabled during soft-start. The LTC3854 has a minimum on-time at 75ns, making it well suited for high step-down ratios. The strong onboard MOSFET drivers allow the use of high power external MOSFETs to produce output currents up to 20A. L, LT, LTC, LTM, Linear Technology, the Linear logo and OPTI-LOOP are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5705919, 6498466, 5408150, 6222231. n n n n n Wide Operating VIN Range: 4.5V to 38V RSENSE or DCR Current Sensing ±1% 0.8V Reference Accuracy Over Temperature 400kHz Switching Frequency Dual N-channel MOSFET Synchronous Drive Very Low Dropout Operation: 97% Duty Cycle Starts Up Into Pre-Biased Output Adjustable Output Voltage Soft-Start Output Current Foldback Limiting (Disabled During Soft-Start) Output Overvoltage Protection 5V LDO for External Gate Drive OPTI-LOOP® Compensation Minimizes COUT Low Shutdown IQ: 15µA Tiny Thermally Enhanced 12-Pin 2mm × 3mm DFN and MSOP Packages applicaTions n n n n Automotive Systems Telecom Systems Industrial Equipment Distributed DC Power Systems Typical applicaTion High Efficiency Synchronous Step-Down Converter VIN TG 0.1µF LTC3854 RUN/SS 2200pF 10k 8.06k FB 42.2k SENSE– SENSE+ GND BG 100pF INTVCC 4.7µF 4.99k 92 1 2 3 4 5 LOAD CURRENT (A) 5.49k 0.22µF ITH BOOST SW 4.7µH AT 8.2mΩ DCR VOUT 5V 7A 150µF 93 VIN = 12V 6 7 3854 TA01b Efficiency and Power Loss vs Load Current 97 EFFICIENCY 2.0 47µF 50V 0.1µF VIN 6V TO 38V 96 EFFICIENCY (%) 1.6 POWER LOSS (W) 95 1.2 + 94 POWER LOSS 0.8 0.4 0 3854 TA01 3854fa  LTC3854 absoluTe MaxiMuM raTings (Note 1) Input Supply Voltage (VIN) ........................ 40V to –0.3V Top Side Driver Voltage (BOOST) .............. 46V to –0.3V Switch Voltage (SW) .................................. 40V to –5.0V INTVCC, BOOST-SW .................................... 6V to –0.3V SENSE+, SENSE– ......................................... 6V to –0.3V RUN/SS........................................................ 6V to –0.3V ITH, FB Voltages ....................................... 2.7V to –0.3V INTVCC Peak Output Current (Note 8) ....................40mA Operating Temperature Range (Notes 2, 3) ..........................................–40°C to 85°C Maximum Junction Temperature ...................... 125°C Storage Temperature Range .................. –65°C to 125°C pin conFiguraTion TOP VIEW FB 1 ITH 2 RUN/SS 3 BOOST 4 TG 5 SW 6 13 12 SENSE+ 11 SENSE– 10 VIN 9 INTVCC 8 BG 7 GND FB ITH RUN/SS BOOST TG SW 1 2 3 4 5 6 TOP VIEW 12 11 10 9 8 7 SENSE+ SENSE– VIN INTVCC BG GND 13 DDB PACKAGE 12-LEAD (3mm 2mm) PLASTIC DFN TJMAX = 125°C, θJA = 76°C/W, θJC = 10°C/W EXPOSED PAD (PIN 13) IS GND, MUST BE SOLDERED TO PCB MSE PACKAGE 12-LEAD PLASTIC MSOP TJMAX = 125°C, θJA = 40°C/W, θJC = 16°C/W EXPOSED PAD (PIN 13) IS GND, MUST BE SOLDERED TO PCB orDer inForMaTion LEAD FREE FINISH LTC3854EDDB#PBF LTC3854IDDB#PBF LTC3854EMSE#PBF LTC3854IMSE#PBF TAPE AND REEL LTC3854EDDB#TRPBF LTC3854IDDB#TRPBF LTC3854EMSE#TRPBF LTC3854IMSE#TRPBF PART MARKING* LDPC LDPC 3854 3854 PACKAGE DESCRIPTION 12-Lead (3mm × 2mm) Plastic DFN 12-Lead (3mm × 2mm) Plastic DFN 12-Lead Plastic MSOP 12-Lead Plastic MSOP TEMPERATURE RANGE –40°C to 85°C –40°C to 85°C –40°C to 85°C –40°C to 85°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3854fa  LTC3854 elecTrical characTerisTics SYMBOL VIN VFB IFB VREFLNREG VLOADREG PARAMETER Operating Input Voltage Range Regulated Feedback Voltage Feedback Current Reference Voltage Line Regulation Output Voltage Load Regulation (Note 4); ITH Voltage = 1.2V (Note 4) VIN = 6V to 38V (Note 4) (Note 4) Measured in Servo Loop; ∆ITH Voltage = 0.7V to 1.2V Measured in Servo Loop; ∆ITH Voltage = 1.2V to 2V ITH = 1.2V; Sink/Source = 5µA (Note 4) ITH = 1.2V; (Guaranteed by Design) (Note 5) RUN = 0V VIN Ramping Down; Measured at INTVCC VIN Ramping Down then Up; Measured at INTVCC Measured at FB VSENSE– = VSENSE+ = 3.3V In Dropout RUN/SS = 0V RUN/SS Pin Must be Taken Below this Value to Reset Part (or Put into Shutdown Mode) Soft-Start Mode FB = 0.7V, VSENSE– = 3.3V, VIN = 6V TG High TG Low BG High BG Low (Note 6) CLOAD = 3300pF CLOAD = 3300pF (Note 6) CLOAD = 3300pF CLOAD = 3300pF CLOAD = 3300pF Each Driver CLOAD = 3300pF Each Driver (Note 7) 6V < VIN < 38V ICC = 0 to 20mA 360 4.8 40 97 0.6 l l l l l The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN = 5V, unless otherwise noted. CONDITIONS MIN 4.5 0.792 0.8 ±5 0.002 0.1 –0.1 2.0 3 2 10 3.0 0.86 3.5 350 0.88 ±0.5 98 1.25 0.4 1.2 50 2.5 1.2 2.5 2.1 25 25 25 25 30 30 75 5.0 0.2 400 5.2 1.0 440 65 2.0 0.90 ±1 3 25 TYP MAX 38 0.808 ±50 0.02 0.5 –0.5 UNITS V V nA %/V % % mmho MHz mA µA V mV V µA % µA V V mV Ω Ω Ω Ω ns ns ns ns ns ns ns V % kHz Main Control Loop gm gmGBW IQ UVLO UVLOHYST VOVL ISENSE DFMAX IRUN/SS VRUN/SS_SD VRUN/SS_ON VSENSE(MAX) TG RUP TG RDOWN BG RUP BG RDOWN TG tr TG tf BG tr BG tf TG/BG t1D BG/TG t2D tON(MIN) VINTVCC VLDO INT Oscillator fSW Transconductance Amplifier gm Transconductance Amplifier GBW Input DC Supply Current Normal Mode Shutdown Undervoltage Lockout Undervoltage Lockout Hysteresis Feedback Overvoltage Lockout Sense Pins Source Current Maximum Duty Factor Soft-Start Charge Current Shutdown Threshold Soft-Start Threshold Maximum Current Sense Threshold TG Driver Pull-Up On Resistance TG Driver Pull-Down On Resistance BG Driver Pull-Up On Resistance BG Driver Pull-down On Resistance TG Transition Time: Rise Time Fall Time BG Transition Time: Rise Time Fall Time Top Gate Off to Bottom Gate On Delay Synchronous Switch-On Delay Time Bottom Gate Off to Top Gate On Delay Top Switch-On Delay Time Minimum On-Time Internal VCC Voltage INTVCC Load Regulation Switching Frequency INTVCC Linear Regulator 3854fa  LTC3854 elecTrical characTerisTics Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3854E is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3854I is guaranteed to meet performance specifications over the full –40°C to 85°C operating temperature range. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC3854DDB: TJ = TA + (PD • 76°C/W) LTC3854MSE: TJ = TA + (PD • 40°C/W) Note 4: The LTC3854 is tested in a feedback loop that servos VITH to a specified voltage and measures the resultant VFB. Note 5: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 6: Rise and fall times are measured using 10% and 90% levels. Delay times are measured using 50% levels. Not 100% tested in production. Note 7: The minimum on-time condition is specified for an inductor peakto-peak ripple current 40% of IMAX (see Minimum On-Time Considerations in the Applications Information section). Note 8: The LTC3854 maximum LDO current specification assumes there is no external DC load current being pulled from INTVCC pin. Typical perForMance characTerisTics Efficiency vs Output Current 100 98 96 EFFICIENCY (%) 94 92 90 88 86 (SEE FIGURE 9) 1 2 3 4 6 5 7 8 LOAD CURRENT (A) 9 10 VIN = 6V VIN = 12V EFFICIENCY (%) VIN = 24V VIN = 32V VOUT = 5V 99 98 97 IOUT = 5A 96 95 94 93 (SEE FIGURE 9) 5 10 15 25 20 VIN (V) 30 35 40 3854 G02 Efficiency vs Input Voltage VOUT = 5V IOUT = 10A 3854 G01 5A Load Step VOUT = 5V, VIN = 24V 5VOUT 200mV/DIV ILOAD 10A/DIV 5A/DIV LOAD STEP 50µs/DIV 3854 G03 Top Gate and Bottom Gate in Forced Continuous Mode Top Gate and Bottom Gate in Dropout BG BG TG 4µs/DIV 3854 G04 TG 4µs/DIV 3854 G05 3854fa  LTC3854 Typical perForMance characTerisTics Switching Waveforms at No Load VOUT = 5V TG BG IL 1µs/DIV 3854 G06 Switching Waveforms at High Duty Cycle, No Load VOUT = 2.5V TG BG IL 2µs/DIV 3854 G07 1.17 1.16 QUIESCENT CURRENT (mA) 1.15 1.14 1.13 1.12 1.11 1.10 1.09 Quiescent Current vs VIN Quiescent Current vs Temperature 1.30 1.25 QUIESCENT CURRENT (mA) 1.20 INTVCC (V) 1.15 1.10 1.05 1.00 0.95 5.00 5.05 INTVCC vs Input Voltage 4.95 4.90 4.85 6 11 16 26 21 VIN (V) 31 36 3854 G08 0.90 –45 –25 –5 15 35 55 75 TEMPERATURE (°C) 95 3854 G09 4.80 4 9 14 19 24 VIN (V) 29 34 39 3854 G10 Maximum Current Sense Threshold vs Sense Common Mode Voltage 60 58 56 VSENSE(MAX) (mV) VSENSE(MAX) (mV) 54 52 50 48 46 44 42 40 0 1 3 5 2 4 6 COMMON MODE VOLTAGE (V) 7 3854 G11 Maximum Current Sense Threshold vs Duty Cycle MAXIMUM CURRENT SENSE VOLTAGE (mV) 65 60 55 50 45 40 35 60 50 40 30 20 10 0 Maximum Current Sense Voltage vs Feedback Voltage (Current Foldback) 0 20 60 40 80 DUTY CYCLE (%) 100 120 3854 G12 0 0.05 0.15 0.25 0.35 0.45 0.55 0.65 0.75 FB (V) 3854 G13 3854fa  LTC3854 Typical perForMance characTerisTics Oscillator Frequency vs Temperature 440 430 FREQUENCY (kHz) 420 410 400 390 380 –40 –20 UVLO (V) 4.0 3.9 3.8 FREQUENCY (kHz) 3.7 3.6 3.5 3.4 3.3 3.2 3.1 0 20 40 60 80 TEMPERATURE (°C) 100 120 3854 G15 Undervoltage Lockout Falling Threshold vs Temperature 440 430 420 410 400 390 380 370 0 20 40 60 80 TEMPERATURE (°C) 100 120 3854 G16 Oscillator Frequency vs VIN 3.0 –40 –20 360 5 10 15 25 20 VIN (V) 30 35 40 3854 G17 Current Sense Threshold vs ITH Voltage 60 CURRENT SENSE THRESHOLD (mV) 50 40 30 20 10 0 –10 0 0.5 1 1.5 VITH (V) 2 2.5 3 3854 G18 Shutdown Current vs Input Voltage 30 25 SHUTDOWN CURRENT (µA) SHUTDOWN CURRENT (µA) 20 15 10 5 0 20 19 18 17 16 15 14 13 12 11 4 8 12 16 20 24 VIN (V) 28 32 36 40 Shutdown Current vs Temperature VIN = 15V 10 –45 –30 –15 0 15 30 45 60 75 90 105 TEMPERATURE (°C) 3854 G20 3854 G19 5.20 5.15 5.10 INTVCC (V) INTVCC Load Regulation VIN = 15V 5.20 INTVCC vs Temperature RUN/SS Shutdown Threshold vs Temperature 1.40 1.35 1.30 RUN/SS (V) 1.25 1.20 1.15 1.10 –45 –30 –15 0 15 30 45 60 75 90 105 TEMPERATURE (°C) 3854 G23 ILOAD = 10mA 5.15 VIN = 15V 5.10 INTVCC (V) 5.05 5.00 4.95 4.90 4.85 5.05 5.00 4.95 4.90 4.85 4.80 0 5 10 15 20 25 30 LDO CURRENT (mA) 35 40 4.80 –45 –30 –15 0 15 30 45 60 75 90 105 TEMPERATURE (°C) 3854 G22 3854 G21 3854fa  LTC3854 pin FuncTions FB (Pin 1): Error Amplifier Feedback Input. This pin receives the remotely-sensed feedback voltage from an external resistor divider across the output. ITH (Pin 2): Error Amplifier Output and Switching Regulator Compensation Point. The current comparator trip point increases with this control voltage. RUN/SS (Pin 3): Run Control, Soft-Start. If the voltage on this pin is held below 0.4V, the part is in shutdown. If the pin is released the capacitance to ground at this pin sets the soft-start ramp rate. An internal 1.25µA soft-start current is always charging this pin. BOOST (Pin 4): Bootstrapped Supply to the Top Side Floating Driver. A low ESR capacitor is connected between the BOOST and SW pins and an external Schottky diode is tied between the BOOST and INTVCC pins. The voltage swing on the BOOST pin is INTVCC to (VIN + INTVCC). TG (Pin 5): High Current Gate Drive for Top N-channel MOSFET. This is the output of floating driver with a voltage swing equal to INTVCC superimposed on the switch node voltage. SW (Pin 6): Switch Node Connection to Inductor. The voltage swing on this pin is from a Schottky diode (external) forward voltage (when this diode is added across the Nchannel synchronous MOSFET) below ground to VIN. GND (Pin 7): Small Signal and Power Ground. This is the high current ground for the gate driver. The internal signal ground is Kelvin connected to this pin for noise suppression. BG (Pin 8): High Current Gate Drive for Bottom (Synchronous) N-channel MOSFET. The voltage swing at this pin is from ground to INTVCC. INTVCC (Pin 9): Output of the Internal 5V Low Dropout Regulator. The driver and control circuits are powered from this voltage. Must be decoupled to power ground with a minimum of 2.2µF low ESR ceramic capacitor (X5R or better). VIN (Pin 10): Main Supply Pin. A bypass capacitor should be tied between this pin and the signal ground pin. SENSE– (Pin 11): The (–) Input to the Differential Current Comparator. SENSE+ (Pin 12): The (+) Input to the Differential Current Comparator. The ITH pin voltage and controlled offsets between the SENSE– and SENSE+ pins in conjunction with RSENSE (or RDCR) set the peak current trip threshold. SGND (Exposed Pad Pin 13): The exposed pad must be soldered to PCB ground for electrical contact and rated thermal performance. 3854fa  LTC3854 FuncTional DiagraM VIN 1.2V 0.88V 0.80V BANDGAP/ LDO INTVCC BOOST OSCILLATOR INTVCC UVLO DRIVE CONTROL AND ANTI-SHOOT THROUGH ON CLK SQ R SLOPE COMP GENERATOR CURRENT SENSE COMPARATOR TG_OFF OV MAIN SWITCH DRIVER TG SW SYNCHRONOUS SWITCH DRIVER BG GND + – + RUN 0.88V VFB SENSE– SENSE+ V TO I CONVERTER VSENSE ICS + ICMP – + SS – 1.2V – SD ITH ERROR AMP VITH  – + + + – 0.4V 1.25µA LEVEL SHIFT 0.8V RUN/SS ITH 3854 FD 3854fa LTC3854 operaTion Main Control Loop The LTC3854 is a constant-frequency, peak current mode step-down controller. During normal operation, the top MOSFET is turned on when the clock sets the RS latch, and is turned off when the main current comparator, ICMP, resets the RS latch. The peak inductor current at which ICMP resets the RS latch is controlled by the voltage on the ITH pin, which is the output of the error amplifier EA. The VFB pin receives the voltage feedback signal, which is compared to the internal reference voltage by the EA. When the load current increases, it causes a slight decrease in VFB relative to the 0.8V reference, which in turn causes the ITH voltage to increase until the average inductor current matches the new load current. After the top MOSFET has turned off, the bottom MOSFET is turned on until the beginning of the next cycle. INTVCC Power Power for the top and bottom MOSFET drivers and most other internal circuitry is derived from the INTVCC pin. An internal 5V low dropout linear regulator supplies INTVCC power from VIN. The top MOSFET driver is biased from a floating bootstrap capacitor CB , which recharges during each off cycle through an external Schottky diode when the top MOSFET turns off. If the input voltage VIN decreases to a voltage close to VOUT, the loop may enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector then forces the top MOSFET off for 1/10 of the clock period every fourth cycle to allow CB to recharge. Shutdown and Start-Up (RUN/SS) The LTC3854 is shut down using the RUN/SS pin. Pulling this pin below 1.2V disables the controller and most of the internal circuitry, including the INTVCC regulator. However, for RUN/SS>0.8V the internal bandgap is functional and the input current will be greater than the minimum shutdown current. To keep the part in a true shutdown mode the RUN/SS pin should be held below 0.4V. Releasing RUN/SS pin allows an internal 1.25µA current to pull up the pin and enable the controller. Alternatively, the RUN/SS pin may be externally pulled up or driven directly by logic. Be careful not to exceed the Absolute Maximum Rating of 6V on this pin. The start-up of the controller’s output voltage VOUT is governed by the voltage on the RUN/SS pin until RUN/SS > 2V. When the voltage on the RUN/SS pin is greater than 1.2V and less than 2V the LTC3854 regulates the VFB voltage to 1.2V below the RUN/SS pin voltage. The RUN/SS pin programs the soft-start period through an external capacitor from the RUN/SS pin to GND. An internal 1.25µA pull-up current charges this capacitor creating a voltage ramp on the RUN/SS pin. As the RUN/SS voltage rises linearly from 1.2V to 2V, VOUT rises smoothly from zero to the target output voltage. When the LTC3854 is in undervoltage lockout the external MOSFETs are held off. Frequency of Operation The LTC3854 operates at a fixed frequency of 400kHz. Output Overvoltage Protection An overvoltage comparator, OV, guards against transient overshoots (>10%) as well as other more serious conditions that may overvoltage the output. In such cases, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. 3854fa  LTC3854 applicaTions inForMaTion The LTC3854 can be configured to use either DCR (inductor winding resistance) sensing or low value resistor sensing. The choice of the two current sensing schemes is largely a design tradeoff between cost, power consumption, and accuracy. DCR sensing is becoming popular because it eliminates expensive current sensing resistors and is more power efficient, especially in high current applications. However, current sensing resistors provide the most accurate current limits for the controller. Other external component selection is driven by the load requirement, and begins with the selection of RSENSE (if RSENSE is used) and inductor value. Next, the power MOSFETs and Schottky diodes are selected. Finally, input and output capacitors are selected. The Typical Application shown on the first page can be configured for operation up to 38V on VIN. SENSE+ and SENSE– Pins The SENSE+ and SENSE– pins are the inputs to the current comparator. The common mode input voltage range of the current comparator is 0V to 5.5V. Both SENSE pins are high impedance inputs with small input bias currents of less than 1μA. When the SENSE pins ramp up from 0V to 1.4V, small bias currents flow out of the SENSE pins. When the SENSE pins ramp down from 5.5V to 1.1V, the small bias currents flow into the SENSE pins. The high impedance inputs to the current comparator allow accurate DCR sensing. VIN INTVCC BOOST LTC3854 TG SW BG GND SENSE+ SENSE– RSENSE VOUT Using a Sense Resistor for Current Sensing A typical sensing circuit using a discrete resistor is shown in Figure 1. RSENSE is chosen based on the required output current. The current comparator has a maximum threshold of 50mV. The input common mode range of the current comparator is 0V to 5.5V. The current comparator threshold sets the peak of the inductor current, yielding a maximum average output current IMAX equal to the peak value less half the peak-to-peak ripple current, ∆IL. Allowing a margin of 20% for variations in the IC and external component values yields: VSENSE(MAX) ∆I IMAX + L 2 Inductor DCR Sensing RSENSE = 0.8 • For applications requiring the highest possible efficiency, the LTC3854 is capable of sensing the voltage drop across the inductor DCR, as shown in Figure 2. The DCR of the inductor represents the small amount of DC copper winding resistance, which can be less than 1mΩ for today’s low value, high current inductors. When the external R1||R2•C1 time constant is chosen to be equal to the L/DCR time constant, the voltage drop across the external FILTER COMPONENTS PLACED NEAR SENSE PINS 3854 F01 Figure 1. Using a Resistor to Sense Current with the LTC3854 3854fa 0 LTC3854 applicaTions inForMaTion VIN VIN RUN/SS CSS CC CC2 RC RFB1 RFB2 TG CB ITH BOOST LTC3854 SW INTVCC CVINT BG GND M2 R1 C1 DBOOST INDUCTOR L DCR VOUT COUT M1 CIN FB SENSE– SENSE+ R2 3854 F02 Figure 2. Buck Regulator Using DCR Current Sense capacitor is equal to the voltage drop across the inductor DCR • R2/(R1+R2). R2 may be used to scale the voltage across the same terminals when the DCR is greater than the target sense resistance. Check the manufacturer’s datasheet for specifications regarding the inductor DCR, in order to properly dimension the external filter components. The DCR of the inductor can also be measured using a precision RLC meter. Slope Compensation and Inductor Peak Current Slope compensation provides stability in constant-frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor current signal. Normally, this results in a reduction of maximum inductor peak current for high duty cycles. However, the LTC3854 uses a novel scheme that allows the maximum inductor peak current to remain unaffected throughout all duty cycles. Inductor Value Calculation The inductor value has a direct effect on ripple current. The inductor ripple current ∆IL decreases with higher inductance or frequency and increases with higher VIN. LMIN =   V 1 • VOUT  1− OUT  ∆IL • fSW  VIN(MAX)  Accepting larger values of ∆IL allows the use of low value inductors, but results in a higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is IL = 0.4 • (IMAX). The maximum ∆IL occurs at the maximum input voltage. Option 1: DCR within desired range R1• C1= L (R2 not used) DCR L (at 20°C) DCR Option 2: DCR > desired RSENSE R1||R2 • C1 = RSENSE (EQ) = DCR(MAX) • R2 R1+ R2 3854fa  LTC3854 applicaTions inForMaTion Inductor Core Selection Once the value for L is determined, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite or molypermalloy cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses decrease. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred at high switching frequencies; allowing design goals to concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Power MOSFET and Schottky Diode (Optional) Selection Two external power MOSFETs must be selected for the LTC3854 controller: one N-channel MOSFET for the top (main) switch, and one N-channel MOSFET for the bottom (synchronous) switch. The peak-to-peak drive levels are set by the INTVCC voltage. This voltage is 5V during start-up. Consequently, logiclevel threshold MOSFETs can be used in most applications. The only exception is if low input voltage is expected (VIN < 5V); then, sub-logic level threshold MOSFETs (VGS(TH) < 3V) should be used. Pay close attention to the BVDSS specification for the MOSFETs as well; most of the logic level MOSFETs are limited to 30V or less. Selection criteria for the power MOSFETs include the on-resistance RDS(ON), Miller capacitance CMILLER, input voltage and maximum output current. Miller capacitance, CMILLER, can be approximated from the gate charge curve usually provided on the MOSFET manufacturers’ data sheet. CMILLER is equal to the increase in gate charge along the horizontal axis while the curve is approximately flat divided by the specified change in VDS. This result is then multiplied by the ratio of the applied VDS to the gate charge curve specified VDS. When the IC is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: Main Switch Duty Cycle = VOUT =D VIN VIN − VOUT = 1− D VIN Synchronous Switch Duty Cycle = The MOSFET power dissipations at maximum output current are given by: PMAIN = VOUT (IMAX )2 (1+ δ ) RDS(ON) + VIN ( VIN )2  IMAX  (RDR )(CMILLER ) • 2    1 1 +   (f)  VINTVCC − VTH(MIN) VTH(MIN)    V −V 2 PSYNC = IN OUT (IMAX ) (1+ δ ) RDS(ON) VIN where δ is the temperature dependency of RDS(ON) and RDR (approximately 2Ω) is the effective driver resistance at the MOSFET’s Miller threshold voltage. VTH(MIN) is the typical MOSFET minimum threshold voltage. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For VIN < 20V, the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during short-circuit when the synchronous switch is on close to 100% of the period. The term (1 + δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs Temperature curve, but δ = 0.005/°C can be used as an approximation for low voltage MOSFETs. 3854fa  LTC3854 applicaTions inForMaTion An optional Schottky diode connected from GND (anode) to the SW node (cathode) conducts during the dead time between the conduction of the two power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on, storing charge during the dead time and requiring a reverse recovery period that could cost as much as 3% in efficiency at high VIN. A 1A to 3A Schottky is generally a good size due to the relatively small average current. Larger diodes result in additional transition losses due to their larger junction capacitance. Soft-Start When the LTC3854 is configured to soft-start by itself, a capacitor must be connected to the RUN/SS pin. The LTC3854 is in the shutdown state if the RUN/SS pin voltage is below 1.2V. The RUN/SS pin has an internal 1.25µA pull-up current and should be externally pulled low ( 1 8fSW RSENSE The maximum RMS capacitor current is: IRMS = This formula has a maximum at VIN = 2 •VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturers’ ripple The first condition relates to the ripple current into the ESR of the output capacitance while the second term guarantees that the output capacitance does not significantly discharge during the operating frequency period due to ripple current. The choice of smaller output capacitance increases the ripple voltage due to the discharging term but can be compensated with capacitors of very low ESR to maintain the ripple voltage at or below 50mV. The ITH pin OPTI-LOOP compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected. The selection of output capacitors for applications with large load current transients is primarily determined by the voltage tolerance specifications of the load. The resistive component of the capacitor, ESR, multiplied by the load current change plus any output voltage ripple must be within the voltage tolerance of the load. 3854fa  LTC3854 applicaTions inForMaTion The required ESR due to a load current step is: ESR ≤ ∆V ∆I in cost-driven applications, provided that consideration is given to ripple current ratings, temperature and long-term reliability. A typical application will require several aluminum electrolytic capacitors in parallel. A combination of the above mentioned capacitors will often result in maximizing performance and minimizing overall cost. Other capacitor types include Nichicon PL series, NEC Neocap, Panasonic SP and Sprague 595D series. Consult manufacturers for other specific recommendations. Like all components, capacitors are not ideal. Each capacitor has its own benefits and limitations. Combinations of different capacitor types have proven to be a very cost effective solution. Remember also to include high frequency decoupling capacitors. They should be placed as close as possible to the power pins of the load. Any inductance present in the circuit board traces negates their usefulness. Setting Output Voltage The LTC3854 output voltage is set by an external feedback resistive divider carefully placed across the output, as shown in Figure 3. The regulated output voltage is determined by:  R VOUT = 0.8  1+ B   RA  VOUT LTC3854 FB RA 3854 F03 where ∆I is the change in current from full load to zero load (or minimum load) and ∆V is the allowed voltage deviation (not including any droop due to finite capacitance). The amount of capacitance needed is determined by the maximum energy stored in the inductor. The capacitance must be sufficient to absorb the change in inductor current when a high current to low current transition occurs. The opposite load current transition is generally determined by the control loop OPTI-LOOP components, so make sure not to over compensate and slow down the response. The minimum capacitance to assure the inductors’ energy is adequately absorbed is: L ( ∆I) COUT > 2 ( ∆V ) VOUT 2 Manufacturers such as Nichicon, United Chemi-Con and Sanyo should be considered for high performance through-hole capacitors. The OS-CON semiconductor electrolyte capacitor available from Sanyo has the lowest (ESR)(size) product of any aluminum electrolytic at a somewhat higher price. An additional ceramic capacitor in parallel with OS-CON capacitors is recommended to reduce the inductive effects. In surface mount applications, ESR, RMS current handling and load step specifications may require multiple capacitors in parallel. Aluminum electrolytic, dry tantalum and special polymer capacitors are available in surface mount packages. Special polymer surface mount capacitors offer very low ESR but have much lower capacitive density per unit volume than other capacitor types. These capacitors offer a very cost-effective output capacitor solution and are an ideal choice when combined with a controller having high loop bandwidth. Tantalum capacitors offer the highest capacitance density and are often used as output capacitors for switching regulators having controlled soft-start. Several excellent surge-tested choices are the AVX TPS, AVX TPSV or the KEMET T51 0 series of surface mount tantalums, available in case heights ranging from 1.5mm to 4.1mm. Aluminum electrolytic capacitors can be used RB CFF Figure 3. Feed-Forward Capacitor on FB Pin To improve the frequency response, a feed-forward capacitor, CFF, may be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. Fault Conditions: Current Foldback The LTC3854 includes current foldback to help limit load current when the output is shorted to ground. If the output falls below 40% of its nominal output level, the maximum 3854fa  LTC3854 applicaTions inForMaTion sense voltage is progressively lowered from its maximum programmed value to 25% of the maximum value. Foldback current limiting is disabled during soft-start. Minimum and Maximum On-Time Considerations Minimum on-time tON(MIN) is the smallest time duration that the LTC3854 is capable of turning on the top MOSFET. It is determined by internal timing delays and the gate charge required to turn on the top MOSFET. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that VOUT >t VIN • fSW ON(MIN) If the duty cycle falls below what can be accommodated by the minimum on-time, the controller will begin to skip cycles. The output voltage will continue to be regulated, but the ripple voltage and current will increase. The minimum on-time for the LTC3854 is approximately 75ns. However, as the peak sense voltage decreases the minimum on-time gradually increases. This is of particular concern in forced continuous applications with low ripple current at light loads. If the duty cycle drops below the minimum on-time limit in this situation, a significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple. Care should also be taken for applications where the duty cycle can approach the maximum given in the data sheet (98%). In all low dropout applications, such as VOUT = 5V and VIN(MIN) = 4.5V, careful selection of the bottom synchronous MOSFET is required. For applications where the input voltage can drop below the targeted output voltage, and subsequently ramp up, a low threshold synchronous MOSFET with a small total gate charge should be chosen. This selection for the bottom synchronous MOSFET will insure that the bottom gate minimum on-time is sufficient in dropout to allow for the initial boost capacitor refresh that is needed to adequately turn on the top side driver and begin the switching cycle. Another method to guarantee performance in this type of application is to increase the minimum output load to 50mA. This minimum load will allow the user to choose larger MOSFETs for delivery of large currents when VIN is in the normal operating range yet still provide an adequate safety margin and good overall performance in dropout with a slow ramping VIN. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC3854 circuits: 1) IC VIN current, 2) INTVCC regulator current, 3) I2R losses, 4) Topside MOSFET transition losses. 1. The VIN current is the DC supply current given in the Electrical Characteristics table, which excludes MOSFET driver and control currents. VIN current typically results in a small (1μF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can alter its delivery of current quickly enough to prevent this 3854fa  LTC3854 applicaTions inForMaTion sudden step change in output voltage if the load switch resistance is low and it is driven quickly. If the ratio of CLOAD to COUT is greater than 1:50, the switch rise time should be controlled so that the load rise time is limited to approximately 25 • CLOAD. Thus a 10μF capacitor would require a 250μs rise time, limiting the charging current to about 200mA. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3854. These items are also illustrated graphically in the layout diagram of Figure 4. Check the following in your layout: 1) Are the signal and power grounds segregated? The LTC3854 GND pin should tie to the ground plane close to the output capacitor(s). The low current or signal ground trace should make a single point connection directly to the GND pin. The synchronous MOSFET source pins should connect to the input capacitor(s) ground. 2) Does the VFB pin connect directly to the feedback resistors? The resistive divider R1, R2 must be connected between the (+) plate of COUT and signal ground. The 47pF to 100pF capacitor should be as close as possible to the LTC3854. Be careful locating the feedback resistors too far away from the LTC3854. The VFB line should not be routed close to any other nodes with high slew rates. + SW LTC3854 TG CSS CC CC2 3 2 1 11 1000pF 12 RUN/SS ITH VFB SENSE– SENSE+ BOOST VIN INTVCC BG GND 6 5 4 VIN 10 9 8 7 DB CB D1 M2 CIN RC 47pF + 4.7µF + M1 – L1 – R1 COUT RSENSE VOUT R2 Figure 4. LTC3854 Layout Diagram  + 3854 F04 + 3854fa LTC3854 applicaTions inForMaTion 3) Are the SENSE– and SENSE+ leads routed together with minimum PC trace spacing? The filter capacitor between SENSE+ and SENSE– should be as close as possible to the LTC3854. Ensure accurate current sensing with Kelvin connections as shown in Figure 5. Series resistance can be added to the SENSE lines to increase noise rejection. 4) Does the (+) terminal of CIN connect to the drain of the topside MOSFET(s) as closely as possible? This capacitor provides the AC current to the MOSFET(s). 5) Is the INTVCC decoupling capacitor connected closely between INTVCC and GND? This capacitor carries the MOSFET driver peak currents. 6) Keep the switching node (SW), top gate node (TG), bottom gate node (BG) and boost node (BOOST) away from sensitive small-signal nodes, especially from the voltage and current sensing feedback pins. All of these nodes have very large and fast moving signals and therefore should be kept on the “output side” (Pins 4,5,6 and 8) of the LTC3854 GND and occupy minimum PC trace area. PC Board Layout Debugging It is helpful to use a DC-50MHz current probe to monitor the current in the inductor while testing the circuit. Monitor the output switching node (SW pin) to synchronize the oscilloscope to the internal oscillator and probe the actual output voltage as well. Check for proper performance over the operating voltage and current range expected in the application. The frequency of operation should be maintained over the input voltage range down to dropout. The duty cycle percentage should be maintained from cycle to cycle in a well-designed, low noise PCB implementation. Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs or inadequate loop compensation. Overcompensation of the loop can be used to tame a poor PC layout if regulator bandwidth optimization is not required. A 1Ω to 10Ω boost resistor may help to improve noise immunity. This resistor is placed between the BOOST pin and the node formed by the cathode of the boost Schottky and the positive terminal of the boost capacitor. Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems coincide with high input voltages and low output currents, look for capacitive coupling between the BOOST, SW, TG, and possibly BG connections and the sensitive voltage and current pins. The capacitor placed across the current sensing pins needs to be placed immediately adjacent to the pins of the IC. This capacitor helps to minimize the effects of differential noise injection due to high frequency capacitive coupling. If problems are encountered with high current output loading at lower input voltages, look for inductive coupling between CIN, Schottky and the top MOSFET components to the sensitive current and voltage sensing traces. In addition, investigate common ground path voltage pickup between these components and the GND pin of the IC. CURRENT SENSE RESISTOR (RSENSE) SENSE+ SENSE– 3854 F04 Figure 5. Kelvin Sensing RSENSE 3854fa  LTC3854 applicaTions inForMaTion Design Example Consider the design of a 1.2V, 15A buck regulator with a VIN range of 4.5V to 28V using a DCR sensing scheme. Inductor Selection Assuming an inductor ripple of 40% of IOUT, L can be calculated for the worst case of VIN = VIN(MAX). LMIN = LMIN =   V 1 • VOUT  1− OUT  ∆IL • fSW  VIN(MAX)  also help insure the minimum on-time requirement of 75ns is not violated). tON(MIN) = tON(MIN) = VOUT VIN(MAX) • fSW 1.2V 20V • 400kHz tON(MIN) = 150ns To choose R1 for DCR sensing we use: R1• C1= L at 25°C DCR 1  1.2V  • 1.2V •  1−   0.40 • 15A • 400kHz 20V  LMIN = 0.47µH Next, determine the DCR of the inductor. When provided, use the manufacturer’s maximum value, usually given at 25°C. Increase this value to account for the temperature coefficient of resistance, which is approximately 0.4%/°C. A conservative value for TLMAX is 100°C which corresponds to a delta of 0.3. To allow the converter to source 15A with an inductor temperature of 100°C without hitting maximum current limit we need a DCR at 25°C of: DCR(25°C) = 0.8 • VSENSE(MAX) ∆IL    IMAX + 2  • (1+ δ)   0.8 • 50mV DCR(25°C) = 15A • 0.4    15A +  • (1+ 0.3)   2 DCR(25°C) = 1.7mΩ The 0.56µH inductor from the IHLP4040DZ-01 series has a typical DCR of 1.7mΩ and a maximum of 1.8mΩ and as ISAT of 49A. The saturation current is well above our operating current maximum. The maximum inductor will be the DC value plus one half the ripple current. Using this inductor gives an inductor ripple current of 6A (keeping the ripple current high will Choosing C1 = 100nF and using the maximum DCR value at 25°C, we get: 0.56µH 1.8mΩ • 100nF R1= 3.11k R1= Choose 3.09k. Output Capacitor Selection The output voltage AC ripple due to capacitive impedance and ESR in normal continuous mode operation can be calculated from:   1 ∆VOUT = ∆IL  ESR + 8 • fSW • COUT    The second term is the AC capacitive impedance part of the above equation and used alone will yield a minimum COUT of: COUT > COUT > ∆IL 8 • fSW • ∆VOUT 0.4 • 15A 8 • 400kHz • 0.01• 1.2V COUT > 156µF 3854fa 0 LTC3854 applicaTions inForMaTion However, the amount of capacitance needed is determined not only by the allowed ripple in steady state but by the maximum energy stored in the inductor. The capacitance must be sufficient in value to absorb the change in inductor current when a high current to low current transient occurs. The minimum capacitance to assure the inductor’s energy is adequately absorbed during a 5A load step for a maximum overshoot of 2% is: L • ∆IL 2 COUT ≥ 2 • ∆VOUT • VOUT 0.56µH • (5A)2 0.02 • 1.2V COUT ≥ 583µF COUT ≥ A maximum overshoot or undershoot of 2% for a 5A load step will require an ESR of: ESR < 0.02 • VOUT 1.2V = 0.02 • ≤ 5mΩ ∆ILOAD 5A Choosing CIN Capacitors CIN is chosen for a RMS current rating of at least IOUT(MAX)/2 = 6A. Again, keeping ESR low will improve performance and reduce power loss (several capacitors in parallel is once again a good choice). We will use an 180µF 25V electrolytic with 2x 10µF 25V low ESR ceramic capacitors connected in parallel. Choosing MOSFETs The power dissipation in the main and synchronous FETs can be easily estimated. Choosing a Renesas RJK0305DPB for the main FET results in the following parameters: BVDSS = 30V RDS(ON) = 13mΩ maximum at 25°C, VGS = 4.5V QGD = 1.5nC at VDS, test 10V results in CMILLER = 1.5nC/10V = 150pF QG = 8nC, typical, at VGS = 4.5V VMILLER = 2.8V At VIN = 20V, IOUT = 15A, estimated TJ = 100°C for the top FET and given VINTVCC = 5.0V RDR,PULLUP = 2.6Ω RDR,PULLDOWN = 1.5Ω the total losses in the main FET will be: PMAIN = 1.2V • (15A)2 • 1+ 0.005 • (100°C – 25°C) 20V 2 15A • 13mΩ + ( 20V ) • • 150pF 2 1.2Ω   2.5Ω • + •f  5V − 2.8V 2.8V  SW  Several quality capacitors are available with low enough ESR. Multilayer ceramic capacitors tend to have very low ESR values. It is also a good practice to reduce the ESL by putting several capacitors in parallel on the output (a parallel bank of larger and smaller capacitors will improve performance in both a DC and a transient condition). To keep ripple very low and design for any possible large excursions in current 2x 330µF (tantalum or polymer surface) and 1x 47µF polymer low ESR type were connected in parallel. Choosing FB Resistors (See Figure 3)  R VOUT = 0.8  1+ B   RA  RB = 0.5R A Using 1% 10.0k for RA gives 1% 4.99k for RB. ( ) PMAIN = 0.55W 3854fa  LTC3854 applicaTions inForMaTion Choosing an RJK0330DPB for the bottom FET will provide: BVDSS = 30V RDS(ON) = 3.9mΩ maximum at 25°C, VGS = 4.5V QG = 27nC, typical, at VGS = 4.5V PSYNC = 20V − 1.2V 2 • (15A ) 20V • (1+ 0.005 • (100°C – 25°C)) • 3.9mΩ Some airflow may be required for higher ambient temperatures. A maximum MOSFET junction temperature of 110°C at worst case ambient generally provides adequate margin. Given a typical QG of 8nC for the RJK0305DPB and 27nC for the RJK0330DPB and the 400kHz switching frequency, the current supplied by INTVCC will be: IGATECHG = (8nC + 27nC) • 400kHz = 14mA The resulting controller temperature at 60°C and a 20V input will be: TJ = 60°C + 20V • 14mA • 76°C/W = 81°C which is well under the maximum junction temperature of 125°C. VIN 4.5V TO 28V 10µF 25V 2 VOUT 1.2V 15A COUT1 330µF 2 PSYNC = 1.1W Assuming a thermal resistance of 40°C/W for the main and synchronous FETs, the resulting junction temperatures at an ambient of 60°C will be 82°C and 104°C, respectively. VIN 0.1µF LTC3854 TG RUN/SS 2200pF 3.9k 10k FB 4.99k SENSE– BG 100pF ITH BOOST SW M1 0.1µF L 0.56µH 1.8m DCR 3.09k 180µF 25V 0.1µF D1 INTVCC 4.7µF + M2 COUT2 47µF SENSE+ GND D1 = CMDSH-3 M1 = RENESAS RJK0305DPB M2 = RENESAS RJK0330DPB L = VISHAY IHLP4040DZ-01 0.56µH COUT1 = SANYO 2R5TPE330M9 COUT2 = MURATA GRM31CR60J476K 3854 F06 Figure 6. 1.2V/15A Converter from Design Example 3854fa  LTC3854 Typical applicaTions VIN 0.1µF LTC3854 TG RUN/SS 2200pF 3.9k 20k FB 17.4k 1nF SENSE+ 47 47 3854 TA02 M1 0.1µF 180µF 25V VIN 4.5V TO 14V 47µF 25V 2 ITH 100pF BOOST SW D1 INTVCC 4.7µF BG M2 COUT2 47µF 2 L 0.56µH RSENSE 2m VOUT 1.5V 15A COUT1 330µF 2 + SENSE– GND D1 = CMDSH-3 M1 = RENESAS RJK0305DPB M2 = RENESAS RJK0330DPB L = TOKO FDA1055-R56M COUT1 = SANYO 2R5TPE330M9 COUT2 = MURATA GRM31CR60J476K Figure 7. 1.5V/15A RSENSE Application 3854fa  LTC3854 Typical applicaTions VIN TG 0.1µF LTC3854 RUN/SS 2200pF 15k 32.4k FB 8.06k SENSE– SENSE+ GND BG 100pF INTVCC 4.7µF M2 ITH BOOST SW D1 0.1µF M1 180µF 25V L 0.22µH, 1.8m DCR 0.1µF VIN 4.5V TO 14V 47µF 25V 2 VOUT 1V 20A COUT1 330µF 2 1.21k + COUT2 47µF 2 D1 = CMDSH-3 M1, M2 = VISHAY Si7866ADP L = SUMIDA CDEP104NP-0R2NC COUT1 = SANYO 2RSTPE330M9 COUT2 = MURATA GRM31CR60J476K 3854 TA03 Figure 8. 1.0V/20A DCR Sense Application VIN 0.1µF LTC3854 TG RUN/SS 2200pF 7.5k 8.06k FB 42.2k 4.7nF SENSE+ 16 16 GND SENSE– BG 100pF ITH BOOST SW D1 INTVCC 4.7µF M2 L 3.6µH 0.1µF M1 150µF 50V VIN 6V TO 38V 47µF 50V 2 RSENSE 3m + VOUT 5V 10A COUT1 220µF D1 = ZETEX ZLLS1000 M1, M2 = INFINEON BSC093N04LS L = COILTRONICS HC1-3R6-R COUT1 = SANYO 6TPE220MI 3854 TA04 Figure 9. 5.0V/10A RSENSE Application 3854fa  LTC3854 package DescripTion (Reference LTC DWG # 05-08-1723 Rev Ø) 0.64 ± 0.05 (2 SIDES) 0.70 ± 0.05 2.55 ± 0.05 1.15 ± 0.05 PACKAGE OUTLINE 0.25 ± 0.05 2.39 ± 0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 3.00 ± 0.10 (2 SIDES) R = 0.05 TYP 2.00 ± 0.10 (2 SIDES) 0.64 ± 0.10 (2 SIDES) 6 0.23 ± 0.05 2.39 ± 0.10 (2 SIDES) BOTTOM VIEW—EXPOSED PAD R = 0.115 TYP 7 0.40 ± 0.10 12 0.45 BSC DDB Package 12-Lead Plastic DFN PIN 1 BAR TOP MARK (SEE NOTE 6) 0.200 REF 1 PIN 1 R = 0.20 OR 0.25 × 45° CHAMFER (DDB12) DFN 0106 REV Ø 0.75 ± 0.05 0.45 BSC 0 – 0.05 NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3854fa  LTC3854 package DescripTion MSE MSE Package Package 12-Lead Plastic MSOP Exposed Pad , 12-Lead Plastic MSOP, Exposed Die Die Pad (Reference LTC # 05-08-1666 Rev B) (Reference LTC DWG DWG # 05-08-1666 Rev B) BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 (.112 0.102 .004) 0.889 (.035 0.127 .005) 2.845 (.112 1 0.102 .004) 6 0.35 REF 5.23 (.206) MIN 1.651 (.065 0.102 3.20 – 3.45 .004) (.126 – .136) 0.12 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY 7 NO MEASUREMENT PURPOSE 0.406 0.076 (.016 .003) REF 12 0.65 0.42 0.038 (.0256) (.0165 .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT DETAIL “A” 0 – 6 TYP 4.039 0.102 (.159 .004) (NOTE 3) 12 11 10 9 8 7 0.254 (.010) GAUGE PLANE 4.90 0.152 (.193 .006) 3.00 0.102 (.118 .004) (NOTE 4) 0.53 0.152 (.021 .006) DETAIL “A” 0.18 (.007) 1.10 (.043) MAX 123456 0.86 (.034) REF SEATING PLANE NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 0.22 – 0.38 (.009 – .015) TYP 0.650 (.0256) BSC 0.1016 (.004 0.0508 .002) MSOP (MSE12) 0608 REV B 3854fa  LTC3854 revision hisTory REV A DATE 10/09 DESCRIPTION Edits to Typical Application Updated Efficiency Graph Edit to Electrical Characteristics and Notes Text Changes to Pin Functions Change to Functional Diagram Updated Related Parts Table PAGE NUMBER 1 1 3, 4 7 8 28 3854fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.  LTC3854 relaTeD parTs PART NUMBER DESCRIPTION LTC3851A/ LTC3851A-1 LTC3878 LTC3879 LTC3850/ LTC3850-1/ LTC3850-2 LTC3853 LTM®4600HV LTM4601AHV LTC3601 LTC3603 LTC3605 LTC3608 LTC3609 LTC3610 LTC3611 LTC3824 LTC3834/ LTC3834-1 LT®3845 No RSENSE™ Wide VIN Range Synchronous Step-Down DC/DC Controller No RSENSE Constant On-Time Synchronous Step-Down DC/DC Controller No RSENSE Constant On-Time Synchronous Step-Down DC/DC Controller Dual 2-Phase, High Efficiency Synchronous Step-Down DC/DC Controllers, RSENSE or DCR Current Sensing and Tracking COMMENTS Phase-Lockable Fixed Operating Frequency, 250kHz to 750kHz, 4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 5.25V, MSOP-16E, 3mm × 3mm QFN-16, SSOP-16 Very Fast Transient Response, tON(MIN) = 43ns, 4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 0.9VIN, SSOP-16 Very Fast Transient Response, tON(MIN) = 43ns, 4V ≤ VIN ≤ 38V, 0.6V ≤ VOUT ≤ 0.9VIN, MSOP-16E, 3mm × 3mm QFN-16 Phase-Lockable Fixed Operating Frequency, 250kHz to 780kHz, 4V ≤ VIN ≤ 30V, 0.8V ≤ VOUT ≤ 5.25V Triple Output, Multiphase Synchronous Step-Down DC/DC Phase-Lockable Fixed Operating Frequency, 250kHz to 750kHz, 4V ≤ VIN ≤ Controller, RSENSE or DCR Current Sensing and Tracking 24V, VOUT Up to 13.5V 10A DC/DC µModule® Complete Power Supply 12A DC/DC µModule Complete Power Supply 1.5A, 4MHz, Monolithic Synchronous Step-Down DC/DC Converter 2.5A, 3MHz, Monolithic Synchronous Step-Down DC/DC Converter 5A, 4MHz, Monolithic Synchronous Step-Down DC/DC Converter 8A, 1MHz, Monolithic Synchronous Step-Down DC/DC Converter 6A, 1MHz, Monolithic Synchronous Step-Down DC/DC Converter 12A, 1MHz, Monolithic Synchronous Step-Down DC/DC Converter 10A, 1MHz, Monolithic Synchronous Step-Down DC/DC Converter Low IQ, High Voltage DC/DC Controller, 100% Duty Cycle Low IQ, Synchronous Step-Down DC/DC Controller Low IQ, High Voltage Synchronous Step-Down DC/DC Controller High Efficiency, Compact Size, Ultrafast Transient Response, 4.5V ≤ VIN ≤ 28V, 0.8V ≤ VOUT ≤ 5V, 15mm × 15mm × 2.8mm High Efficiency, Compact Size, Ultrafast Transient Response, 4.5V ≤ VIN ≤ 28V, 0.8V ≤ VOUT ≤ 5V, 15mm × 15mm × 2.8mm High Efficiency, Phase Lockable, IQ = 300µA, 4V ≤ VIN ≤ 15V, VOUT(MIN) 0.6V, 3mm × 3mm QFN-16, MSOP-16E High Efficiency, Phase Lockable, IQ = 75µA, 4.5V ≤ VIN ≤ 15V, VOUT(MIN) 0.6V, 4mm × 4mm QFN-20 High Efficiency, Adjustable Frequency, 800kHz to 4MHz, 4V ≤ VIN ≤ 15V, VOUT(MIN) 0.6V, 4mm × 4mm QFN-24 High Efficiency, Adjustable Constant On-Time, 4V ≤ VIN ≤ 18V, VOUT(MIN) 0.6V, 7mm × 8mm QFN-52 High Efficiency, Adjustable Constant On-Time, 4V ≤ VIN ≤ 32V, VOUT(MIN) 0.6V, 7mm × 8mm QFN-52 High Efficiency, Adjustable Constant On-Time, 4V ≤ VIN ≤ 24V, VOUT(MIN) 0.6V, 9mm × 9mm QFN-64 High Efficiency, Adjustable Constant On-Time, 4V ≤ VIN ≤ 32V, VOUT(MIN) 0.6V, 9mm × 9mm QFN-64 Selectable Fixed Operating Frequency, 4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ VIN, IQ = 40µA, MSOP-10E Phase-Lockable Fixed Operating Frequency, 140kHz to 650kHz, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, IQ = 30µA, Adjustable Fixed Operating Frequency, 100kHz to 500kHz, 4V ≤ VIN ≤ 60V, 1.23V ≤ VOUT ≤ 36V, IQ = 30µA, TSSOP-16 µModule is a registered trademark of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation. 3854fa  Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● LT 1209 REV A • PRINTED IN USA www.linear.com  LINEAR TECHNOLOGY CORPORATION 2009